This document describes the design of a DC/DC flyback converter project. It details the initial specifications of the converter including an input voltage of 10V DC and adjustable output voltage range of 5-15V DC. It outlines the preliminary calculations done to determine component values for the open-loop design. Simulation results are presented showing the converter can operate in buck and boost modes. The design of the closed-loop controller using a type 2K compensator is described and simulation waveforms are shown verifying stable voltage regulation. Hardware test plans are laid out to characterize the open-loop and closed-loop performance of the built converter.

1.
EE 452 Power Electronics Design
Final Project
Flyback Converter
Section AB
Nasir Elmi 1468579
Daniel Park 1271113
Ki Hei Chan 1368010
December 18, 2015

2. Abstract
We have constructed a DC/DC Flyback Converter, which is able to perform buck
and boost behavior from 10V input to 5­15V output in a open loop circuit.
Additionally, the converter is able to output a 15 V DC in close loop circuit with
varying load magnitude or input voltage magnitude. This document explains the
design process, and compares the capabilities of the final product to the original
specifications.
1

3. Table of Contents:
1. Introduction
1.1. Initial Design Specifications………………………………………….3
2. Pre­Simulation Parameter Calculations
2.1. Open­Loop Design Calculations……………………………………...4
2.2. Average Model………………………………………………………..6
2.3. Type 2K Factor Calculations………………………………………....7
2.4. Closed Feedback Simulation Test…………………………………...10
3. Changes to Specifications
4. Hardware and Circuit Designs
4.1. Flyback Closed­Loop Circuit Schematic…………………………....12
4.2. HPH6­0158L Transformer…………………………………………..13
4.3. SG3425 Controller chip……………………………………………..13
4.4. MOSFET Driver……………………………………………...……..14
4.5. Optocoupler NTE3220……………………………………………....14
5. Open­Loop Hardware Tests
5.1. High Load…………………………………………………………...15
5.2. Low Load…………………………………………………………....16
6. Closed­Loop Hardware Tests
6.1. Varying Load (No Dynamic Changes) ……………………………..17
6.2. Dynamic Load……………………………………………………….19
6.3. Dynamic Input Voltage and Transient Analysis…………………….20
6.4. Improving Transient Analysis……………………………………….21
7. Required Parts
7.1. Cost………………………………………………………………….24
7.2. Engineering Standards………………………...………………..…...25
7.3. Flyback converter Manufacturing…………………………………...25
8. Conclusion and Encountered Problems
2

4. 1 Introduction
The purpose of this final project is to use the theory of the flyback (buck­boost) converter to
build one. This lab report will focus on the preliminary design, which includes calculations and
measurements to extract values and parts necessary for the open­loop and closed­loop design.
With all simulation information intact, the hardware open­loop will be tested and then finally the
hardware closed­loop design will be tested. The goal overall is to simulate a working open and
closed­loop version of the converter and have all the information to build a sufficient closed­loop
version of the flyback converter that meets the specifications.
1.1 Initial Design Specifications
The following are the design specifications/rules that are going to be followed in order to
complete the project. Some specifications might be tweaked to meet the requirements:
Input voltage magnitude: 10 V DC
Adjustable Output voltage magnitude: 5V – 15V DC
Power Rating:
Minimum load: 5 W
Maximum load: 10 W
Maximum output ripple: ± 5%
Efficiency: > 60 % at Maximum load
Converter Frequency: 100 kHz
Output spiking: Less than 25% of Output voltage
Steady state regulation: Less than 3%
Overshoot: Less than 15% of output voltage
Settling time: Less than than 0.3 seconds
Startup transient magnitude: Less than 25% of output voltage
Startup transient duration: Less than 0.4​ ​seconds
3

5. 2 Pre­Simulation Parameter Calculations
2.1 Open­Loop Design Calculations
Figure 1. Open­Loop Flyback Converter Design
For our initial calculations, the inductor value was set first based off of the wattage and the
current of the output. With this, the inductor value that is necessary to be sufficient and operate
at DCM mode was found, which came out to be 12.5 uH. For the capacitance value, this value
was found using the inductor value. From this, C = 2 uF, but for this design we will use a higher
capacitor in order to account for the ripple; thus C = 100 uF was used for the design.
Figure 2. Transient Output Voltage in Buck Settings/Mode
4

6. Figure 2 above shows the buck mode of the converter when the switching voltage is on for 37%
of the time. This 5 DC V shows that for a given range of appropriate resistance values, the
converter will still act like a buck converter. Note: this is not dynamic load varying test; just tests
if the converter can take different loads and output the same voltage.
Figure 3. Transient Output Voltage in Boost Setting/Mode
Figure 3 above shows the boost mode of the converter when the switching voltage is on for 55%
of the time. This 15 constant DC V shows that for a given range of appropriate resistance values,
the converter will still act like boost converter. Note that the zener diode is used to make sure
that the voltage doesn’t exceed 15 V at low loads during boost mode.
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7. 2.2 Average Model
Figure 4. Average Model of the Open­Loop Flyback Converter
In order to obtain values for the controls, the bode plot of the average model from Figure 4 was
used. Figures 5 and 6 shows the magnitude and the phase of the bode plot with cursor at the cross
frequency that will be used in the 2K control calculations in later part of the report.
Figure 5. Magnitude of the Bode Plot of the Average Model at average settings
6

8.
Figure 6. Phase of the Bode Plot of the Average Model at average settings
2.3 Type 2K Factor Calculations
For the 2K Factor Method, we used the following given values from the specifications and
chosen values from the bode plots to calculate the transfer function of the controller circuit. Note
that for every results that follows for each equation is based off of the chosen/given parameters:
Desired Phase Margin: 40 degrees
Cross Frequency: 2990 Hz
Power Stage Gain: 34.114 dB
k Feedback: 1/6
Because the PWM duty cycle is dependent on the voltage at pin 9 of the SG3524 and its pin 9
voltage ranges from specifications are from 1.0 V to 3.5 V, the transfer is as the following:
.4GPWM = 1
3.5−1 = 0
Using these values from the bode plots, the following equation was used to find the phase boost
and the K factor:
PM 80 0ϕboost = − 1 − ϕcross + 9
tan( 5)K = 2
ϕcross
+ 4
With the K factor, we are then able to find the pole and zero frequency using the following
equations:
2πf /KwZero = cr
2πfwPole = cr * K
7

9. Then using the feedback constant, the PWM gain, and the power stage gain, the desired
compensator gain at crossover frequency was found using the following equation (note all of
these terms are linear magnitude gains):
Gcr = 1
k G Gfb PWM PS
Additionally, the controller gain was found using the zero frequency and the desired kc
compensator gain in following equation:
G wkc = cr Zero
Using the calculated parameter values, the transfer function for a 2K was used in the following
equation format:
Gc = s
kc (1+s/wz)
(1+s/wp)
With all the values needed, the two capacitance values and the resistance value of the controller
(shown on Figure x) using the following equations:
C1 = k wc P
g wm Z
C2 = kc
gm
− C1
R = 1
w CZ 2
Figure 7. 2K Factor Controller
With everything calculated and compiled together, the 2K controller is ready to be built into the
closed­loop boost converter design. The following table are the resulting values from the
equation:
8

10. Table 1: Parameter Values from 2K Controller Calculations
Parameter Values
Phase Boost 68 degrees
wZero 3651 rad/s
wZero 96649 rad/s
Desired Gcr 0.2954 (linear gain)
kc 1078
C1 70.1 nF
C1 1.78 uF
R 153.49 Ω
Note: ​DISREGARD ​any of the controller circuit element values in the future figures as they are
just there as placeholders to know where the circuit elements are inserted. Most of the results
were taken using values near the calculated values from small tweaks and trial­and­error.
9

11. 2.4 Closed Feedback Simulation Test
Figure 8. Closed­Loop Average Model of the Flyback Converter
With the controller values computed and obtained, the average model for the closed loop was
built in the simulation and tested. Figures 9 shows the transient response of the closed­loop in
boost mode, which shows that the voltage is constant at 15V for a sufficient amount of time,
proving that the control design is stable.
Additionally, Figure 10 shows the transient response of the closed­loop in buck mode. This also
shows that after a sufficient time, the voltage stays stable at around 5 V, which also proves that
the control design is sufficient.
10

12.
Figure 9. Transient of the Closed­Loop Average Model in Boost Mode
Figure 10. Transient of the Closed­Loop Average Model in Buck Mode.
11

13. 3 Changes to Specifications
With numerous testing, there were some issues that followed and some changes to the design had
to be made. The following changes are on the table below:
Table 2: Changes of some of the specifications
Specification Initial Final
Input Voltage (DC) 10 V 10 V ­ 15 V
Output Voltage (DC) 5 V ­ 15 V 15 V
Converter Frequency 100 kHz 54 kHz
Instead of a range of output voltages, only one output voltage was chosen for the converter to
handle and regulate in closed­loop. Additionally, the converter is made to handle multiple input
voltages to convert to one output voltage with a range of output loads. Finally, the frequency had
to be decreased in order for the MOSFET driver and the opto­isolator to handle the switching.
4 Hardware and Circuit Designs
With all of the simulation data compiled and the necessary information of the converter studied,
the next phase of the project will begin, which is the hardware testing. Below, we will talk about
the schematic of the closed­loop design and then discuss some of the important parts of the
design.
4.1 Flyback Closed­Loop Circuit Schematic
Figure 11: Circuit Schematic of Hardware Implementation
12

14.
Figure 11 shows the baseline of the closed­loop flyback converter. Note that the parts and
element values are subject to change and if so, will be discussed further on in the report.
4.2 HPH6­0158L​ ​Transformer
The calculated transformer inductance that was needed for this project was 12.5 ​µ​H with
turn ratio of 1:1. However, we ordered a transformer with higher inductance of 14.7 µ​H
with 15% leakage inductance. With this leakage, the measured inductance would turn out
to be close to the desired value. During our research for an ideal transformer to our
circuit, there were multiple transformers that would meet inductance specifications, but
they did not meet the other specifications which included current rating and output
voltage range. Thus, we used this transformer in our simulation and test in order to satisfy
our design.
Figure 12.​ ​HPH6­0158 Transformer Design.[1]
4.3 SG3425 Controller chip
Based on our calculation of the average model simulation bode plot, as mentioned earlier,
we needed a type 2 K­factor controller because the phase boost was within 90 degrees.
Then, we agreed to implement the SG3425 controller chip that we used in lab 3 since we
are familiar with the chip functionality with a type 2 K­factor controller. With our design
parameters, the controller is able to adjust the duty ratio to whatever switching frequency
that the circuit needed. This enables our circuit to regulate its output voltage with
differing loads and input voltages.
13

15.
Figure 13. Functional Block Diagram for SG3425 PWM chip
4.4 MOSFET Driver
The PWM output then feeds the MOSFET driver circuit. The driver turns ON and OFF
the MOSFET by applying a high voltage with a relatively low current to the gate
depending on the duty cycle. We tried two different MOSFET driver: the MC34151 and
the UC2710. The MC34151 lacked the ability to switch quickly, which resulted in the
MOSFET on all the time. Thus we went with the UC2710, which was able to take in
higher switching cycles and operate much better than the MC34151 with the circuit as a
whole.
4.5​ ​Optocoupler NTE3220
The purpose of the optocoupler circuit is to provide feedback from the output to the
reference pins of the SG3524. The optocoupler reflect voltage from the secondary to the
primary winding of the transformer while isolating the two sides. This separates the
grounds between the two sides. Thus this devices protects the low side components from
having high power that might transfer from the high side. Thus, we used the NTE3220
optocoupler to control the feedback of our closed loop controller circuit.
Figure 14. Functional Block Diagram for Optocoupler NTE3220
14

16.
5 Open­Loop Hardware Tests
With all the important parts discussed, we then will discuss the open­loop hardware testing.
5.1 High Load
The following oscilloscope waveforms were obtained from testing our circuit and show
how the final product performs. With the first version of the circuit, we were able to meet
many of our desired specifications. The first specification we checked was whether the
flyback could output 15V with 10V input and a 10 W load.
Figure 15. 15V output, 10V input, 22.4Ω load with ripple voltage=0.2 V
Figure 16. 5V output, 10V input, 22.4Ω load with ripple voltage=0.2 V
The test was performed with Vin=10 V, Iin= 1.68 A (Boost), Iin= 0.1751 A (Buck) and 10W
high load (22.4 Ω)
Table 3: High Load Output & Specifications
Output Voltage(V) Power Output(W) Ripple Voltage (%) Efficiency(%)
15

17. 5 .01 1.12 1.42 64
15.20 10.04 1.30 61.5
5.2 Low Load
The following oscilloscope waveforms were obtained from testing our circuit and show how the
final product performs. With the first version of the circuit we were able to meet many of our
desired specifications. The first specification we checked was whether the flyback could output
15V and 5V with 10V input for a 5W load.
Figure 17. 15.1V output, 10V input, 44.98Ω load with ripple voltage=0.2 V
Figure 18: 5V output, 10V input, 44.98Ω load with ripple voltage=0.2 V
16

18. The following test was performed with Vin=10 V, Iin= 0.7947 A (Boost), Iin= 0.0858 A (Buck)
and 5W high load (44.98 Ω) to find the resulting measurements and calculations from Table 4.
Table 4: Low Load Output & Specifications
Output Voltage(V) Power Output(W) Ripple Voltage (%) Efficiency(%)
5 .00 0.56 1.45 64.8
15.10 5.07 1.52 63.8
We met all the initial specification for our open loop circuit as you can see from the above two
tables. Also, we can see that the switching frequency is 54.30 kHz, which is close to the
specified frequency 54 kHz.
6 Closed­Loop Hardware Tests
The following oscilloscope traces were obtained from actual circuit, and show how the final
product performed compared to its specifications. We were able to meet some of the
specifications, yet some of them still need some time to fix it​.
6.1 Varying Load (No Dynamic Changes)
Figure 19: 15.8V output, 10V input, 45.1Ω load with ripple voltage=0.35 V
17

19.
Figure 20: 15.3V output, 10V input, 31.4Ω load with ripple voltage=0.28 V
Figure 21: 14.9V output, 10V input, 22.7Ω load with ripple voltage=0.20 V
The following measurements and calculations of the efficiencies was taken when Vin=10 V
Table 5: Different Load Efficiency
Input Current(A) Load (Ω) Output
Voltage(V)
Power
Output(W)
Efficiency(%)
1.55 22.7 14.9 9.78 63.1
1.24 31.4 15.1 7.45 58.6
18

20. 1.04 45.1 15.3 10.04 50.0
As we can see from the above table, our efficiency was low for one of the results because of the
snubber circuits and the capacitors to deal with the spikes. It is still good comparing to our first
trial where we had our efficiency below 30%. We believed that this low in efficiency was due to
slight error values from the snubber circuit.
6.2 Dynamic Load
We met the dynamic load specification especially when we are changing from high load
to low load quickly.
Figure 22. Voltage regulation with rapid load variation from 10W to 5W
From Figure 22, The rapid load change was from a ​10W ​load ​to a 5W ​load. The output
voltage remained around 15V throughout, with an overshoot of about 1.75V,
approximately ​11.7% which meets our specifications. The settling time was ​39 ms​, of
this rapid change,​ ​which also met our goal.
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21. 6.3 Dynamic Input Voltage and Transient Analysis
Figure 23. Rapid change from 10V to 15V input voltage
Figure 23 shows the dynamic change in input voltage from 10V to 15V. From the
waveforms, the output voltage changed from 15.1 V to 15.0 V, which shows that our
design is able to regulate output well with respect to dynamic changes in the input
voltage.
The first hardware result from the closed loop shows steady state at 15V output, 5W and
an ​12V​ input voltage as shown in Figure 24
Figure 24. 15V output Voltage steady state regulation, 12V input, 5W load
20

22.
As we can see in Figure 24, the output voltage is still 15V even with the different input
voltage value, which demonstrates good steady state voltage regulation.
The second hardware result from the closed loop shows steady state at 15V output, 5W
and an ​13V​ input voltage (Figure 25).
Figure 25. 15V output Voltage steady state regulation, 13V input, 5W load
The third hardware result from the closed loop shows steady state at 15V output, 5W and
an ​15V​ input voltage (Figure 26).
Figure 26. 15V output Voltage steady state regulation, 15V input, 5W load
21

23. The following measurements and calculations were taken when test was performed with
load=44.9 Ω.
Table 6: Dynamic Input voltage Efficiency
Input
Voltage(V)
Input
Current(A)
Load (Ω) Output
Voltage(V)
Power
Output(W)
Efficiency(%
)
12.0 0.72 44.9 15.0 5.01 58.0
13.0 0.74 44.9 15.0 5.01 52.1
15.0 0.80 44.9 15.1 5.1 42.5
As we can see from the above data, our efficiency was low because of the snubber circuits. We
did a lot of improvements to increase our efficiency, which is now close to 60% after using some
of suggestions from the industry reviewers after the presentation.
6.4 Improving Transient Analysis
The next specification we met was the startup transient time after we did our demo. We
tried to improve our start up transient by implementing the industrial and professor
recommendation of getting better snubber values. This process involved getting proper
values from the snubbers. We realized that we needed to match the RC time constant
from the snubbers with the oscillation period of the ringing across the MOSFET and the
diode. Figure 24 below is the original signal. Blue waveform is the voltage across the
MOSFET gate and the yellow waveform is the output voltage.
Figure 27. Output and gate voltage of the circuit at 10W operation (without snubbers)
22

24. Because there are major oscillations for every switching cycle, there are significant
spikes at the output, of the circuit, which can spike up to additional 10V from the 15V
output. With the snubbers implemented, the resulting waveforms are shown Figure 25
(zoomed in more than the one in Figure 24)
Figure 28. Output and gate voltage of the circuit at 10W operation (with snubbers)
With the help of the snubbers, the oscillations during the transition between the on and
off cycle of the circuit has decreased. As a result, the spiking has reduced immensely
from the maximum of 10V peak from the output DC to approximately to 2.5 V peak from
the output DC. This would help meeting the transient requirements, which will be
discussed further on.
Figure 29. 10W Load start up transient time with 15V output and 10 V input voltages
23

25. We can see from Figure 29 that the startup transient time was only 12ms, which was
much faster than anticipated in the specifications (500ms). Additionally, there is barely
any overshoot in the transient, which definitely meets our specifications.
7. Required Parts
7.1 Cost
With all the data compiled, the following table will list out the parts and their cost that will be
needed for the design:
Table 7. Cost for each component of the Flyback Converter Design
Element Type Part Number/Value Unit Price Quantity Total Price
PWM Controller LM3524 1.60 1 1.60
Power Diode MUR420 0.80 1 0.80
Power MOSFET MTP3055VL 1.00 1 1.00
Transformer Coilcraft O4343­BL 4.71 1 4.71
Opto­Isolator PS25001­2 1.70 1 1.70
1/4W 5% Resistors Varying Values 0.10 8 0.80
1/2W 5% Resistors Varying Values 0.10 4 0.40
1 W 1% Resistors 10Ω 0.10 1 0.10
5W Resistors 100Ω 0.50 2 1.00
Potentiometers T34­100k
T34­10k
1.00
1.00
1
1
1.00
1.00
Electrolytic
Capacitor
1000μF
470μF
100μF
22μF
10μF
33μF
0.20
0.20
0.20
0.20
0.20
0.20
1
1
1
1
1
1
0.20
0.20
0.20
0.20
0.20
0.20
Ceramic Disk
Capacitor
100nF
0.00122μF
0.1μF
0.20
0.20
0.10
1
1
1
0.20
0.20
0.10
24

26. From Table 4, the total cost of the design comes out to be $ ​15.81​. The market price is around 30
to 60 dollars for a typical flyback converter integrated chip, which would make our design much
cheaper. However, these market designs have better specifications. Thus, our design output and
specifications fairly reflect the price of creating our design.
7.2 Engineering Standards
Our group met many Engineering and IEEE standards like:
● Citing resource wherever we used to maintain circuit design integrity.
● Having two isolated DC circuits using opto­coupler.
● Temperature Limits in the Rating of Electronics Equipment and for the Evaluation of
Electrical Insulation. In our hardware implementation the MTP3055l Power Mosfet
would get hot.
7.3 Flyback converter Manufacturing
First, the most expensive and important component of the flyback converter is the transformer.
We think that the cost of our flyback converter can be decreased by looking and finding more
different transformers designs. Also, we could find a local distributor for the transformer, which
will decrease the cost of the shipping especially at this time of the year.
Second, our flyback converter will function better if we were able to find a faster MOSFET gate
driver that will work better with higher frequency. On the other hand, we found this problem
after we ordered our parts. This meant that it will take more time to prepare the manufacturing,
which was not in our favor since we had limited time to get the project ready.
Our flyback converter cost is reasonable comparing with the market values. After reducing the
cost as much as possible, we will then find a way to design and manufacturing. We found this
online tool called PCB 123 construction software, which is designed to smooth transition from
design and manufacturing. With this, with sufficient design, we will be able to make the
manufactured design.
8. Conclusion and Encountered Problems:
At the end, our team met many of our initial design specifications, but fell short on some of the
most significant ones. We learned from the process of buying our HPH6­0158L transformer the
importance of looking at datasheet before buying components. We also learned how difficult it
can be to deal with shipping time especially at this time of the year. Despite all of the difficulties
however, we are confident that given one more week, our flyback will meet most of the
25

27. specifications. One of the problems that we had was how to integrate the optoisolator in our
circuit to get the closed­loop working. Finding a new optocoupler, implement it into the circuit,
and ordering will take more than one week because of the holiday season.
If we had a little more time, our circuit would meet the specifications and also be safer. Finally,
this project has taught us all a lot about actual engineering lessons by being more precise with
our specifications, examining all ratings for devices, and presenting us with valuable life
experience that could be applied real world projects.
Overall, we felt that we had success at the beginning by getting our open loop working.
However, our closed­loop only worked for a little bit and overheated to the point where the
circuit stopped working. We then had to replace the MOSFET and add snubber circuit. Even
then, we had high current throughout, resulting in a little bit of overheating and efficiency loss.
Then, we decided to recalculate our snubber circuit values by using different design circuit by
adding diode. We spent hours and days troubleshooting our circuit, finding different parts and
resolving issues with the functionality of our design. Although we had our closed­loop working,
some of the specifications were not met.
26