This document discusses the design philosophy of integrated circuits and biasing techniques. It begins with an overview of constraints in IC design such as minimizing resistors and replacing them with transistors. It then discusses current sources, current mirrors, and current steering circuits which are used to generate constant currents for biasing multiple amplifier stages. The document compares MOSFET and BJT transistors. It explains that current sources use a transistor connected as a diode to generate a constant current, and current mirrors replicate this current. Current steering circuits distribute current from current sources to multiple locations. The document concludes with a brief section on the high frequency response of IC amplifiers.

1. Single Stage IC Amplifier
Mr. A. B. Shinde

2. A. B. Shinde
Contents…
 IC Design philosophy,
 Comparison of MOSFET and BJT,
 IC Biasing: Current sources,
 IC Biasing: Current mirrors and
 IC Biasing: Current steering circuits,
 High frequency response
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3. A. B. Shinde
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IC Design Philosophy

4. A. B. Shinde
IC Design Philosophy
• Constraints and opportunities & features of the IC design philosophy:
• Resistors:
• To minimize the chip area, large and even moderate-size resistors are to
be avoided.
• Transistors can be made small and cheaply, and the use of transistors in
preference to resistors is appreciated.
• As a result, the classical biasing arrangement, is abandoned in IC
amplifiers, rather constant-current sources are implemented with
transistors operating in the active mode for biasing.
• The collector and drain resistors in amplifiers are replaced with constant-
current sources that have much higher incremental resistance.
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5. A. B. Shinde
IC Design Philosophy
• Constraints and opportunities & features of the IC design philosophy:
• Capacitors:
• Chip-area considerations make it impossible to fabricate large-valued
capacitors such as those employed for signal coupling and bypass in
discrete-circuit amplifiers.
• Therefore, IC amplifiers are all direct coupled.
• Small-size capacitors, in the picofarad and fraction-of-a-picofarad range,
are easy to fabricate in IC MOS technology.
• Such capacitors can be combined with MOS amplifiers and MOS
switches to realize a wide variety of analog and digital signal-processing
functions.
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6. A. B. Shinde
IC Design Philosophy
• Constraints and opportunities & features of the IC design philosophy:
• Power Supplies:
• CMOS process technologies are capable of producing devices with a
12-nm channel length.
• To avoid breaking down the thin oxide layers (less than 1 nm) used in
these devices, power supplies are limited to 1 V.
• Low power-supply voltages keeps the power dissipation within
acceptable limits.
• However, the use of such low dc power-supply voltages presents the
circuit designer with a lost of challenges. For instance, MOS transistors
must be operated with overdrive voltages of only 0.1 V to 0.2 V.
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7. A. B. Shinde
IC Design Philosophy
• Constraints and opportunities & features of the IC design philosophy:
• Device Variety:
• The designer of discrete circuits, has limit to use the transistors.
• But, the IC designer has the freedom to specify the device dimensions
and to utilize device matching and arrays of devices having dimensions
with specified ratios.
• For instance, one can utilize an array of bipolar transistors whose
emitter–base–junction areas have binary-weighted ratios.
• CMOS technology provides even more flexibility, with the W and L
values of MOS transistors selected to fit a very wide range of design
requirements.
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8. A. B. Shinde
IC Design Philosophy
• Constraints and opportunities & features of the IC design philosophy:
• Bipolar Technology:
• BJTs are still used in special analog applications, such as high-quality
general-purpose op-amp packages.
• Bipolar circuits can also be combined with CMOS circuits in innovative
and exciting ways in what is known as BiCMOS technology.
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9. A. B. Shinde
IC Design Philosophy
• Constraints and opportunities & features of the IC design philosophy:
• CMOS Technology:
• Currently majority of analog integrated circuits are designed using
CMOS technology.
• This was initially motivated by the need to be compatible with digital
circuits.
• Now, the richness and the versatility that CMOS provides the analog
designers has even stronger reason for its dominance.
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10. A. B. Shinde
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MOSFET Vs BJT

11. A. B. Shinde
MOSFET Vs BJT
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Parameter MOSFET BJT
Types n Channel, p Channel npn, pnp
Output Current is controlled by the input gate
voltage.
is controlled by the input base
current.
ESD Risk Easily damaged by ESD
Electrostatic Discharge.
ESD is rarely a problem

12. A. B. Shinde
MOSFET Vs BJT
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Parameter MOSFET BJT
Gain Very high current gain which is
nearly constant for varying drain
currents.
Lower current gain and it is not
constant. It decreases when the
collector current increases.
Gain increases as temperature
increases.
Input
Resistance
Very high.
For AC signals, much lower due
to the capacitance of the device.
Low
Input Current Picoamps (approximately zero). Microamps or Milliamps
Saturation VDS = 20 mV
Even lower heat dissipation
when saturated.
VCE = 200 mV
Low heat dissipation when
saturated.
Switching
Speed
Faster than Bipolar Slower than MOSFETs.
Frequency
Response
Better frequency response Inferior frequency response

13. A. B. Shinde
MOSFET Vs BJT
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Parameter MOSFET BJT
Voltages When fully turned on, the
potential drop across the device
(VDS) is about 20 mV.
When fully turned on, the
potential drop across the device
(VCE) is about 200 mV.
Bias (input)
Voltages
N Channel MOSFETS need +2
to +4 volts to turn them on.
Base current starts to flow with an
input voltage of about +0.7V
Thermal
Runaway
When MOSFETS heat up, the
current flowing through them
decreases.
They are less likely to be
destroyed by overheating.
When bipolar transistors heat up,
the gain increases and so the
current through them increases
too.
This can cause thermal runaway.
Cost More Expensive Lower Cost

14. A. B. Shinde
MOSFET Vs BJT
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15. A. B. Shinde
MOSFET Vs BJT
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16. A. B. Shinde
MOSFET Vs BJT
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17. A. B. Shinde
MOSFET Vs BJT
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18. A. B. Shinde
MOSFET Vs BJT
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19. A. B. Shinde
MOSFET Vs BJT
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20. A. B. Shinde
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IC Biasing

21. A. B. Shinde
IC Biasing
• On an IC chip with a number of amplifier stages, a constant dc current is
generated at one location and is then replicated at various other
locations for biasing the various amplifier stages through a process
known as current steering.
• This approach has the advantage that the effort expended on generating
a predictable and stable reference current, usually utilizing a precision
resistor external to the chip or a special circuit on the chip, need not be
repeated for every amplifier stage.
• Furthermore, the bias currents of the various stages track each other in
case of changes in power-supply voltage or in temperature.
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22. A. B. Shinde
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IC Biasing:
Current Source Circuit

23. A. B. Shinde
Current Source
• Figure shows the circuit of a simple
MOSFET constant-current source.
• For transistor Q1, the drain is shorted to
its gate, forcing it to operate in the
saturation mode with
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Basic MOSFET
constant current source
The drain current of Q1 is supplied by
VDD through resistor R, which in most
cases would be outside the IC chip.
Since the gate currents are zero,

24. A. B. Shinde
Current Source
• Consider Q2: It has the same VGS as Q1;
thus, if we assume that it is operating in
saturation, its drain current, which is the
output current IO of the current source,
will be
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Basic MOSFET
constant current source
Relation between the output current IO to
the reference current IREF as follows:

25. A. B. Shinde
Current Source
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Output characteristic of the
current source

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IC Biasing:
Current Mirror Circuit

27. A. B. Shinde
Current Mirror
• Here, output current IO is related to the IREF
by the aspect ratios of the transistors i. e.
the relationship between IO and IREF is
determined by the geometries of the
transistors.
• For identical transistors, IO = IREF, and the
circuit simply replicates or mirrors the
reference current in the output terminal.
• Therefore, circuit composed of Q1 and Q2 is
called as current mirror
• It is irrespective of the ratio of device
dimensions.
• Figure shows the current-mirror circuit with
the input reference current shown as being
supplied by a current source.
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28. A. B. Shinde
Current Mirror
• Effect of VO on IO:
• In constant current source circuit, we assumed Q2 is operating in
saturation mode; so as to supply a constant-current output.
• To ensure that Q2 is saturated, the circuit to which the drain of Q2 is to
be connected must establish a drain voltage VO that satisfies the
relationship
VO ≥ VGS −Vtn
or, in terms of the overdrive voltage VOV of Q1 and Q2,
VO ≥ VOV
The current source will operate properly with VO as low as VOV, which is
a few tenths of a volt.
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29. A. B. Shinde
Current Mirror
• Consider, identical devices Q1 and Q2.
• The drain current of Q2, IO, will equal the current in Q1, IREF, at the value
of VO that causes the two devices to have the same VDS, that is, at
VO = VGS.
As VO is increased above this value, IO will increase according to the
incremental output resistance ro2 of Q2.
• In summary, the current source circuit and the current mirror circuit have
a finite output resistance Ro
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where IO is output current and VA2 is the Early voltage of Q2which is
proportional to the transistor channel length.

30. A. B. Shinde
Current Mirror
• Example: Given VDD = 3V and
using IREF =100 μA, design the
circuit shown in figure to obtain an
output current whose nominal
value is 100 μA. Find R if Q1 and
Q2 are matched and have channel
lengths of 1 μm, channel widths of
10 μm, Vt = 0.7 V, and kn = 200
μA/V2. What is the lowest possible
value of VO? Assuming that for this
process technology, the Early
voltage VA =20 V/μm, find the
output resistance of the current
source. Also, find the change in
output current resulting from a
+1 V change in VO.
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31. A. B. Shinde
Current Mirror
• Solution:
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For the transistors used, L = 1 μm. Thus
Therefore
and

32. A. B. Shinde
Current Mirror
The output current will be 100 μA at VO = VGS = 1 V.
If VO changes by +1 V, the corresponding change in IO will be
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33. A. B. Shinde
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IC Biasing:
Current Steering Circuit

34. A. B. Shinde
Current Steering Circuit
• Constant current source once generated can be replicated to provide dc
bias or load currents for the various amplifier stages in an IC.
• This process is known as current steering.
• Current mirrors can also be used to implement these current-steering
function.
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35. A. B. Shinde
Current Steering Circuit
• Here Q1 together with R determine the reference current IREF. Transistors
Q1, Q2, and Q3 form a two-output current mirror,
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current-steering circuit.

36. A. B. Shinde
Current Steering Circuit
• To ensure operation in the saturation region, the voltages at the drains of
Q2 and Q3 are constrained as follows:
VD2,VD3 ≥ −VSS +VGS1 −Vtn
or, equivalently,
VD2,VD3 ≥ −VSS +VOV1
where VOV1 is the overdrive voltage at which Q1, Q2, and Q3 are
operating. In other words, the drains of Q2 and Q3 will have to remain
higher than −VSS by at least the overdrive voltage, which is usually a few
tenths of a volt.
• Here, current I3 is fed to the input side of a current mirror formed by
PMOS transistors Q4 and Q5.
• This mirror provides
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where I4 = I3

37. A. B. Shinde
Current Steering Circuit
• To keep Q5 in saturation, its drain voltage should be
VD5 ≤ VDD −|VOV5 |
where VOV5 is the overdrive voltage at which Q5 is operating.
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38. A. B. Shinde
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High Frequency Response

39. A. B. Shinde
High Frequency Response
• The amplifier circuits do not employ bypass capacitors.
• The various stages in an integrated-circuit cascade amplifier are directly
coupled; that is, they do not utilize large coupling capacitors.
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The frequency response of these direct-coupled or d c amplifiers takes the
general form shown in figure, where gain remains constant at midband
value AM down to zero frequency (dc).
The gain, however, falls off at the high-frequency end due to the internal
capacitances of the transistor.

40. A. B. Shinde
High Frequency Response
• High-Frequency Gain Function
• The amplifier gain, taking into account the internal transistor
capacitances, can be expressed as a function of the complex-frequency
variables in the general form
A(s) = AM FH (s)
where AM is the midband gain.
• The value of AM can be determined by analyzing the amplifier equivalent
circuit while neglecting the effect of the transistor internal
capacitances—that is, by assuming that they act as perfect open
circuits.
• By taking these capacitances into account, the gain acquires the factor
FH(s), which can be expressed in terms of its poles and zeros, which are
usually real, as follows:
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41. A. B. Shinde
High Frequency Response
• Determining the 3-dB Frequency fH:
• In amplifier high-frequency band that is close to the midband is more
important because the designer needs to estimate – and if needed
modify the value of the upper 3-dB frequency fH (or ωH; fH = ωH/2π).
• In many cases the zeros are either at infinity or such high frequencies as
to be of little significance to the determination of ωH.
• If in addition one of the poles, say ωP1, is of much lower frequency than
any of the other poles, then this pole will have the greatest effect on the
value of the amplifier ωH.
• In other words, this pole will dominate the high-frequency response of
the amplifier, and the amplifier is said to have a dominant-pole response.
• In such cases the function FH(s) can b e approximated by
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which is the transfer function of a first-order low-pass network.

42. A. B. Shinde
High Frequency Response
• Determining the 3-dB Frequency fH:
• A dominant pole exists if the lowest-frequency pole is at least two
octaves (a factor of 4) away from the nearest pole or zero.
• If a dominant pole does not exist, the 3-dB frequency ωH can be
determined from a plot of |FH (jω)|.
• An approximate formula for ωH can be derived as follows:
Consider, for simplicity, the case of a circuit having two poles and two
zeros in the high-frequency band; that is,
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Substituting s = j ω and taking the squared magnitude gives

43. A. B. Shinde
High Frequency Response
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Since ωH is usually smaller than the frequencies of all the poles and
zeros, we may neglect the terms containing ω4
H and solve for ωH to
obtain
Determining the 3-dB Frequency fH:

44. A. B. Shinde
High Frequency Response
• Determining the 3-dB Frequency fH:
• This relationship can be extended to any number of poles and zeros as
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45. A. B. Shinde
Unit-III: Single Stage IC amplifiers
• CS and CF amplifiers with loads,
• High frequency response of CS and CF amplifiers,
• CG and CB amplifiers with active loads,
• High frequency response of CG and CB amplifiers,
• Cascade amplifiers.
• CS and CE amplifiers with source (emitter) degeneration source and
emitter followers,
• Some useful transfer parings,
• Current mirrors with improved performance.
• SPICE examples.
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46. This presentation is published only for educational purpose
abshinde.eln@gmail.com