Chapter 5 fm receivers

FIGURE 5.1 Double-conversion FM receiver block diagram Basically - - - > similar to AM receivers Double-Conversion Superheterodyne FM Receivers

Prevent mixer saturation when strong RF signals are received 8) AGC Remove info signal from FM wave 7) Detector/demodulator Clipping amplitude varied (noise) 6) Limiter Provide gain & sensitivity 5) IF Amplifier Down converts 1 st IF to 2 nd IF * Normally 2 nd IF low - - > 455 KHz 4) 2 nd mixer/converter Down converts RF to 1 st IF * Normally 1 st IF high - - > 10.7 MHz 3) 1 st mixer/converter Establish SNR & NF 2) RF Amplifier Reject f image 1) Preselector Main Function Main Stage / Block

FM DEMODULATION Demodulation should provide an output signal whose amplitude is dependent on the instantaneous carrier frequency deviation and whose frequency is dependent on the rate of the carrier frequency change Figure 5.2 : FM Characteristics curve

Slope Detector simplest form of tuned-circuit frequency discriminator Single ended slope has the most nonlinear voltage-versus-frequency characteristics  seldom used Convert FM to AM - - > demodulate AM envelope with conventional peak detectors Circuit consist of a tuned circuit and an AM detector Disadvantages Gain is reduced Difficult to achieve a linear slope response curve

Figure 5.3: Slope detector (a) schematic diagram (b) voltage-versus-frequency curve. Linear portion AM out AM peak detector

Circuit Operation (Figure 5.3) L a & C a (tuned circuit) produce o/put voltage (amplitude varies) which is proportional to the i/put freq. (FM in) - - -> AM characteristic Maximum o/put voltage occurs at resonant freq of tank circuit, f o and its o/put decrease proportionately as the i/put freq deviates below & above f o IF center frequency (f c ) falls in the center of the most linear portion of the voltage-versus-frequency curve (Figure 5.3(b)) When IF deviates above f c , output voltage increase and when IF deviated below f c , output voltage decrease. The tuned circuit converts frequency variations to amplitude variations (FM-to-AM conversion). D i , C i and R i - - -> simple peak detector that converts amplitude variations to o/put voltage (operate as AM Diode Detector) o/put voltage varies at a rate equal to i/put frequency Amplitude of o/put voltage depend on magnitude of freq changes

Balanced slope detector Tuned circuit Balanced peak detector f a > f c f a < f c Figure 5.4: Balanced slope detector (a) schematic diagram (b) voltage-versus-frequency response curve.

made up of single-ended slope detector connected in parallel and fed 180 o out of phase (phase inversion). the 2 tuned circuit perform the FM to AM conversions At resonant freq, the resultant output voltage is 0 V. As the IF deviates above the center freq, top tuned circuit produces higher voltage than the lower tank circuit (+Vout) and vice versa. Figure 5.4(b) shows the output versus frequency response curve. diode detector circuit (D 1 ,R 1 ,C 1 & D 2 ,R 2 ,C 2 ) recover the original signal. advantage : simple. disadvantages : poor linearity, difficult in tuning and lack of provisions for limiting.

Foster Seeley Discriminator Also called phase shift discriminator (tuned-circuit frequency discriminator)  operation very similar to the balanced slope detector Is tuned by injecting a frequency equal to the IF center frequency and tuning C 0 for 0 volts out. Output voltage is directly proportional to the magnitude and direction of the frequency deviation. Output voltage-versus-frequency deviation curve is more linear than that of a slope detector because there is only one tank circuit, - - -> easier to tune. For distortion less demodulation, the frequency deviation should be restricted to the linear portion of the secondary tuned-circuit frequency response curve. Responds to amplitude as well as frequency variations and, therefore, must be preceded by a separate limiter circuit

Figure 5.5: Foster Seeley discriminator (a) schematic diagram (b)vector diagram, f in = f o ; (b) f in > f o ; (c) f in < f o

Principle Circuit Operation Have 2 tank circuits : i) L p & C p (primary side of T1) ii) L a ,L b & C o (secondary side of T1) C c , C 1 & C 2 are chosen so that short circuits for the IF centre frequency IF signal is fed directly (in phase) across L3 via center tapping of T1 - - - > IF voltage, V p appears directly across L3 with no phase inversion V p ∠ θ o <= = > V L3 ∠ θ o Incoming IF signal is inverted 180 o by T1 and divided equally between V La & V Lb V La ∠ θ o <= = > V Lb ∠ θ o + 180 o Vs (secondary winding of T1, V La + V Lb ) have 180 o phase inversion with Vp because characteristic of centre tapping of T1 Vp ∠ θ o <= = > Vs∠ θ o + 180 o Both tuned exactly to the IF centre frequency V p V s V La V Lb

6. Is (secondary winding T1) always have 90 o phase inversion with V La & V LB Is ∠ θ o <= = > VLb ∠ θ o ± 90 o & VLb ∠ θ o ± 90 o 7. V D1 is the vector sum of V L3 & V La V D2 is the vector sum of V L3 & V Lb Previously, know that V L3 is fed directly by Vin - - - > same phase & value. 8. If input frequency ( f in ) same with resonant freq of the secondary tank circuit (IF centre freq), I s is in the phase with total secondary voltage (V s ) : f in = f o (IF centre frequency) : C 1 & C 2 charge to equal magnitude voltage but opposite polarities : V D1 & V D2 will have equal voltages : V out = V C1 - V C2 = 0 The phase relation can be represented as Figure 5.5 (b) 9. If IF goes above resonance (X L > X C ), tank circuit impedance become inductive & I s lags the V s by some angle, θ ’ o which is proportional to the magnitude of the ∆ f : f in > f o (incoming IF signal freq > IF centre freq) : C 1 charges & C 2 discharges : V D1 > V D2 (sum vector of V D1 > sum vector of V D2 ) : V out = V C1 – V C2 = +ve value The phase relation can be represented as Figure 5.5 (c) V La V Lb Is V D2 V La V Lb Vin V D1 V s I s Θ ’ o

10. If IF goes below resonance (X L < X C ), I s lead the V s by some angle, θ ’ o which is proportional to the magnitude of the ∆ f : f in < f o (incoming IF signal freq < IF centre freq) : C 1 discharges & C 2 charges : V D1 < V D2 : V out = V C1 – V C2 = -ve value The phase relation can be represented as Figure 5.5 (d) 11. Figure 5.6 shows a typical voltage-versus-frequency response curve for a Foster-Seeley discriminator, called an S-curve. Figure 5.6 : Discriminator voltage-versus-frequency response curve. V s I s Θ ’ o

Ratio detector Advantage : relatively immune to amplitude variations in its input signal. Figure 5.7 shows the schematic diagram for a ratio detector. Same as the Foster-Seeley discriminator but with 3 limiting changes. D 2 has been reversed  current (I d ) flow through the outermost loop of the circuit Shunt capacitor, C s charges to approximately the peak voltage across the secondary winding of T 1 . The reactance of C s is low, and R s simply provides a dc path for diode current. Therefore, the time constant for R s and C s is sufficiently long so that rapid changes in the amplitude of the input signal due to thermal noise or other interfering signals are shorted to ground and have no effect on the average voltage across C s .

Figure 5.7 : Ratio detector (a) schematic diagram; (b) voltage-versus-frequency response curve

Consequently, C 1 and C 2 charge and discharge proportional to frequency changes in the input signal and are relatively immune to amplitude variations. Also, the output voltage from a ratio detector is taken with respect to ground, and for the diode polarities shown in Figure 5.7(a), the average output voltage is positive. At resonance, the output voltage is divided equally between C 1 and C 2 and redistributed as the input frequency is deviated above and below resonance. Therefore, changes in Vout are due to the changing ratio of the voltage across C 1 and C 2 , while the total voltage is clamped by C s . Figure 5.7(b) shows the output frequency response curve for the ratio detector shown in Figure 5.7(a). It can be seen that at resonance, Vout is not equal to 0 V but, rather, to one-half of the voltage across the secondary windings of T 1 . Because a ratio detector is relatively immune to amplitude variations, it is often selected over a discriminator. However, a discriminator produces a more linear output voltage-versus-frequency response curve.

PHASE LOCKED LOOP (PLL) FM demodulation can be accomplished quite simply with a phase-locked loop (PLL). PLL FM demodulator is probably the simplest and easiest to understand. A PLL frequency demodulator requires no tuned circuits and automatically compensates for changes in the carrier frequency due to instability in the transmit oscillator. Figure 5.8(a) shows the simplified block diagram for a PLL FM demodulator.

Figure 5.8 (a) Block diagram for a PLL FM demodulator.

If the IF amplitude is sufficiently limited prior to reaching the PLL and the loop is properly compensated, the PLL loop gain is constant and equal to K V . after frequency lock had occurred the VCO would track frequency changes in the input signal by maintaining a phase error at the input of the phase comparator. Therefore, if the PLL input is a deviated FM signal and the VCO natural frequency is equal to the IF center frequency, the correction voltage produced at the output of the phase comparator and fed back to the input of the VCO is proportional to the frequency deviation and is, thus, the demodulated information signal. Therefore, the demodulated signal can be taken directly from the output of the internal buffer and is mathematically given as V out = Δ f K d K a

Figure 5.8(b) shows a schematic diagram for an FM demodulator using the XR-2212. R 0 and C 0 are course adjustments for setting the VCO's free-running frequency. R x is for fine tuning, and RF and R c set the internal op-amp voltage gain (K a ). The PLL closed-loop frequency response should be compensated to allow un-attenuated demodulation of the entire information signal bandwidth. The PLL op-amp provides voltage gain and current drive stability PLL is the best frequency demodulator, because the filtering circuit removes noise and interference and its linear output reproduce the output signal

Figure 5.8 (b) PLL FM demodulator using the XR-2212 PLL

LIMITER If noise or other interference perturbs the amplitude of the carrier, then the carrier amplitude variations will cause distortion at the output. This distortion can be removed by passing through a limiter prior to the differentiator (demodulator) The limiter output must then be converted to a sinusoid by passing it through a band-pass filter with center frequency and sufficient bandwidth to pass the varying fundamental. The result is a constant amplitude sinusoid, which is differentiated and passed through the envelope detector to produce the desired signal. Improve S/N as much as 20 dB.

Figure 5.9 Amplitude limiter input and output waveforms: (a) input waveform; (b) output waveform

FIGURE 5.10 Limiter output: (a) captured by noise; (b) captured by signal.

FM STEREO TRANSMITTER Broadcast of high fidelity stereo began in 1961 of FM radio The transmission of 2 channels of sound information (stereophonic) on a single carrier require compatibility with existing high-fidelity monophonic FM receiver FDM is used to combine the two audio Figure 5.11

FIGURE 5.12: Stereo FM transmitter using frequency-division multiplexing

From Figure 5.12 Both L & R channel will be added to the 2 adder circuits 1 st adder : (L + R) channel (50Hz ~ 15KHz) - -> total mono audio signal for compatibility 2 nd adder : (L – R) channel (50Hz ~ 15KHz) R channel is inverted and then added to adder circuit Delay network (L – R) channel have delay which is introduced by signal path as it propagates via the balanced modulator (L + R) channel must maintain its phase integrity with (L – R) channel to prevent phase error at further process. Balanced modulator AM DSB-SC which modulate (L – R) channel with frequency subcarrier 38KHz Produce ( L – R) sidebands with bandwidth 23KHz ~ 53KHz Multiplier x2 Multiply 19KHz oscillator by 2 to produce 38KHz subcarrier Linear combining network Combine (L + R) channel, (L – R) channel, 19KHz stereo pilot and SCA (Subsidiary Communication Authorization) 60KHz ~ 74KHz Produce composite baseband to be modulated with main station carrier SCA is used to broadcast uninterrupted music to private subscriber (not compulsory to be combined by fundamental of FM Broadcast)

The 19kHz pilot frequency fits between (L + R) and DSBSC (L – R) signals in the baseband frequency spectrum All three signals are added together to form the composite stereo broadband signal, which will modulate the main station carrier The pilot frequency is made small enough so that its FM deviation of the carrier is only ≈ 10% of the total ±75kHz maximum deviation allowed After the FM stereo is received and demodulated to baseband, the 19kHz pilot is used to phase lock an oscillator, which provides the 38 kHz subcarrier for the demodulation of the (L – R) signal.

FM Baseband Spectrum FIGURE 5.13 : Composite baseband spectrum L-R LSB L+R 50Hz 15KHz L+R Stereo Channel Stereo Pilot L-R Stereo Channel Subcarrier L-R USB 23KHz 37.95 KHz 38 KHz 38.05 KHz 53 KHz 19 KHz SCA 60KHz 74KHz

FM STEREO RECEIVER All new FM broadcast receivers are being built with provision for receiving stereo, or two-channel broadcasts. The left (L) and right (R) channel signals from the program material are combined to form two different signals, one of which is the left-plus-right signal and one of which is the left-minus-right signal The (L - R) signal is double-sideband suppressed carrier (DSBSC) modulated about a carrier frequency of 38 kHz, with the LSB in the 23- to 38-kHz slot and the USB in the 38- to 53-kHz slot. The (L + R) signal is placed directly in the 0- to 15-kHz slot, and a pilot carrier at 19 kHz is added to synchronize the demodulator at the receiver. The output from the FM detector is a composite audio signal containing the frequency-multiplexed (L + R) and (L - R) signals and the 19-kHz pilot tone. This composite signal is applied directly to the input of the decode matrix.

The composite audio signal is also applied to one input of a phase-error detector circuit, which is part of a phase locked loop 38-kHz oscillator. The output drives the 38-kHz voltage-controlled oscillator, whose output provides the synchronous carrier for the demodulator. The oscillator output is also frequency divided by 2 (in a counter circuit) and applied to the other input of the phase comparator to close the phase locked loop. The phase-error signal is also passed to a Schmitt trigger circuit, which drives an indicator lamp on the panel that lights when the error signal goes to zero, indicating the presence of a synchronizing input signal (the 19-kHz pilot tone). The outputs from the 38-kHz oscillator and the filtered composite audio signals are applied to the balanced demodulator, whose output is the (L - R) channel.

The ( L + R) and (L - R) signals are passed through a matrix circuit that separates the L and R signals from each other. These are passed through de-emphasis networks and low-pass filters to remove unwanted high-frequency components and are then passed to the two channel audio amplifiers and speakers. On reception of a monaural signal, the pilot-tone indicator circuit goes off, indicating the absence of pilot tone, and closes the switch to disable the (L - R) input to the matrix. The (L + R) signal is passed through the matrix to both outputs. An ordinary monaural receiver tuned to a stereo signal would produce only the (L + R) signal, since all frequencies above 15 kHz are removed by filtering, and no demodulator circuitry is present. Thus the stereo signal is compatible with the monaural receivers.

FIGURE 5.15 FM stereo and mono receiver

Chapter 5 fm receivers

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Chapter 5 fm receivers