The document describes a high-resolution time-to-digital converter (TDC) implemented in a 90nm CMOS process. It uses two gated ring oscillators (GROs) as delay lines in a Vernier structure to achieve both a high time resolution and first-order noise shaping. The TDC achieves less than 10ps coarse resolution, consumes 3.6mA from a 1.2V supply, and occupies an active area of 0.18mm by 0.15mm. Simulation results show the quantization noise is pushed to high frequencies, improving the in-band noise performance for applications like all-digital phase-locked loops.

1. A High-resolution Vernier Gated-Ring-Oscillator TDC
in 90-nm CMOS
Ping Lu, Pietro Andreani
Department of Electrical and Information Technology, Lund University, Sweden
Email: {Ping.Lu, Pietro.Andreani }@eit.lth.se
Abastract—A Vernier Gate-Ring-Oscillator (GRO) Time
to Digital Converter (TDC) is proposed and implemented
in 90-nm CMOS process technology. It utilizes two GRO
chains as the delay lines. The time resolution is
determined by the difference between two delays, so not
limited by the process. Moreover, the quantization noise
can be first-order shaped by the gated behavior in the
oscillators, which further improves the in-band TDC noise
contribution for an ADPLL. Operating at 1.2-V supply
with 250MHz clock, the chip achieves a less-than-10ps
coarse resolution (varies with digital control bits) and
consumes only 3.6-mA.
INTRODUCTION
All Digital Phase-Locked Loop (ADPLL) is now a hot
research topic for more flexibility and programmability Fig.1 GRO principle and noise shaping
compared with its analog counterpart. In an ADPLL,
Digitally Controlled Oscillator (DCO) phase noise and then developed [5].
TDC quantization noise dominate the in-band and Noise shaping is another method for reducing in-band
out-band noise respectively. Thus in some ADPLL TDC noise contribution [6]. This type of TDC has still a
applications which need a wide bandwidth, the coarse resolution of an inverter delay, but the
quantization noise of TDC is important to the total noise corresponding large quantization noise can be first-order
performance. shaped. It pushes most of the noise to high-frequency
The TDC noise contribution, within the loop bandwidth, region which is then filtered by the loop filter in ADPLLs.
at the ADPLL RF output is [1] As shown in Fig.1, the cyclic behavior of GRO helps
(2π ) 2 Δtdelay 2 1 reuse the staggered clocks circularly and the enable
STDC = ( )
12 TDCO f REF transistors (NMOS and PMOS) in the delay cell hold the
(1)
oscillation node state between measurements. At next
where Δtdelay denotes the delay time of an delay cell in the
measurement interval, the oscillator starting phase
TDC chain, TDCO denotes the period of RF output, and fREF
corresponds to the stopping phase of the current
is frequency of the reference clock. As can be seen, a
measurement interval. The variable starting phase
smaller Δtdelay leads to a smaller TDC quantization noise.
effectively scrambles the quantization noise across the
In a traditional TDC where the time resolution provided
different measurement intervals and also introduces a
by a self-loaded inverter, the quantization noise is limited
first-order noise shaping.
by a single inverter, so is limited by a process technology.
Considering that the coarse resolution in GRO based
Some advanced approaches like vernier structure [2],
TDC is still limited by an inverter delay, a new TDC
pulse shrinking [3] and sub-exponent TDC [4] have
combining vernier and GRO structures is proposed in this
therefore been proposed to achieve much shorter
work. It uses two GROs in a classic vernier TDC to
resolvable time than an inverter itself.
achieve both process-independent high time resolution and
A vernier TDC uses two delay lines with respective
first-order shaped noise characteristic.
delay of τ 1 and τ 2 instead of a single line. The time This paper is arranged as follows. Section II describes
resolution Δtdelay becomes a delay difference τ 1 − τ 2 , not the architecture of the vernier GRO based TDC. The
an inverter delay any more. This delay difference can be circuit implementation is detailed in Section III. Section
very small (down to 0) theoretically regardless of IV gives the layout and simulation results. Conclusions are
thermo-noise of devices. Thus TDC noise contribution can drawn in Section V.
be reduced greatly according to (1). Since the time
VERNIER GATED-RING-OSCILLATOR TDC
resolution is determined by a very small delay difference
in the vernier structure, a large number of inverter stages Fig.2 shows the structure of the proposed TDC. It
are required to cover a large detection range, incurring includes the GRO core, Phase and Frequency Detector
large silicon overhead. A ring-oscillator vernier TDC was (enable generator) and a multi-clock counter.
978-1-4244-8971-8/10$26.00 c 2010 IEEE

2. Fig.2 Block Diagram of proposed vernier GRO based TDC
The GRO core consists of two 9-stage ring oscillators rising edges in SGRO will lag their corresponding rising
and a sampling block. As in a vernier TDC, the two edges in FGRO after passing a certain amount of
oscillators have different delay cells. The delay of the first staggered phases. Once the lag happens, RBO will reset
oscillator cell is a little longer than that of the second one. PFD and the measurement window will be closed
The difference between them indicates the time resolution. immediately (EN_REF/ EN_CKV = ”0”).
Use SGRO (slow) and FGRO (fast) to denote those two
CIRCUIT DETAILS
oscillators. EN_REF and EN_CKV are enable signals for
them respectively. The oscillators work at the enable state A. Gated-Ring-Oscillator core
(EN_REF/ EN_CKV=”1”) and hold their states at the Fig.3 shows the detailed GRO core. Two identical ring
disable state (EN_REF/ EN_CKV=”0”). The sampling oscillators are used. Delay difference comes from different
block reads the phase information of the 9 pairs of control inputs (c<1:15>) to them. EN and ENB are enable
one-one correspondence clocks and feeds back a control signals for NMOS and PMOS gate transistors in delay
signal. At initialization, SGRO always leads FGRO and cells. c<1:15> are thermometer-code bits which are
the sampled value RBO is “0”. When any phase in SGRO connected to 15 unit MOS capacitors directly. The delay
lags its counterpart phase in FGRO, RBO sends a positive gain is about 1.28ps/bit. For a vernier TDC application,
pulse to reset PFD. Then EN_REF and EN_CKV return to the inputs c<1:15> are always different for SGRO and
“0” to hold the oscillators’ state. All clock (ck<0:8>) FGRO because a small time difference is needed to
edges from SGRO are also used by a multi-clock counter quantize phase error.
to calculate the digitized time error in each measurement The GRO delay cell is shown at the top right corner in
interval. Since the gate transistors are used in the GROs, Fig.3. NMOS and PMOS enable transistors are controlled
quantization error residue will be accumulated in each by EN and ENB respectively. When both enable transistors
measurement. That means a first-order noise shaping is are in the conducting condition, the delay cell works like a
achieved at the same time, as shown in Fig.1. traditional inverter. Then the inverter ring oscillates as a
It should be noticed that in this work only rising edge of common ring oscillator. However, when EN and ENB cut
all oscillator clocks is used for the sampling block to get off the enable transistors, the ring will stop oscillating and
rid of rising and falling edge mismatch. It suggests the keep the output nodes’ charge unchanged because no
coarse time resolution is double difference between two current path is connected to power supply or ground any
inverter delays because non-inversion neighboring phases more. The oscillation frequency of each GRO can be tuned
are preferred to use. discretely by 15 thermometer-code switches c<1:15>. For
Multi-clock counter needs to count at each rising edge SGRO, these 15 switches are always connected to ground
of ck<0:8>. The count enable signal is SGRO’s gate suggesting a fixed cell delay (also a fixed frequency).
control EN_REF. When EN_REF = ”1”, each rising edge Then the delay difference between SGRO and FGRO are
triggers the accumulator to add “1” to the sum. When determined only by the 15 switches in FGRO. Actually, to
EN_REF = ”0” (the measurement is finished), the counter reduce off-chip controls, the 15 thermometer-code
saves the sum into a latch and then resets the accumulator switches are generated by an additional
preparing for the next measurement. binary-to-thermometer coder.
PFD generates two enable signals (EN_REF, EN_CKV) Sampling flip-flops detect the phase error between each
and their reversal signals (ENB_REF, ENB_CKV) to pair of staggered clocks. Once one of those sampled value
control NMOS and PMOS enable transistors in GRO becomes “1” (that is the staggered lagged clocks catch up
delay cells. EN_REF/EN_CKV is supposed to be a with those leading ones), RBO becomes “1” to reset PFD.
rectangular signal suggesting the phase error information The key point for GRO core is the absolute delay in
between input clocks REF and CKV. When REF rising each oscillator. When phase ck<0(n+1)> catches up with
edge arrives first, for example, EN_REF becomes “1” phase ck<n+1>, as illustrated in lower right of Fig.3,
(ENB_REF becomes “0”) forcing SGRO to start o<n+1> needs transmission time to become “1”. The
oscillating first. When CKV rising edge arrives after a situation for RBO and EN_REF/EN_CKV is the same. Due
while, EN_CKV also becomes “1” forcing FGRO to work. to this kind of gate delay, two GROs keep oscillating for a
In this example, EN_REF leads EN_CKV. It means that short while after ck<n+1> is already lagged. The critical
the first rising edge in SGRO leads the first rising edge in feedback path has a typical delay of 100ps~200ps
FGRO during the measurement interval. If these two (obtained by post layout simulation). No more rising edges
oscillators have the same frequency, the phase error will are supposed to be generated during this additional time
be always kept. However, due to the delay difference, the interval, otherwise wrong sampling may happen in the

3. Fig.3 GRO core and sampling/reset delay
Fig.4 Adaptive PFD
next measurement. That means the absolute delay of each
vernier line (two-inverter delay) should be longer than
200ps. Here 400ps is adopted for some margin.
B. Phase & frequency detector (PFD)
In the GRO core, the SRGO (driven by EN_REF) cell
delay is always longer than FGRO (driven by EN_CKV)
cell delay. It requires EN_REF to lead EN_CKV always.
However for a normal PFD in which EN_REF/EN_CKV is
corresponding to REF/CKV, this precondition cannot be
guaranteed because CKV frequency and phase are adjusted
dynamically in ADPLLs. A SR-latch is therefore used as
an arbiter, as shown in Fig.4. It senses the leading edge
between REF and CKV and gives a sign signal which
Fig.5 Multi-clock counter
controls four multiplexers. When REF is leading, sign is
“0” meaning EN_REF corresponds to REF. On the
clock frequency and hardware consumption. In each group,
contrary, EN_REF corresponds to CKV.
clocks should have a large interval between rising edges,
C. Multi-clock counter which eases the strain of accumulator. The two-inverter
The digitized phase error can be calculated using delay is about 400ps in this work. To make the counter
multiple clocks ck<0:8> from SGRO in each measurement work with a large margin, the nine clocks are grouped into
interval. One approach is to assign one individual 5 groups with each containing two. The two clocks with
accumulator for each clock, which consumes more adders the longest rising interval (e.g. ck<0> and ck<8>) share
and flip-flops. If all clocks could be combined to one that one accumulator. Each clock rising edge produces a
contains all edges, only one accumulator is enough. But narrow pulse by inserting a delay in the reversion-input
the new clock frequency is inevitably very high bearing on path of the first NAND gate, as shown in Fig.5. Then the
the accumulator speed. To tackle the above problem, we second NAND gate combines these two positive pulses to
propose to group those clocks to get a tradeoff between one to clock an accumulator. The sum of each accumulator

4. multi-clock counter
GRO1
PFD
GRO2 (a)
dummy load
Fig.6 Layout of TDC
(b)
Fig.8 Simulated transistor-level TDC output
(a) (b) (a) Bit stream (coded decimal) (b) FFT analysis
Fig.7 Simulated PSD of TDC output in Simulink
(a) Proposed TDC (b) Classic vernier TDC quantization noise due to coarse resolution is well
first-order shaped (see Fig.8(b)).
should be synchronized and then added together. The
above operation runs only in the measurement interval CONCLUSIONS
(EN_REF = ”1”). This paper describes a Vernier-GRO TDC which uses
LAYOUT AND SIMULATION vernier principle to get a high time resolution and GRO’s
to achieve first-order shaped quantization noise. The
The above presented TDC has been implemented using first-order noise shaping pushes much noise to the
90-nm CMOS technology. Fig.6 shows the layout of the high-frequency region which can be filtered by a low-pass
proposed TDC (without I/O pads). The active area is noise transfer function in an ADPLL. The chip is
0.18mm*0.15mm. The multi-clock counter uses staggered implemented in 90-nm CMOS technology and occupies
clocks from SGRO as referred above. Since FGRO has no 0.18mm*0.15mm area.
identical load capacitances as SGRO, some dummy gate
loads are used to compensate delay mismatch. REFERENCES
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