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RF Module Design - [Chapter 4] Transceiver Architecture

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Transceiver Architecture

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RF Module Design - [Chapter 4] Transceiver Architecture

  1. 1. RF Transceiver Module Design Chapter 4 RF Transceiver Architectures 李健榮 助理教授 Department of Electronic Engineering National Taipei University of Technology
  2. 2. Outline • General Considerations • Frequency Conversion • Receiver Architectures Heterodyne Receiver Direct-Conversion Receiver (DCR) Image-Reject and Low-IF Receiver • Transmitter Architectures Direct-Conversion Transmitter (DCT) Heterodyne and Sliding-IF Transmitter Open-loop and Closed-loop PLL-based Transmitter Envelope Tracking and Envelope Following Transmitter Polar Transmitter Department of Electronic Engineering, NTUT2/110
  3. 3. Front-end General Considerations • TX: Adjacent channel leakage • RX: Rejection of inband and out-of-band interference BPF Power Amplifier (PA) Transmitted Channel Adjacent Channels ω BPF Low Noise Amplifier (LNA) Bandpass Filter Response Adjacent Channel Alternate Adjacent Channel 1f f Department of Electronic Engineering, NTUT3/110
  4. 4. Interferer Suppression • High linearity to accommodate interferes without experiencing compression or significant intermodulation. Filtering the interferer can relax RX linearity requirements. • BPF high selectivity is required for near channel rejection. • Variable BPF is required for different carrier frequencies, and it is difficult. 900 900.4 ( )MHzf 20 dB 35 dB BPF Response Hypothetical filter to suppress an interference Department of Electronic Engineering, NTUT4/110
  5. 5. Channel-Selection Filter • All of the stages in the RX chain that precede channel- selection filtering must be sufficiently linear to avoid compression or excessive intermodulation • Since channel-selection filtering is extremely difficult at the input carrier frequency, it must be deferred to some other point along the chain where the center frequency of the desired channel is substantially lower and hence the required filter Q’s are more reasonable. Department of Electronic Engineering, NTUT5/110
  6. 6. Band-Select Filter • A band-select filter selects entire RX band and reject out-of- band interferers, thereby suppressing components that may be generated by users that do not belong to the standard of interest. • Trade-off between selectivity and in-band loss (higher-order filtering sections and arise NF). BPF LNADesired Channel Receive Band f f Band-selection filtering Department of Electronic Engineering, NTUT6/110
  7. 7. TX-RX Feedthrough • TX leakage in a CDMA transceiver (full duplex). The RX must meet difficult linearity requirements. • A BPF following the LNA can alleviate the leakage. Duplexer −20 dBm LNA PA 1 W (+30 dBm) −50 dB Duplexer LNA PA −50 dB f f TX Leakage f BPF Response BPF 10dB/div. 20 MHz/div. 1f2f TX Band RX Band 50 dB 30 dB Department of Electronic Engineering, NTUT7/110
  8. 8. Frequency Conversion (I) • Recall Chapter 1 (double sideband amplitude modulation) ( ) ( )cos2m cs t A t f tπ=t( ) ( )BBs t A t= f f cf0 Hzcf− 0 Hz USBLSB USBLSBLSBUSB cos2 cf tπ “real signal” Real signal f 0 Hz Complex conjugate USBLSB 1f1f− cos2 cf tπ 0 Hz cfcf− USBLSBLSBUSB IF cf f+c IFf f−IF cf f−c IFf f− − Double sideband (DSB) Double sideband (DSB) Department of Electronic Engineering, NTUT8/110
  9. 9. Frequency Conversion (II) • Recall Chapter 1 (linear modulation) • Yes, a modulated signal sm(t) is a real signal. ( ) ( ) ( ) ( )1 12 2 2 2 j t j tj f t j f tA t A t e e e e φ φπ π− − = + ( ) ( ) ( )( )1cos 2ms t A t f t tπ φ= + ( ) ( ) { }12 Re j t j f t A t e e φ π = ⋅ f 1f0 Hz1f− “complex”“complex” “real” Complex conjugate ( )I t 1cos tω 1sin tω− ( )Q t ( )ms t Real signal Complex envelope Department of Electronic Engineering, NTUT9/110
  10. 10. Frequency Conversion (III) 0 Hz 2f2f− 0 Hz 2f2f− USBLSBLSBUSB Real signal f 0 Hz Complex conjugate USBLSB 1f1f− 1 2f f+2 1f f−1 2f f−2 1f f− − 2cos2 f tπ RFIF ( )I t cos IFtω sin IFtω− ( )Q t ( )IFs t Modulated signal (real signal) f 0 Hz USBLSB IFfIFf− cos2 cf tπ RF 0 Hz cfcf− USBLSBLSBUSB IF cf f+c IFf f−IF cf f−c IFf f− − Double sideband (DSB) mixing upconversion upconversion IF LO LO By filtering, you can choose only USB or LSB transmission, which is call the single-sideband (SSB) transmission. Department of Electronic Engineering, NTUT10/110
  11. 11. Frequency Conversion (IV) 0 Hz 2f2f− 0 Hz 2f2f− Real signal f 0 Hz Complex conjugate 1f1f− 1 2f f+1 2f f−2 1f f−2 1f f− −2cos2 f tπ IFRF downconversion 2 1f f< 2f2f− 0 Hz 2f2f− 0 Hz 2f2f− Real signal f 0 Hz 1f1f− 1 2f f+2 1f f−1 2f f−2 1f f− −2cos2 f tπ IFRF downconversion 2 1f f> 2f2f− High-side injection Low-side injection LO LO Department of Electronic Engineering, NTUT11/110
  12. 12. Receiver Architecture • Basic Heterodyne Receiver • Modern Heterodyne Receiver Hetero-dyne Different-freq. Mixing Department of Electronic Engineering, NTUT12/110
  13. 13. Basic Heterodyne Receivers (I) • Translating the desired channel to a much lower center frequency to permit a channel-selection filtering with a reasonable Q. ω inωinω− 0 ω LOωLOω− 0 Downconversion by mixing RF input inω ω Mixer 0 cos LOA tω vLPF IF Output in LOω ω− − in LOω ω− + 0 in LOω ω+in LOω ω− Filtered-outFiltered-out LO ( ) ( ) 1 1 cos cos cos cos 2 2 in LO in LO in LOt t tω ω ω ω ω ω⋅ = + + − Low freq.High freq. Two IF frequencies: Department of Electronic Engineering, NTUT13/110
  14. 14. Basic Heterodyne Receivers (II) • Use of LNA to reduce noise • Variable IF: • Constant IF: Mixer 0 cos LOA tω vLPF IF Output RF input LNA IFj RFj LOf f f= − (Constant LO freq. and variable IF freq.) IF RFj LOjf f f= − (Variable LO freq. and constant IF freq.) Precise LO freq. and steps provided by a “frequency synthesizer” Constant IF approach is more common to simplify the design of IF path; e.g., it does not require a variable-frequency channel selection filter. LO Department of Electronic Engineering, NTUT14/110
  15. 15. Basic Heterodyne Receivers (III) • Constant-LO downconversion mixing • Constant-IF downconversion mixing 1RFf f f f 1LOf IFf0 1RFf f f f LOf 1IFf0 2RFf f f f 2IFf0 LOf 2RFf f f f IFf0 2LOf Department of Electronic Engineering, NTUT15/110
  16. 16. • While each wireless standard impose constrains upon the emissions by its own users, it may have no control over the signals in other bands. The image power can therefore be much higher than that of the desired signal, requiring proper “image rejection.” Image Problem in Heterodyne RX cos LOtω vLPF Desired signal Image inω imω ω IFω ω IFω IFω LOω ω High-side injection ( ) ( )cosd d dA t t tω φ +   ( ) ( )cosim im imA t t tω φ +   ( ) ( ) ( ) ( ) ( ) ( ) ( ) 1 1 cos cos 2 2 IF d LO d LO d d LO d LO dx t A t A t t A t A t tω ω φ ω ω φ   = + + − − +    ( ) ( ) ( ) ( ) ( ) ( ) 1 1 cos cos 2 2 im LO im LO im im LO im LO imA t A t t A t A t tω ω φ ω ω φ   + + + − − +    Department of Electronic Engineering, NTUT16/110
  17. 17. Downconverted Spectrum (I) 1ω−2ω− 1ω+ 2ω+0 ω ω ω 0 LOω+LOω− 0 1ω−2ω− 1ω+ 2ω+0 ω ω ω 0 LOω+LOω− 0 Downconversion for 1LOω ω< Downconversion for 2 1LOω ω ω> > Department of Electronic Engineering, NTUT17/110
  18. 18. Downconverted Spectrum (II) 1ω−2ω− 1ω+ 2ω+0 ω ω ω 0 LOω+LOω− 0 1ω−2ω− 1ω+ 2ω+0 ω ω ω 0 LOω+LOω− 0 Downconversion for 2 1LOω ω ω> > Downconversion for 2LOω ω> 1 2 2 LO ω ω ω + = Department of Electronic Engineering, NTUT18/110
  19. 19. • The most common “image rejection “ approach is to precede the mixer with an “image-rejection filter.” • The filter exhibits a relatively small loss in the desired band and a large attenuation in the image band, two requirements that can be simultaneously met if 2ωIF is sufficiently large. • A filter with high image rejection typically appears between the LNA and the mixer to lower the noise contribution to the RX NF (The NF increases while the filter precedes the LNA). Image Rejection Image Reject Filter inω imω ω 2 IFω cos LOtω v Image Reject Filter LNA Department of Electronic Engineering, NTUT19/110
  20. 20. Image Rejection v.s. Channel Selection Image Reject Filter inω imω ω 2 IFω cos LOtω v Image Reject Filter LNA v Channel Select Filter Desired channel Interference Image Channel Select Filter (high-Q needed) IFω ω 0 0 ω IFω inω imω 2 IFω ω High IF Low IF • If the IF is high, the image can be suppressed but complete channel selection is difficult, and vice versa. Department of Electronic Engineering, NTUT20/110
  21. 21. Image Noise Increases Noise Figure • Even in the absence of interferes, the thermal noise produced by the antenna and the LNA in the image band arrives at the input of the mixer. The thermal noises in the desired channel and image band are downconverted to IF (unless the LNA has a limited bandwidth to suppresses the noise in the image band). LOω LNA inω Thermal Noise LOω inω ω 2 in LOω ω− ω Department of Electronic Engineering, NTUT21/110
  22. 22. Dual IF Receiver (I) • The concept of heterodyning is extended to multiple downconversions, each followed by filtering and amplification, to resolve the trade-off between “image rejection” and “channel selection.” • This technique performs partial channel at progressively lower center frequencies, thereby relaxing the Q required of each filter. 1LOω vBPF2 LNA vBPF1 vBPF3 2LOω vBPF4 Band Select Filter Image Reject Filter RF Mixer MX1 Channel Select Filter IF Mixer MX2 Channel Select Filter IF Amp. A C E GB D F H Department of Electronic Engineering, NTUT22/110
  23. 23. Dual-IF Receiver (II) 1LOω vBPF2 LNA vBPF1 vBPF3 2LOω vBPF4 Band Select Filter Image Reject Filter RF Mixer MX1 Channel Select Filter IF Mixer MX2 Channel Select Filter IF Amp. A C E GB D F H Department of Electronic Engineering, NTUT Desired Channel Image fA C E G B D F H BPF1 BPF2 Image Image BPF3 BPF4 f f f f f f f 23/110
  24. 24. Mixing Spurs (I) • In practice, the mixing is the multiplication of the RF input by all harmonics of the LO. Thus the RF mixer produces components at and IF mixer, where m and n are integers. • For the desired signal, only is of interest. But if an interferer, , is downconverted to the same IF, it corrupts the signal; this occurs if . 1in LOmω ω± 1 2in LO LOm nω ω ω± ± 1 2in LO LOω ω ω− − intω int 1 2 1 2LO LO in LO LOm nω ω ω ω ω ω± ± = − − Department of Electronic Engineering, NTUT24/110
  25. 25. Mixing Spurs (II) • An example of a 2.4 GHz dual conversion RX : 1 1.98 GHzLOω = LNA vBPF 2 400 MHzLOω = 420 MHz2.4 GHz 20 MHz 2.76GHz 2.4 GHz 2.8GHz 4.38 GHz f Received Spectrum 820 MHz 780 MHz 2800 MHz 2760 MHz 22 800 MHzLOω = 12 3.96 GHzLOω = 4380 MHz 20 MHz 420 MHz 2 400 MHzLOω = 20 MHz Department of Electronic Engineering, NTUT25/110
  26. 26. Modern Heterodyne Receivers ω ω 0 0 1IFω− 1IFω+ 2LOω+2LOω− ω ω Desired Channel Interferer 0 0 • Zero Second IF: Avoid secondary image (assume no interferers are downconverted as an image to a zero center frequency). • Interferer appears in the adjacent channel 2LOω ω ω 1IFω ω 02 1LO IFω ω= 2 1LO IFω ω= Department of Electronic Engineering, NTUT26/110
  27. 27. Signal Becomes its Own Image • For symmetrically-modulated signal: • For asymmetrically-modulated signal: LOf f f 0 f LOf ( )S f ( )LOS f f− vVCO ( )BBx t t f cf 1IFω+1IFω− 0 ω ω 0 same information on both side of the carrier downconversion Corruption occurs if the signal spectrum is asymmetric Department of Electronic Engineering, NTUT27/110
  28. 28. Avoid Self-corruption of Asymmetric Signals (I) • One can downconvert signal to an IF equal to half of the signal bandwidth to avoid self-corruption of a signal with asymmetric spectrum. 1IFω+1IFω− 0 ω ω 0 BWω 2 BWω + 2 BWω − 1 2 BW IF ω ω ≥ Department of Electronic Engineering, NTUT28/110
  29. 29. Avoid Self-corruption of Asymmetric Signals (II) • Zero second IF with quadrature downconversion. ( )IFx t ( ),BB Ix t ( ),BB Qx t 2cos LO tω 2sin LO tω 2 1LO IFω ω= Quadrature baseband signal Though xBB,I(t) and xBB,Q(t) exhibit identical spectra, they are separated in phase and together can reconstruct the original information Department of Electronic Engineering, NTUT29/110
  30. 30. Zero Second IF Heterodyne RX • Zero second IF heterodyne RX with quadrature downconverison ( )IFx t ( ),BB Ix t ( ),BB Qx t 2cos LO tω 2sin LO tω 1LOω vBPF LNA No image rejection filter LNA/mixer interface can be optimized (need not 50 Ohms) for gain, noise, and linearity with little concern for the interface impedance values. The lack of image rejection filter requires careful attention to the interferers in the image band, and dictates a narrow-band LNA design (suppress image noise). No channel-selection filter is shown at the first IF, but some “mild” on-chip BPF is usually inserted to suppress out-of-band interferers. Department of Electronic Engineering, NTUT30/110
  31. 31. Sliding-IF Heterodyne RX (I) ( )IFx t ( ),BB Ix t ( ),BB Qx t 2,ILO 1LOω vBPF LNA vLO1 v2÷ 2,QLO t 2,ILO 2,QLO 1LO 90 RF Input 1st LO 1st IF 2nd IF inf f f f f 1LOf 1in LOf f− 1 1 2 LO in LO f f f− − For an input band [f1, f2], the LO must cover a range of [(2/3)f1, (2/3)f2]. 1 1 2 LO LO in f f f+ = 1 2 3 LO inf f= The 1st IF is not constant, because 1 1 3 IF in LO inf f f f= − = Department of Electronic Engineering, NTUT31/110
  32. 32. Sliding-IF Heterodyne RX (II) • As fin varies from f1 to f1, fIF1 goes from f1 /3 to f2 /3 (slide IF). RF Range LO Range 1st IF Range f 1f 2f f f 1 2 3 f 2 2 3 f 1 1 3 f 2 1 3 f ( ) 2 1 2 1 1 1 3 3% *100% 1 1 1 2 3 3 IF f f BW f f − =   +    ( ) ( ) 2 1 1 2 % *100% 1 2 RF f f BW f f − = + Department of Electronic Engineering, NTUT32/110
  33. 33. Sliding-IF Heterodyne RX (III) • Image band of the sliding-IF heterodyne RX Image Band RF Band LO Band 1 1 3 f 2 1 3 f 1f 2f f f 1 2 3 f 2 2 3 f Department of Electronic Engineering, NTUT33/110
  34. 34. Direct-Conversion Receivers • Direct-conversion receiver (DCR) is also called the “zero-IF”, or the “homodyne” receiver. • As mentioned previously, downconversion of an asymmetric- modulated signal to a zero IF leads to self-corruption unless the baseband signals are separated by their phases. I Q cos LOtω sin LOtωvBPF LNA vLPF vLPF inω LO inω ω= Department of Electronic Engineering, NTUT34/110
  35. 35. DCR Advantages • The absence of an image greatly simplifies the design process. • Channel selection is performed by low-pass filters, which can be realized on-chip as active circuit topologies with relatively sharp cut-off characteristics. • Mixing spurs are considerably reduced in number and hence simpler to handle. • The LNA/mixer interface can be optimized for gain, noise, and linearity without requiring a 50- impedance. Department of Electronic Engineering, NTUT35/110
  36. 36. DCR Issues − LO Leakage • A DCR emits a fraction of its LO power from its antenna, and the LO emission is undesirable because it may desensitize other receivers operating in the same band. Typical acceptable values range from −50 to −70 dBm (measured at the antenna). • In heterodyne receivers, since the LO frequency falls outside the band, it is suppressed by the front-end band-select filters in both the emitting receiver and the victim receiver. Pad LNA LO Substrate LO LNA LO leakage Cancellation of LO leakage by symmetry Department of Electronic Engineering, NTUT36/110
  37. 37. DCR Issues − DC Offset • The leakage causes “LO self-mixing” at the mixer to produce a dc component in the baseband (because multiplying a sinusoid by itself results in a dc term). The zero second IF architecture also suffers from this issue. • LO leakage yields a very large dc offset due to the high gain of the receiving chain, and this saturates the baseband circuits (degrades the dynamic range), prohibiting signal detection. • Time-varying dc offset RF LOV kV+ Pad LNA IF DCV V+ Department of Electronic Engineering, NTUT37/110
  38. 38. Effect of DC Offset in Baseband Chain • Since and , thus . Amplified by another 40 dB, this offset reaches 1-V at the baseband output. 1 31.6vA = ( )632 2 VleakV µ= 10 mVdcV = cos LOtω LPF sin LOtω LNA 1 30 dBvA = 2 40 dBvA = 0 cos inV tω ( )cosbb in LOV tω ω− 1 0bb vV A V= cosleak LOV tω 1dc v leakV A V= Department of Electronic Engineering, NTUT38/110
  39. 39. Leakage of Quadrature Phases of LO • The dc offset measured in the baseband I and Q outputs are often unequal. LO LNA ( )cosleak LO leakV tω φ+ ( ), cosdc I leak LO leak circuitV V Vα φ φ= + ( ), sindc Q leak LO leak circuitV V Vα φ φ= − + Department of Electronic Engineering, NTUT39/110
  40. 40. Cancelling DC Offset LNA cos LOtω LPF 1C 1R bv 1A Baseband SignalHPF Response 1f− 1f+0 f • Using a HPF (ac coupling) removes dc offset but also removes a fraction of the signal’s spectrum near zero frequency, thereby introducing distortion. • As a rule of thumb, the corner frequency of the HPF must be less than 1/1000 of the symbol rate for negligible distortion. This may require very large capacitance and thus difficult to integrate on chip (especially for low symbol rate). For the slow response to transient inputs (LO switch, LNA gain change), ac coupling is rarely used in today’s receivers. Department of Electronic Engineering, NTUT40/110
  41. 41. DCR Issues − Even-order Distortion • DCRs are additionally sensitive to even-order nonlinearity in the RF path (IM2 falls around DC to corrupt the desired signal and mixer feedthrough), and so are the heterodyne architectures having a second zero IF . Beat Component cos LOtω 1 2ω ω− Feedthrough LNA Desired Channel Interferers 0 ω 0 ω 1ω ω 2ω Department of Electronic Engineering, NTUT41/110
  42. 42. Mixer Feedthrough – Simple Mixer inV outV LO 1R ( ) ( ) ( ) ( ) ( ) ( ) 1 1 2 2 out in in inV t V t S t V t S t V t   = ⋅ = − + ⋅   where is the RF input and is the Ideal LO toggling between 0 and 1 with 50% duty cycle, and ( ) 1 2 S t   −   DC-free square wave ( ) 1 2 inV t ⋅ is the RF feedthrough to the output ( )inV t ( )S t represents a Department of Electronic Engineering, NTUT42/110
  43. 43. Mixer Feedthrough – Differential Mixer • If the output is sensed differentially, the RF feedthrough Vout1(t) and Vout2(t) are cancelled while the signal components add. • This cancellation is sensitive to asymmetrics, e.g., if the switches exhibits a mismatch between their on-resistance, then a net RF feedthrough arises in the differential output. ( ) ( ) ( )1out inV t V t S t= ⋅ LO LO 1R 1R 1outV 2outV inV ( )S t ( )1 S t− ( ) ( ) ( )2 1out inV t V t S t= ⋅ −   Department of Electronic Engineering, NTUT43/110
  44. 44. Even-order Distortion (I) • The 2nd order nonlinear effect cause the beat amplitude which grows with the square of the amplitude of the input tones. (log scale) (log scale) inA IIP2A IIP2A 1Aα 2 2 Aα ( ) ( ) ( )2 1 2out in inV t V t V tα α= + ( ) ( ) ( )2 2 1 1 2 2 1 2 2 1 2cos cos cos cosA t t A t A tα ω ω α ω ω α ω ω= + + + + + +⋯ ( ) 1 2cos cosinV t A t A tω ω= + Department of Electronic Engineering, NTUT • Since the net feedthrough of the beat depends on the mixer and LO asymmetries, the beat amplitude measured in the baseband depends on the derive dimensions and the layout and is therefore difficult to formulate. 44/110
  45. 45. Even-order Distortion (II) • Even-order distortion may manifest itself even in the absence of interferers. Suppose in addition to frequency and phase modulation, the received signal also exhibits amplitude modulation. • Both of the terms and are low-pass signals and, like the neat component, pass through the mixer with finite attenuation, corrupting the downconverted signal. ( ) ( ) ( )0 cosin cx t A a t t tω φ=  +   +     ( ) ( ) ( )2 1 2out in inV t V t V tα α= + ( ) ( ) ( ) ( )2 2 2 2 2 0 0 1 cos 2 2 2 2 c in t t x t A A a t a t ω φ α α +  +   = + +  ( )2 0A a tα ( )2 2 2a tα Department of Electronic Engineering, NTUT45/110
  46. 46. DCR Issues − Flicker Noise (I) • Linearity requirements limit the cascaded LNA/mixer gain, the downconverted signal in a DCR is still relatively small and hence susceptible to noise in the BB circuits. Since the signal is centered around zero frequency, it can be substantially corrupted by the flicker noise. • The mixers themselves may also generate flicker noise at their output. (logscale) f BWfCf 1000 BWf ( )1 fS f thS , 1 2 BW RF BWf f= 1 fS f α = th c S f α = th cS fα = ⋅ Assume noise components below fBW/1000 are unimportant ( )1 0.001 5.9 ln BW BW f c n BW c th c th BW thf BW f P df f f S f S f S f f α   = + − = + +    ∫ If no flicker noise 2n BW thP f S≈ Flicker noise penalty 1 2 n n P P Department of Electronic Engineering, NTUT46/110
  47. 47. DCR Issues − Flicker Noise (II) • In a good design, the thermal noise at the end of the baseband chain arises mostly from the noise of the antenna, the LNA, and the mixer. Thus, a higher front-end gain directly raises Sth, thereby lowering fc and hence the flicker noise penalty. • Flicker noise penalty: An 802.11g RX with fc of 200 kHz: 10 MHzBWf = 1 2 1.04n n P P = A GSM RX with fc of 200 kHz: 1 2 16.4n n P P = (logscale) BWf200100 ( )1 fS f ( )kHzf thS Downconverted GSM Channel Effect of flicker noise on a GSM channel Flicker noise makes it difficult to employ DCR for a narrow channel bandwidth. In such cases, the “low-IF” architecture proves a more viable choice. Department of Electronic Engineering, NTUT47/110
  48. 48. DCR Issues − I/Q Mismatch • DCR require 90o shift of the RF signal and this generally entails severe noise-power-gain trade-offs. I Q LOV vLPF vLPF 90 RFV I Q LOV vLPF vLPF RFV 90 Shift of RF signal or LO waveform by 90o Department of Electronic Engineering, NTUT48/110
  49. 49. I/Q Mismatch (I) • Errors in the 90o phase shift circuit and mismatches between the quadrature mixers result in imbalance in the amplitudes and phases of the baseband I and Q outputs. • The BB stages themselves may also contribute mismatches. I Q LOV vLPF vLPF RFV 90 Phase and Gain Error Phase and Gain Error Phase and Gain Error Phase and Gain Error Department of Electronic Engineering, NTUT49/110
  50. 50. I/Q Mismatch (II) • I/Q mismatches tend to be larger in DCRs than in heterodyne topologies, because: Propagation of a higher frequency experiences greater mismatches LO quadrature phases suffer from greater mismatches at higher frequencies cos LOtω sin LOtω LNA 5 GHz 10 ps 18 @5 GHzT∆ = ⇒ ( )IFx t ( ),BB Ix t ( ),BB Qx t LNA vLO1 4÷ 5 GHz 4 GHz 1 GHz 10 ps 3.6 @1 GHzT∆ = ⇒ DCR Heterodyne RX Department of Electronic Engineering, NTUT50/110
  51. 51. I/Q Mismatch – QPSK Example (I) ( ),BB Ix tvLPF vLPF ( )inx t 90 ( )LOx t 2 θ + 2 θ − 1 2 ε + 1 2 ε − ( ),BB Qx t ( ) cos sinin c cx t a t b tω ω= + , 1a b = ± ( ), 2 1 cos 2 2 LO I cx t t ε θ ω     = + +        ( ), 2 1 cos 2 2 LO Q cx t t ε θ ω     = − −        ( ), 1 cos 1 sin 2 2 2 2 BB Ix t a b ε θ ε θ    = + − +        ( ), 1 sin 1 cos 2 2 2 2 BB Qx t a b ε θ ε θ    = − − + −        The mismatch causes crosstalk between I and Q BB signals. Department of Electronic Engineering, NTUT51/110
  52. 52. I/Q Mismatch – QPSK Example (II) I Q t t I Q Ideal I Q t t I Q Ideal Only amplitude mismatch : 0, 0ε θ≠ = Only phase mismatch : 0, 0ε θ= ≠ ( ), 1 2 BB Ix t a ε  = +    ( ), 1 cos 2 2 BB Qx t b ε θ  = −    ( ), cos sin 2 2 BB Ix t a b θ θ = − ( ), sin cos 2 2 BB Qx t a b θ θ = − + Department of Electronic Engineering, NTUT52/110
  53. 53. Correction of I/Q Mismatch cos LOtω sin LOtω LPF LPF LNA I Q I Q Phase Mismatch Amplitude Mismatch t LPF LPF ADC ADC Logic φ φ cos LOtω sin LOtω LNA • Calibration of quadrature phase and gain either at power up or continuously is usually needed for high performance system. Test signal Analog adaption Digital adaption is more popular in nowadays systems Department of Electronic Engineering, NTUT53/110
  54. 54. Mixing Spurs • Unlike heterodyne systems, DCRs rarely encounter corruption by mixing spurs. This is because, for an input frequency f1 to fall in the baseband after experiencing mixing with nfLO, we must have f1 ≈ nfLO. • Since fLO is equal to the desired channel frequency, f1 lies far from the band of interest and is greatly suppressed by the selectivity of the antenna, the band-select filter, and the LNA. • The issue of LO harmonics does manifest itself if the receiver is designed for a wide frequency band (greater then two octaves). Examples include TV tuners, “software-defined radios,” and “cognitive radios.” Department of Electronic Engineering, NTUT54/110
  55. 55. Image-Reject Receivers • “Image-reject” architectures are another class of receivers that suppress the image without filtering, thereby avoiding the trade-off between image rejection and channel selection. • Benefits from a 90o shifter (Hilbert transform, −j for f > 0, +j for f <0) Re Im Re Im Re Im 2 A j+ 2 A 2 A 2 A j− cω− cω+ ω ( )cos 2 c cj t j t c A A t e eω ω ω − = + Illustration of 90o phase shift for a cosine ( ) ( )9090 cos 90 2 cc j tj t c A A t e e ωω ω − −− − = +    2 2 c cj t j tA A je jeω ω− = − + sin cA tω= Department of Electronic Engineering, NTUT55/110
  56. 56. Hilbert Transform (I) • Hilbert transformation pair: • The transform means a 90 degree phase shift in time domain, the impulse response of the Hilbert transformation . ( ) ( ) ( ) ˆ x x t d t τ τ π τ ∞ −∞ = −∫ ( ) ( ) ( ) ˆx t x d t τ τ π τ ∞ −∞ = − −∫ ( ) 1h t tπ= ( ) 1 h t tπ =( )x t ( )ˆx t 90( )x t ( )ˆx t Department of Electronic Engineering, NTUT56/110
  57. 57. Hilbert Transform (II) • Simple relation between sine and cosine functions: • It simply shows that if we want to make a transform between cosine and sine waveforms, a 90 degrees phase shift is required. • From Euler’s formula: ( ) ( )cos 90 sint tω θ ω θ+ − = + ( ) ( )sin 90 cost tω θ ω θ+ − = − +and 0 0 0cos 2 j t j t e e t ω ω ω − + = 0 0 0sin 2 j t j t e e t j ω ω ω − − = ( ) ( )0 0 1 2 δ ω ω δ ω ω− + +   ( ) ( )0 0 1 2 j δ ω ω δ ω ω− − +   F.T. F.T. Department of Electronic Engineering, NTUT57/110
  58. 58. Hilbert Transform (III) • We like to find a transfer function, which is able transfer the cosine to since and since to cosine function. ( ) ( ) ( ) ( ) ( )0 0 0 0 1 1 2 2 H j jδ ω ω δ ω ω ω δ ω ω δ ω ω− + + ⋅ = − − − +       ( ) ( ) ( ) ( ) ( )0 0 0 0 1 1 2 2 H j j δ ω ω δ ω ω ω δ ω ω δ ω ω   − − + ⋅ = − − + +    ( ) ( )sgnH j jω ω= − ⋅ ( ) 1 , 0 sgn 0 , 0 1, 0 ω ω ω ω >  = = − < ( ) ( ) 0 0 1 1 sgn 2 2 2 j t j t j tj j h t j e d e d e d t ω ω ω ω ω ω ω π π π π ∞ ∞ −∞ −∞ = − ⋅ ⋅ = − =∫ ∫ ∫ cosine sine negative cosine 0 0ω ω= > ( ) ( ) ( )0 1 1 0 0 2 2 H j jδ ω δ⋅ = − 0 0ω ω= − < ( ) ( ) ( )0 1 1 0 0 2 2 H j jδ ω δ⋅ − = 0 0ω ω= = ( ) ( ) ( ) ( ) ( ) 1 1 0 0 0 0 0 2 2 H jδ δ δ δ+ ⋅ = − −       ( )0 0H = ( )H j jω = − ( )H j jω = : : : sine phase f 90+ 90− Department of Electronic Engineering, NTUT58/110
  59. 59. 90o Phase Shift (I) • 90o phase shift for a modulated signal Re Im ( )X ω ω cω+ cω− ( )X ω( )jX ω ( )jX ω− ( ) ( ) ( )cos cx t A t t tω φ=  +   ( ) ( ) ( ) ( )( ) ( )( ) ( ) ( )( ) ( )( )90 90 cos 90 2 2 c c c c j t t j t t j t t j t t c A t A t A t t t e e je je ω φ ω φ ω φ ω φ ω φ + − − + − + − +    + − = + = − +       ( ) ( )sin cA t t tω φ=  +   ( ) ( ) ( )90 sgnX X jω ω ω= −   Department of Electronic Engineering, NTUT59/110
  60. 60. 90o Phase Shift (II) Re Im 2 A j+ 2 A 2 A j− cω− cω+ ω sin 2 c cj t j t c e e A t A j ω ω ω − − = 2 A − sin cjA tω Re Im 2 A 2 A cω− cω+ ω cos ctω Re Im A cω− cω+ ω cos sinc ct jA tω ω+ • Plot the spectrum of cos sinc cA t jA tω ω+ (SSB spectra) Department of Electronic Engineering, NTUT sine is the Hilbert transform of the cosine To get a SSB spectra: (1) real signal (2) its Hilbert transform (3) = (1)+j(2) 60/110
  61. 61. 90o Phase Shift (III) Re Im ( )S ω ω cω+ cω− ( )S ω( )jS ω ( )jS ω− Re Im ( )S ω ω cω+ cω− ( )S ω− ( )jS ω ( )jS ω− Re Im ( ) ( )ˆS jSω ω+ ω cω+ • A narrowband signal with a real spectrum is shifted by 90o to produce . Plot the spectrum of which is called the “analytic signal,” or the “pre-envelope” of . ( )s t ( )ˆs t ( ) ( )ˆs t js t+ ( )s t ( )ˆS ω ( )ˆjS ω Department of Electronic Engineering, NTUT61/110
  62. 62. 90o Phase Shift (IV) • Use RC-CR network to implement the 90o phase shifter 1C 1R 1R 1C 1outV 2outV inV HPFH LPFH 1 1 2 1 1 1 R C ω 1 1 1tan 2 HPFH R C π ω− ∠ = − 1 1 1tanLPFH R C ω− ∠ = − ω 2 π 2 π − 0 2 π ( ) 1 1 1 1 1 1 out HPF in V R C s H s V R C s = = + ( ) 2 1 1 1 1 out LPF in V H s V R C s = = + We can consider Vout2 as the Hilbert transform of Vout1 at frequencies close to (R1C1)−1 Department of Electronic Engineering, NTUT Amplitude response Phase response 62/110
  63. 63. Quadrature Downconversion (I) • Quadrature downconversion translate the spectrum to a nonzero IF as a 90o phase shifter. RFV IFI cos LOtω sin LOtω IFQ 0cω− cω+ ω 0LOω− LOω+ ω 1 2 + 1 2 + 1 2 + 1 2 + 0IFω− IFω+ ω 0cω− cω+ ω 2 j + 2 j − LOω+ 0LOω− ω 0IFω− IFω+ ω 2 j − 2 j + High-side injection Department of Electronic Engineering, NTUT63/110
  64. 64. Quadrature Downconversion (II) • Quadrature downconversion translate the spectrum to a nonzero IF as a 90o phase shifter. 2 j + 2 j − LOω+ 0LOω− ω 0 IFω− IFω+ ω 2 j − 2 j + 1 2 + 1 2 + 0IFω− IFω+ ω Low-side injection RFV IFI cos LOtω sin LOtω IFQ 0cω− cω+ ω 0cω− cω+ ω 0LOω− LOω+ ω 1 2 + 1 2 + Department of Electronic Engineering, NTUT64/110
  65. 65. Distinguish Desired Signal and Its Image ( )sig im IF I I+ cos LOtω sin LOtω ( )sig im IF Q Q+ Re Im sigI IFω− sigQ sigQ sigI IFω+ ω Re Im imI imQ imQ imI IFω+ ω Re Im cω+ ω imω+ imω− cω− Signal Components Image Components IFω− Department of Electronic Engineering, NTUT Signal: low-side injection Image: high-side injection LOω+ LOω− 65/110
  66. 66. Image Reject RX – Hartley Architecture (I) • Negate image Department of Electronic Engineering, NTUT C A LPF LPF B cos LOtω sin LOtω 90 sigQ imQ ,90im Q ,90sig Q sigI imI IF Output ω imω+ cω+cω− imω− 0 sigI sigQ imI imQ Re Im IFω− IFω+ ω Re Im IFω− IFω+ ω Re Im IFω− IFω+ ω Re Im IFω− IFω+ ω 66/110
  67. 67. Image Reject RX – Hartley Architecture (II) C A LPF LPF B cos LOtω sin LOtω 90 sigQ imQ ,90im Q ,90sig Q sigI imI IF Output Department of Electronic Engineering, NTUT sigI imI Re Im IFω− IFω+ ω Re Im IFω− IFω+ ω ,90sig Q ,90im Q Re Im IFω− IFω+ ω Re Im IFω− IFω+ ω Re Im IFω− IFω+ ω 67/110 • Negate image
  68. 68. Image Reject RX – Hartley Architecture (III) • Realization of 90o phase shift in Hartley receiver 1cos tω 1sin tω RF Input LPF LPF IF Output 1R 1R 1C 1C Department of Electronic Engineering, NTUT68/110
  69. 69. Image Reject RX – Hartley Architecture (IV) • Downconversion of Hartley receiver output to baseband LPF LPF 1cos LO tω 1sin LO tω 90 2sin LO tω 2cos LO tω I Q RF Input Department of Electronic Engineering, NTUT69/110
  70. 70. Image Reject RX – Weaver Architecture (I) A 1cos tω 1sin tω RF Input LPF LPF 2cos tω 2sin tω LPF LPF B C D E F − + IF Input Department of Electronic Engineering, NTUT ω imω+ cω+cω− imω− 0 sigI imI Re Im 1IFω− 1IFω+ ω Re Im 1IFω− 1IFω+ ω sigQ imQ Re Im 1IFω− 1IFω+ ω Re Im 1IFω− 1IFω+ ω 70/110 • Negate image
  71. 71. Image Reject RX – Weaver Architecture (II) A 1cos tω 1sin tω LPF LPF 2cos tω 2sin tω LPF LPF B C D E F − + IF Department of Electronic Engineering, NTUT sigI imI Re Im 2IFω− 2IFω+ ω Re Im 2IFω− 2IFω+ ω sigQ imQ Re Im 2IFω− 2IFω+ ω Re Im 2IFω− 2IFω+ ω Re Im IFω− IFω+ ω Low-side inj. 71/110 • Negate image
  72. 72. Image Reject RX – Weaver Architecture (III) 1inω ω− ω 2ω 2 12 inω ω ω− + 0 First IF ω 1 2inω ω ω− − 0 Second IF ω 2 12 2inω ω ω− + 1ω inω0 RF Input Secondary Image Desired Channel Department of Electronic Engineering, NTUT • Problem of secondary image 72/110
  73. 73. Image Reject RX – Weaver Architecture (IV) Department of Electronic Engineering, NTUT 1cos tω 1sin tω RF Input LPF LPF 2÷ ( ),BB Qx t − +− + ( ),BB Ix t • Double quadrature downconversion Weaver architecture to produce BB outputs. The second downconverion produces zero IF to avoid secondary image. 73/110
  74. 74. Low-IF Receiver (I) cf GSM Adjacent Channel Spec. 9 dB ff 0 100 kHz cf f LOf f 200 kHz • It is undesired to place image within the signal band because the overall NF would raise by approximately 3 dB. • In “low-IF” RXs, the image indeed falls in the band but can be suppressed by image rejection techniques. • For a GSM RX, signal would be corrupted by flicker noise in a zero-IF architecture. The noise penalty can be lower by using low-IF architecture (attractive for narrow-channel standards). Moderate IRR is ok. Department of Electronic Engineering, NTUT74/110
  75. 75. Low-IF Receiver (II) 2sin tω LPF 2cos tω LPF 1R 1R 1C 1C IF OutputRF Input Quadrature Phases of Image and Signal sin LOtω LPF cos LOtω IF OutputRF Input LPF ADC ADC 90 Department of Electronic Engineering, NTUT • Adopt image cancellation technique with low-IF architecture 75/110
  76. 76. Low-IF Receiver (III) • Low-IF receiver with double quadrature downconverter Department of Electronic Engineering, NTUT sin ctω cos ctω IF Output RF Input ( ),IF Qx t − + + + ( ),IF Ix t90 76/110
  77. 77. Polar Receiver • Using the oscillator injection locking technique to accomplish magnitude and phase extraction of the complex envelope. This technique was also published to perform envelope elimination and restoration in the Kahn EER transmitter. LPF LPF ILO1 ILO2 Magnitude Phase Department of Electronic Engineering, NTUT77/110
  78. 78. Transmitter Architectures • Basic Direct Conversion Transmitter (DCT) • Modern DCT • Heterodyne Transmitters • OOK Transceivers • Open-loop Phase Modulation Techniques • Closed-loop Phase Modulation Techniques • Polar Transmitter Department of Electronic Engineering, NTUT78/110
  79. 79. • Quadrature upconverter: Server as the modulator. • Power amplifier: Amplify the signal. • Matching network : Provide maximum power delivery to antenna and filter out-of-band components that result from the PA nonlinearity. • xBB,I(t) and xBB,Q(t) are generated by BB circuits and hence has a sufficiently large amplitude, the noise of the mixers is much less critical here than in receivers. • A predriver is typically interposed between the upconverter and the PA to serve as a buffer. Direct-Conversion Transmitter (DCT) cos ctω Matching Network PA sin ctω− ( ) ( ) ( )cos cx t A t t tω φ=  +   ( ) ( ) ( ) ( )cos cos sin sinc cA t t t A t t tφ ω φ ω= − ( ) ( ) ( ), cosBB Ix t A t tφ= ( ) ( ) ( ), sinBB Qx t A t tφ= Department of Electronic Engineering, NTUT79/110
  80. 80. I/Q Mismatch • I/Q mismatch in the DCT: • Constellation: ( ) ( ) ( )1 2cos sinc c c c cx t A A t A tα ω θ α ω= + ∆ + ∆ + ( ) ( )1 2 1cos cos sin sinc c c c c c cA A t A A A tα θ ω α α θ ω = + ∆ ∆ + − + ∆ ∆  1 2, 1α α = ± 1 21 cos , 1 1 sinc c c c A A A A β θ β θ    ∆ ∆ = + + ∆ = − + ∆        1 21 cos , 1 1 sinc c c c A A A A β θ β θ    ∆ ∆ = + + ∆ = − − + ∆        1 21 cos , 1 1 sinc c c c A A A A β θ β θ    ∆ ∆ = − + ∆ = + + ∆        1 21 cos , 1 1 sinc c c c A A A A β θ β θ    ∆ ∆ = − + ∆ = − + + ∆        Department of Electronic Engineering, NTUT80/110
  81. 81. I/Q Calibration (I) • Apply a single sinusoidal to both inputs of the upconverter to reveal phase mismatch. It can be shown that the output contains two sidebands of equal amplitudes and carries an average power equal to We observe that ε is forced to zero as described above, then . Thus, the calibration of phase mismatch proceeds to drive this quantity to zero. cos ctω sin ctω0 cos inV tω 3outV + − ( ) ( ) ( )3 0 01 cos cos cos sinout in c in cV t V t V tε ω ω θ ω ω= + + ∆ − ( )0 cos 1 cos cosin cV tω ε θ ω= + ∆ ( ) ( )2 2 3 0 1 1 sinoutV t V ε θ=  + + ∆   2 2 3 1 sinout outV V θ− = ∆ ( )0 cos 1 sin 1 sinin cV t tω ε θ ω− ⋅  + ∆ +   Department of Electronic Engineering, NTUT81/110
  82. 82. I/Q Calibration (II) • Applying a sinusoidal to one BB input while the other is set to zero for gain mismatch calibration. The gain mismatch can be adjusted so as to drive this difference to zero. 0 cos inV tω cos ctω sin ctω 1outV cos ctω sin ctω 0 cos inV tω 2outV ( ) ( ) ( )1 0 1 cos cosout in cV t V t tε ω ω θ= + ⋅ + ∆ ( ) 2 2 20 1 0 2 out V V t V ε= + ( )2 0 cos sinout in cV t V t tω ω= ⋅ ( ) 2 2 0 2 2 out V V t = ( ) ( )2 2 2 1 2 0out outV t V t V ε− = Department of Electronic Engineering, NTUT82/110
  83. 83. Carrier Leakage (I) • The analog BB circuitry producing the quadrature signals exhibits dc offsets, and so does the baseband port of each upconvertion mixer: • The upconverter therefore contains a fraction of the unmodulated carrier, called “carrier leakage”: ( ) ( ) ( )1 2cos cos sin sinout OS c OS cV t A t V t A t V tφ ω φ ω=  +  −  +     ( ) ( ) ( ) 1 2cos cos sinout c OS c OS cV t A t t V t V tω φ ω ω= + + − ( ) 2 2 1 2 2 Relative Carrier Leakage OS OSV V A t + = Department of Electronic Engineering, NTUT83/110
  84. 84. Carrier Leakage (II) • RX BB outputs suffer from dc offsets. • In the presence of carrier leakage, if the TX power is controlled by varying BB signals, it is difficult forThe base station to measure the actual signal power. 0 2OSV V+ + Q I 0 1OSV V+ +0V− 0V+ 0V− 0V+ cos ctω sin ctω PA Receiver Base Station Baseband Processor Carrier Leakage cω ω Department of Electronic Engineering, NTUT84/110
  85. 85. Reduction of Carrier Leakage Baseband Processor DACQ DACI Register ADC Power Detector cos ctω sin ctω • The BB swing, A(t), must be chosen sufficiently larger to reduce carrier leakage. However, as A(t) increases, the input port of mixers becomes more nonlinear. A compromise is therefore necessary. ( ) 2 2 1 2 2 Relative Carrier Leakage OS OSV V A t + = • Use BB offset control to reduce the carrier leakage. During carrier leakage cancellation, the BB processor produces a zero output so that the detector measures only the leakage. Thus, the loop can use the DACs to drive the leakage toward zero. Department of Electronic Engineering, NTUT85/110
  86. 86. Transmitter Linearity (I) • Upconversion mixers in TX sense no interferers, however, excessive nonlinearity in the mixer BB port can corrupt the signal or raise the adjacent channel power. • In most cases, as the BB signal swings increase, the PA output begins to compress before the mixer nonlinearity manifests itself. • Power back-off is required for variable envelope signal to avoid spectrum regrowth at PA output. 1-dB Compression Point outV inV 0V t Department of Electronic Engineering, NTUT86/110
  87. 87. Transmitter Linearity (II) • In the TX chain, the signal may experience compression in any of the stages. Since the largest voltage swing occurs at the output of the PA, this stage dominates the compression of the TX. In a good design, the preceding stages must remain well below compression as the PA output approaches P1dB. To ensure this, we must maximize the PA gain and minimize the output swing of the predriver and the stage preceding it. cos ctω sin ctω ,BB IV ,BB QV PAPredriver XV drV outV XV drV outV BBV Department of Electronic Engineering, NTUT87/110
  88. 88. Oscillator Pulling • The PA output (very large swing) would couple to various parts of the system through the silicon substrate, package parasitics, and traces on the printed-circuit board. Thus, it is likely that an appreciable fraction of PA output couples to the LO to pull the oscillator. outφLO ω∆ LOω ω LOω ω ω∆ PA LO I LOω Q Output Spectrum injω ω Department of Electronic Engineering, NTUT88/110
  89. 89. Avoid LO Pulling (I) • Most of today’s DCTs avoid an oscillator frequency to the PA output frequency by using frequency division and mixing. • Since the PA nonlinearity produces a finite amount of power at the second harmonics of the carrier, the LO may still be pulled by using the following architecture. • Very high speed divider is needed, but even a substantial effort on divider design to enable this architecture is well justified. I 2LO cω ω= Q LO 2÷ PA cω 2 cω ω Department of Electronic Engineering, NTUT89/110
  90. 90. Avoid LO Pulling (II) • Use a frequency double is possible to avoid LO pulling, but the doubler typically dose not provide quadrature phases, necessitating additional quadrature generation stages such as the poly phase filter. • Advantage: no harmonic can pull the LO. • Disadvantage: the doubler and polyphase filter suffer from a high loss, requiring the use of power-hungry buffers. I 2 c LO ω ω = Q LO 2X Polyphase Filter PA cω ω Department of Electronic Engineering, NTUT90/110
  91. 91. Avoid LO Pulling (III) • Use of mixing to avoid LO pulling. • It is difficult to remove the unwanted carrier by means of filtering because the two frequencies are only differ by only a factor of 3. Even the filter is applied, the unwanted sideband would corrupt other channels or bands. 1 2 ω LO 2÷ 1ω 1 2 ω 13 2 ω ω PA Quadrature Upconverter ( ),BB Ix t ( ),BB Qx t chI chQ LO 1 2 ω 13 2 ω ω 1 2 ω 13 2 ω ω Department of Electronic Engineering, NTUT91/110
  92. 92. Suppress Unwanted Sideband (I) • Use the single-sideband (SSB) mixing technique to suppress the unwanted sideband instead of filtering. • The harmonics of the input frequencies also corrupt the output of an SSB mixer. 2cos tω 2sin tω 1cosA tω 1sinA tω outV 1ω 2ω outV ( )1 2 1 2 1 2cos cos sin sin cost t t t tω ω ω ω ω ω− = + Symbol of a SSB mixer 2ω 1ω1 2ω ω−0 2 13ω ω− 1 2ω ω+ 1 23ω ω− ω Department of Electronic Engineering, NTUT92/110
  93. 93. Suppress Unwanted Sideband (II) • For use in a DCT, the SSB mixer must provide the quadrature phases of the carrier. This is accomplished by nothing that 2cos tω 2sin tω 1cos tω 1sin tω 2cos tω 2sin tω + − + + ( )1 2sin tω ω+ ( )1 2cos tω ω+ ( )1 2 1 2 1 2sin cos cos sin sint t t t tω ω ω ω ω ω− = + ( )1 2 1 2 1 2cos cos sin sin cost t t t tω ω ω ω ω ω− = + SSB mixer providing quadrature outputs Department of Electronic Engineering, NTUT93/110
  94. 94. Suppress Unwanted Sideband (III) • Carrier provided by SSB mixing for a DCT. • While suppressing the carrier sideband at ω1/2, this architecture presents two drawbacks: (1) the spurs at 5ω1/2 and other harmonic-related frequencies prove troublesome, and (2) the LO must provide quadrature phases, a difficult issue. ( ),BB Ix t ( ),BB Qx t 12 3 ω 1 3 ωI/QI/Q LO 2÷ PA 1ω DCT using SSB mixing in LO path Department of Electronic Engineering, NTUT94/110
  95. 95. Heterodyne Transmitter • Another approach to avoiding injection pulling involves performing the signal upconversion in two steps so that the LO frequency remains far from the PA spectrum. • Smaller I/Q mismatch 1sin tω 1cos tω 2cos tω I BPF PA Q 1ω ω 2ω ω 1 2ω ω+ ω Department of Electronic Engineering, NTUT95/110
  96. 96. Sliding-IF TX • The carrier frequency is equal to 3ω1/2. 1ω PA BPF LO2÷ I Q 1 1 2 ω 1 3 2 ω RF Mixer Department of Electronic Engineering, NTUT96/110
  97. 97. Carrier Leakage • The dc offsets in BB yield a component at ω1/2 at the output of the quadrature upconverter, and the dc offset at the input of the RF mixer produces another component at ω1. The former can be minimized, and the latter (lower sideband) at ω1/2 must be removed by filtering. The leakage at ω1 is closer to the upper sideband than the lower sideband is, but it is also much smaller than the lower sideband. Thus, the filter following the RF mixer must be designed to attenuate both to acceptably low levels. 1 2 ω IF Output 1 2 ω 13 2 ω 1ω ω RF Output Carrier leakage in heterodyne TX Department of Electronic Engineering, NTUT97/110
  98. 98. Mixing Spurs (I) • Heterodyne TX displays various mixing spurs that must be managed properly. The spurs arise from the mechanism with 1st LO and 2nd LO. 1 2 ω + IF Output 13 2 ω + 15 2 ω +015 2 ω − 13 2 ω − 1 2 ω − ω 1 2 ω + 13 2 ω + 15 2 ω +0 17 2 ω +13 2 ω − 1 2 ω − ω RF Output 1 2 ω + 13 2 ω +17 2 ω − 015 2 ω − 13 2 ω − 1 2 ω − ω LO mixed with 2LO, 5LO IF mixed with 2nd LO Department of Electronic Engineering, NTUT98/110
  99. 99. Mixing Spurs (II) • Effect of harmonics of 2nd LO on TX output. Upon mixing with +3ω1, the IF sideband at −3ω1/2 is translated to +3ω1/2, thereby corrupting the wanted sideband (if the modulation is asymmetric). Similarly, the IF sideband at −5ω1/2 is mixed with +5ω1 and falls atop the desired signal. 13 2 ω + ω 13 2 ω − 015 2 ω − 13 2 ω + ω 0 Department of Electronic Engineering, NTUT99/110
  100. 100. Reduce Unwanted Components • Use of BB quadrature SSB mixing and IF SSB mixing to reduce the unwanted component. ( ) ( ), cosBB Ix t A t θ= ( ) ( ), sinBB Qx t A t θ= + − + + 1ω LO2÷ PA RF SSB Mixer I Q 13 2 ω Department of Electronic Engineering, NTUT100/110
  101. 101. OOK Transceivers • On-off keying (OOK) modulation is a special case of ASK where the carrier amplitude is switched between zero and maximum. • Less bandwidth-efficient as unshaped binary pulses modulated on one phase of the carrier occupy a wide spectrum. LO PA LO PA LNA Envelope Detector Direct LO switching PA switching OOK RX OOK TX Department of Electronic Engineering, NTUT101/110
  102. 102. Open-loop Modulation • Open-loop modulation based-on a frequency synthesizer (or phase-locked loop). • Wideband (high data rate). • Poor accuracy due to VCO frequency drifting. reff DAC VCO PFD Loop Filter Div-by-N [ ]BBs n ( )BBs t ( )ms t Department of Electronic Engineering, NTUT102/110
  103. 103. Closed-loop Modulation (I) • Closed-loop modulation based-on a frequency synthesizer (or phase-locked loop). • Narrowband (low data rate). • Good frequency accuracy. • No DACs required. ∆ − ∑ VCO PFD Loop Filter [ ]BBs n / 1N N÷ + ( )ms t reff Modulator Department of Electronic Engineering, NTUT103/110
  104. 104. Closed-loop Modulation (II) • Use the compensated filtering to increase the data rate. ∆ − ∑ VCO PFD Loop Filter [ ]BBs n / 1N N÷ + ( )ms t reff Modulator Compensated Filter Department of Electronic Engineering, NTUT104/110
  105. 105. Closed-loop Modulation (III) • Use the two-point ∆-Σ modulation to increase the data rate. ∆ − ∑ Two-point VCO PFD Loop Filter [ ]BBs n / 1N N÷ + ( )ms t reff Modulator DAC Department of Electronic Engineering, NTUT105/110
  106. 106. Envelope Detector Envelope Following/Tracking Transmitter • Dynamically adjusting bias to improve efficiency. ( )BBA t′ ( )ms t Linear PA Antenna Matching ( )BBA t I/Q Modulator Amplitude Modulator/ Regulator Department of Electronic Engineering, NTUT ( )I t cos ctω sin ctω− ( )Q t 106/110
  107. 107. Polar Transmitter (I) • Envelope Elimination and Restoration Scheme (Kahn EER TX, 1952): Department of Electronic Engineering, NTUT Envelope Detector ( )BBA t′ ( )ms t Switching-mode PA Antenna Matching ( )BBA t I/Q Modulator Amplitude Modulator/ Regulator ( )I t cos ctω sin ctω− ( )Q t Limiter 107/110
  108. 108. Polar Transmitter (II) • Polar Transmitter Department of Electronic Engineering, NTUT ( )BBA t cos ctω ( )ms t Switching-mode PA Antenna Phase Modulator Matching ( )BBA t ( )BB tφ Baseband Processor Amplitude Modulator ( ) { }2 Re c BBj f t t e π φ+   • Linear modulator to generate PM signal • Frequency synthesizer or PLL-based PM modulator 108/110
  109. 109. Polar Transmitter (III) • Hybrid Quadrature and Polar Modulation TX (HQPM-TX): Department of Electronic Engineering, NTUT Baseband Processor ( )BBA t′ ( )ms t Switching-mode PA Antenna Matching ( ),BB DSMA t I/Q Modulator Amplitude Modulator/ Class-S ( )I t cos ctω sin ctω− ( )Q t 109/110
  110. 110. Summary • In this chapter, many receiver and transmitter architectures were introduced. For receiving or transmitting, there are two main categories including heterodyne and direct conversion architectures. • For these transceivers, the modulation and demodulation can be classified as “I/Q” and “polar” schemes. I/Q modulator is an universal modulator with high linearity and signal quality, and the polar modulator is adopted for improving power efficiency. I/Q demodulator is the conventional scheme to demodulate signals, and the polar demodulator is proposed for low-cost and low-power applications. Department of Electronic Engineering, NTUT110/110

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