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RF Module Design - [Chapter 8] Phase-Locked Loops

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Phase-Locked Loops

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RF Module Design - [Chapter 8] Phase-Locked Loops

  1. 1. RF Transceiver Module Design Chapter 8 Frequency Synthesis and Phase-Locked Loops 李健榮 助理教授 Department of Electronic Engineering National Taipei University of Technology
  2. 2. Outline • Frequency Synthesis Techniques • Frequency Synthesizers based on the Phase-Locked-Loop • Loop Analysis and Stability • Settling Time • Loop Filters • Noise Analysis • PLL Architectures • Summary 2/55 Department of Electronic Engineering, NTUT
  3. 3. Generic Transceiver Front End • Local oscillator (LO) provides the carrier signal for both the receive and transmit paths. • If the LO output contains phase noise, both downconverted and upconverted signals are corrupted. BPF LNA Duplexer Antenna v Frequency Synthesizer LO PA BPF Department of Electronic Engineering, NTUT3/55
  4. 4. Effect of Phase Noise in Receivers f 0f Wanted Signal LO Output Wanted Signal Downconverted Signal f Downconverted Signals ff 0f Wanted Signal LO Output Interferer • Reciprocal mixing Department of Electronic Engineering, NTUT4/55
  5. 5. Effect of Phase Noise in Transmitters f 1f Wanted Signal Nearby Transmitter 2f f 0f Multi-carrier signal (or OFDM) f 0f • Receiver desensitization • Orthogonality Department of Electronic Engineering, NTUT5/55
  6. 6. • Meaning of frequency • Meaning of frequency synthesis Generation of a frequency or frequencies that are exact multiples of a reference frequency. Usually the reference is very precise and the synthesized frequencies are selectable over some range of whole-number multiples of a submultiple of . Frequency Synthesis out ref n f f M = where n and M are integers, n varies from Nmin to Nmax, and M is constant. ( )v t t 1T 1 1 f ( )V f f 1f reff Department of Electronic Engineering, NTUT6/55
  7. 7. Transformation To and From Voltage or Current A B1f Frequency Discriminator ∫ C D d dt Voltage Controlled Oscillator (VCO) Phase Detector Phase Modulator 1 1v Af= 2 1f Bv= 1 1f dtφ = ∫ 2 1v Cφ= 2 2Dvφ = 3 2 d f dt φ= V/Hz Hz/V V/rad or V/cycle rad/V or cycle/V Department of Electronic Engineering, NTUT7/55
  8. 8. Demonstration of the Transfer Functions rms rms rms rms rms 1 rad 2 V rad rad 1 1 V 2 V 2 V V 0.32 V V cycle cycle 2 rad AV π π   = ⋅ = = = =    RMS voltage at point A: 1 rms MHz 0.32 V 1.5 0.48 MHz V f∆ = ⋅ = rms 2 0.32 V MHz 1 0.32 MHz 1 k mA f∆ = ⋅ = Ω ( ) rms 5 V 0.48 MHz-0.32 MHz 0.8 V MHz DV = ⋅ = Phase Modulator 1 rad/V Phase Detector 2 V/Cycle 220 MHz VCO 1.5 MHz/V 1 kΩ 200 MHz ICO 1 MHz/mA 100 MHz signal Low-pass Filter 50 MHz Cut-off Frequency Discriminator 5 V/MHz A B D Modulation voltage (1 Vrms at 10 kHz) C and Department of Electronic Engineering, NTUT8/55
  9. 9. • Frequency addition and subtraction: Mathematical Operations on Frequency ( ) ( )cos 2RF RF RFv t A f tπ θ= + ( ) ( ) ( ) ( ) ( )cos 2 cos 2IF RF LO RF RF LO LOv t v t v t A f t B f tπ θ π θ= ⋅ = + × + For the practical mixer with nonlinear operation: IF RF LOf mf nf= + ( ) ( )cos 2LO LO LOv t B f tπ θ= + ( ) ( ) ( ) ( ){ }cos 2 cos 2 2 RF LO RF LO RF LO RF LO AB f f t f f tπ θ θ π θ θ= + + + + − + −       ( ) ( ){ } ( )cos2 cos2 for 0 2 RF LO RF LO RF LO AB f f t f f tπ π θ θ= + + − = = ( ) ( ), cosIF m nv t K m nα β= + 2 1f Bv= 2 LO LOf tβ π θ= +where and or we can say the intermediate frequency is: RF LO IF Mixer Department of Electronic Engineering, NTUT9/55
  10. 10. Frequency Synthesis Techniques • Direct Analog Synthesis (DAS) • Direct Digital Synthesis (DDS) • Indirect Synthesis : Phase-Locked Loops (PLLs) • Hybrid DDS/PLL Department of Electronic Engineering, NTUT10/55
  11. 11. • Frequency generated by mixed frequencies Direct Analog Synthesis (DAS) (I) 1f 2f 3f 2Nf − 1Nf − Nf outf filter1 filter2 filter3 filterN-2 filterN-1 filterN out a bf mf nf= ± Department of Electronic Engineering, NTUT11/55
  12. 12. • More stages are required for flexibly frequency planning. Direct Analog Synthesis (DAS) (II) 1f 2f 3f 2Nf − 1Nf − Nf filter1 filter2 filter3 filterN-2 filterN-1 filterN filter1 filter2 filter3 filterN-2 filterN-1 filterN outf 1f ′ 2f ′ 3f ′ 2Nf − ′ 1Nf − ′ Nf ′ ( )out a b cf p mf nf qf= ± ± Department of Electronic Engineering, NTUT12/55
  13. 13. Direct Digital Synthesis (DDS) (I) ref cf f= • Waveform construction is based on the lookup table (LUT) and a digital to analog converter (DAC). • Direct synthesis • Generated frequency is lower than input frequency Ref. Clk Phase Accumulator Amplitude/Sin Conv. Algorithm DAC Tuning word N Digital Analog 2 o cN M f f= ⋅ M Jump size 0000…0 1111…1 1111…0 Department of Electronic Engineering, NTUT13/55
  14. 14. Direct Digital Synthesis (DDS) (II) • Hardware technique to reduce the spur level of a DDS • Reduce bandwidth 1000 MHz 100−150 MHz 1100−1150 MHz 110−115 MHz div-by-10 DDS Filter Frequency Divider 0f outf BW=50 MHz BW=15 MHz reff Department of Electronic Engineering, NTUT14/55
  15. 15. Hybrid DDS/DAS 1f ′ 2f ′ 3f ′ 2Nf − ′ 1Nf − ′ Nf ′ outf filter1 filter2 filter3 filterN-2 filterN-1 filterN DDS Frequency Divider reff • Scheme to increase a DDS output bandwidth Department of Electronic Engineering, NTUT15/55
  16. 16. • The main goal of the PLL is to sync the divided oscillator frequency with the reference frequency . Indirect Frequency Synthesis ( )outf N reff ( )out reff N f= out reff N f= ⋅ PFD: Phase frequency detector LPF: Loop filter VCO: Voltage controlled oscillator /N: Divided-by-N frequency divider Frequency divider reff outf /N PFD LPF VCO Department of Electronic Engineering, NTUT16/55
  17. 17. Fractional-N Frequency Synthesis • Lower division ratio N to reduce inband phase-noise gain • Effectively produce a fractional division value • Generally employee a delta-sigma modulator for division ratio dithering PFD LPF Dual-modulus Frequency Divider reff outf /N, (N+1) FCW Department of Electronic Engineering, NTUT17/55
  18. 18. Phase-Locked Loop Analysis Department of Electronic Engineering, NTUT18/55
  19. 19. Feedback System ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( )o i o i oV s V s V s H s G s G s V s V s G s H s= − = −   ( ) ( ) ( ) ( ) ( ) ( )1 o i V s G s T s V s G s H s = = + Closed-loop transfer function T(s) G(s)H(s) is the open-loop transfer function ( )iV s ( )oV s( )G s ( )H s error • Feedback loop: Department of Electronic Engineering, NTUT19/55
  20. 20. Loop Analysis – Use Frequency as I/O ( ) ( )p v K G s F s K s = ( ) 1 H s N = ( ) ( ) ( ) ( ) ( ) ( ) ( ) 11 1 p v out pref v K F s Kf s G s s Kf s G s H s F s K N s = = + + • Relation between input and output frequencies: Phase differenceFrequency difference Frequency Divider ( )reff s ( )outf s 1 s pK vK 1 N ( )outf s N PFD LPF VCO ( )F s Department of Electronic Engineering, NTUT20/55
  21. 21. • Relation between input and output phases: Loop Analysis – Use Phase as I/O ( ) ( ) v p K G s K F s s = ( ) 1 H s N = ( ) ( ) ( ) ( ) ( ) ( ) ( ) 11 1 v p out vref p K K F ss G s s Ks G s H s K F s N s φ φ = = + + Frequency Divider ( )ref sφ ( )out sφpK ( )F s vK s 1 N Phase difference Frequency to Phase PFD LPF VCO Department of Electronic Engineering, NTUT21/55
  22. 22. Loop Transfer Functions ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( ) 0 0 0 01 1 p po i p p K F s K s K F s K G s T s K F s K Ns Ns K F s K G s H s φ φ = = = = + + + T(s) : closed-loop PLL transfer function G(s) : forward-path transfer function F(s) : loop filter transfer function Kp: phase detector gain K0/s: VCO transfer function ( ) ( )1 0G s H s+ = ( ) ( ) 1 0 dB@ 180G s H s = − = ∠ − H(s) : feedback-path transfer function G(s)H(s) : open-loop transfer function or • A PLL is unstable when The condition of unity open loop gain and a phase angle of 180 degrees must be avoided. Department of Electronic Engineering, NTUT22/55
  23. 23. PLL Response w/o a Loop Filter (I) ( ) 0 0 0 01 F p LPFo Fi p LPF K K K K s NT s N N KK K K Ns ss N φ ω φ ω = = = = + ++ ( ) LPFF s K= 0ω logω dB 0 FK N ω ω= = 20log N 3 dB 0ω • Closed-loop transfer function: Without the loop filter, the feedback loop is equivalent to a DC gain of N plus a low-pass filter (in dB) with cutoff at . 3 dB cutoff frequency is KF/N = KpKLPFK0/N . With Department of Electronic Engineering, NTUT23/55
  24. 24. • The simplest PLL is called a type-I loop because the open-loop gain has one pole at DC (pure integration). It is also a first- order loop because the open-loop gain has one significant pole: • The open loop gain has a slope of −6 dB/octave or −20 dB/decade for all frequencies. • The phase angle is always −90 degrees at all frequencies. Hence with no low-pass filter, the PLL is always stable. But the main drawback is that designers loose control over the loop. PLL Response w/o a Loop Filter (II) ( ) ( ) ( )p vK F s K G s H s Ns = Department of Electronic Engineering, NTUT24/55
  25. 25. • The function of the LPF is to filter out any high frequency harmonics in the loop. • Adding a LPF also affects the loop response including parameters such as the loop time response, bandwidth, and the damping factor. • If we add a low-pass filter with a pole located at , the loop will be still type-I, but it will become a second-order loop. Single Pole Loop Filter ( ) ( )1 LPF p K F s s ω = + ( ) ( ) 1 1 p LPF v F p p K K K K N NG s H s s s s s ω ω = =     + +           pω Department of Electronic Engineering, NTUT25/55
  26. 26. Bode Plot of Open-Loop Transfer Function ( ) ( ) ( ) 1 G s G s H s N = = ( )G s N= (forward-loop gain ) log mω ( ) ( )m mG Hω ω 6 dB/oct− 1020log FK N 0 dB Lω pω1mω = 90− 135− 180− ( )mG ω∠ 12 dB/oct− Phase margin • Where the curves cross, the open-loop gain equals unity. Department of Electronic Engineering, NTUT26/55
  27. 27. • Prototype 2nd-order equation: • Natural frequency: • Damping ratio: Natural Frequency and Damping Ratio ( ) ( ) ( ) ( ) ( ) ( ) 0 0 1 1 1 1 1 1 F ppo F Fi p F p p K ss K F s K sG s NK T s KG s H s K F s K Ns sNs K sNs ωφ φ ω ω   +   = = = = = + +  + + +    +    0 F p n p K N ω ω ω ω= = 0 1 1 1 2 2 2 p p p n F N K ω ω ς ω ω ω = = = It is geometric mean of the loop bandwidth in the absence of a filter and the filter corner frequency. 2 2 2 2 2 2 2 2 p F p F n p F n n n n p K K N K s s s s s s N ω ω ω ω ςω ω ςω ω ω = = = + + + + + + Characteristics equation Department of Electronic Engineering, NTUT27/55
  28. 28. • The poles of the closed-loop function: • As long as damping ratio is greater than one, the poles are real and a tangential plot of closed loop gain looks like: Closed-Loop Gain for Large Damping 20log N 6 dB/oct− 12 dB/oct− log mω ( )out ref f s f 2 2 0 2 1 1 1ω ω ς ς = − −   2 1 1 1 2 pω ω ς = + −   The characteristics are similar to the case with no loop filter, except for the increasing rate of attenuation in fout/fref beyond approximately the filter corner frequency . 2 1p n ns ςω ω ς= − ± − 0ω pω pω Department of Electronic Engineering, NTUT28/55
  29. 29. • As decreases toward , the damping ratio decreases and the phase shift at increases. Correspondingly, the transient response of the loop becomes less damped (more ringing) and the response peaks near . Close-Loop Responses 0 F p n p K N ω ω ω ω= = 0 1 1 1 2 2 2 p p p n F N K ω ω ς ω ω ω = = = pω 0ω 0ω nω m nω ω 0.25 0.5 1 2 4 12 dB 6 dB 0 dB −6 dB −12 dB −18 dB −24 dB 0.1ς = 0.3 0.5 0.7 1 ( ) 1 out ref f N f ω 0mω ω 0.1ς =12 dB 6 dB 0 dB −6 dB −12 dB −18 dB −24 dB −30 dB 0.25 0.5 1 2 4 ( ) 1 out ref f N f ω 0.3 2 0.5 0.7 1 Tangent for no filter Department of Electronic Engineering, NTUT29/55
  30. 30. Relative Stability – Phase Margin • Right figure shows the phase margin (relative stability) as a function of the damping factor. More highly damped loops are safer, in that more parameter variation is allowable before instability occurs. log mω ( ) ( )m mG Hω ω 6 dB/oct− 1020log FK N 0 dB Lω pω1mω = 90− 135− 180− ( )mG ω∠ 12 dB/oct− Phase margin • With a single-pole low-pass filter, the loop is inherently stable, sine −180o phase shift cannot be attained for any finite frequency. (not always true practically) Department of Electronic Engineering, NTUT ς 0.2 0.3 0.5 0.7 1.0 2.00.1 10 20 30 40 50 60 70 80 0 90 1 4 1 1 90 tan 1 1 2 4 PM ς −   = − + −     PhaseMargin(degrees) 30/55
  31. 31. Transient Response 0t = t oldf newf Synthesizer output frequency A B C D Overshot Ringing • Lower damping ratio brings a higher percent overshoot can cause the loop to go out of lock. (more unstable) • Narrower bandwidth with smaller damping ratio and longer settling time. Department of Electronic Engineering, NTUT 1N 2N 1 refN f 2 refN f ( )2 1N N> e 1 2 1 ref N f N   −    t reff outf( )G s 1 N e 31/55
  32. 32. Settling Time ( ) ( )0 02.3 log f T f Tω ∆ ∆ ≃ oldf newf ( )0f∆ 0t = t T= ( )f T∆ 3 -1 1 MHz/cycle 10 secFK = = 10 kHzreff = 11 MHz 10 MHzoutf = → Find the settling time for the output frequency of 10.1 MHz is attained: 6 0 10 1000 1000 FK N ω = = = 0 2.3 1 log 2.3 ms 0.1 T ω =≃ Department of Electronic Engineering, NTUT • Settling time: The frequency error changes one decade approximately each 2.3 time constant, i.e., • Example (no filter): 32/55
  33. 33. • The addition of a pole in the transfer function causes the slope to drop at a rate of −6 dB/oct whereas the addition of a zero has the opposite effect. • The open loop transfer function is: • The closed-loop transfer function is: A Pole-Zero Filter ( ) ( ) ( ) 1 1 z p s F s s ω ω + = + ( ) ( ) ( ) ( ) ( ) 1 1 p v zp v p K K sK F s K G s H s Ns Ns s ω ω +  = =  +  pω zω 6 dB/oct− 12 dB/oct− log mωzωpω 6 dB/oct− ( ) ( ) ( ) 0 01 po i p K F s K s T s K F s K Ns θ θ = = + Department of Electronic Engineering, NTUT33/55
  34. 34. Open-Loop Gain with a Pole-Zero LPF (I) ( ) ( ) 2 1 1 p F z p F p F p z K s NT s N K K s s N N ω ω ω ω ω ω + =   + + +    0 F n p p K N ω ω ω ω= = 1 2 p n n z ω ω ς ω ω   = +    6 dB/oct− 12 dB/oct− log mω zωpω 6 dB/oct− ( )2 2 2 1 2 z n n n s N s s ω ω ςω ω + = + + Department of Electronic Engineering, NTUT • Given the pole frequency location, a zero can be placed after the so as to avoid the magnitude from crossing the unity gain axis at a slope of −12 dB/oct, and therefore avoiding instability. To determine the closed loop response, simply plot T(s): 34/55
  35. 35. • From the results: Selecting the pole frequency sets the natural frequency and subsequently the loop bandwidth. Selecting the zero (based on the pole location in the open loop gain response) determines the desired percentage overshoot. • Therefore, a pole-zero filter allows the designer to select the loop bandwidth and the damping factor independently and still achieve stability. Open-Loop Gain with a Pole-Zero LPF (II) 0 F n p p K N ω ω ω ω= = 1 2 p n n z ω ω ς ω ω   = +    Department of Electronic Engineering, NTUT35/55
  36. 36. • The simplest PLL is the type-I loop because the open-loop gain has one pole at DC (pure integration). It is also a first-order loop because the open-loop gain has one significant pole. Type and Order of the Loop ( ) ( ) ( ) ( ) ( ) ( )1 G s N s T s G s H s D s = = + ( ) 1 1 0 n n n nN s a s a s a− −= + + +⋯ ( ) 1 1 0 m m m mD s b s b s b− −= + + +⋯ Order = m (m roots) Type = n (n roots at DC) ( )ref sφ ( )out sφpK ( )F s vK s 1 N PFD LPF VCO Department of Electronic Engineering, NTUT36/55
  37. 37. • First order, type I (no loop filter) First-order Type I Lω log mω0 dB G 6 dB/oct.− 1R iv ov 2R 2 1 2 o LPF i v R G v R R = = + A− 3R iv 4R ov′ A → ∞ 4 3 o LPF i v R G v R ′ ′− = = − Department of Electronic Engineering, NTUT37/55
  38. 38. • Second order, type I (lag-and-lead filter) Second-order Type I (I) 1R iv ov 2R 1C A− 3R iv 5R ov 2C 4R Lω log mω0 dB G 12 dB/oct.− 6 dB/oct.− zωpω 1 1 z LPF p s G s ω ω + = + 2 1 1 z R C ω = ( )1 2 1 1 p R R C ω = + 1 1 z LPF p s G s ω ω + ′ ′− = + ′ 4 5 2 4 5 1 z R R C R R ω′ = + 5 2 1 p R C ω′ = Department of Electronic Engineering, NTUT38/55
  39. 39. • Second order, type I (lag filter) Second-order Type I (II) Lω log mω0 dB G 6 dB/oct.− pω 12 dB/oct.− 2 1 2 1 1 LPF p R G sR R ω − = + + 4 3 1 1 LPF p R G sR ω ′− = − + ′ 1 1 2 1 1 1 p C R R ω   = +    4 2 1 p R C ω′ = Department of Electronic Engineering, NTUT 1R iv ov 2R 1C A− 3R 4R ov′ 2C A → ∞ iv 39/55
  40. 40. • Second order, type II (integrator and lead filter) Second-order Type II Lω log mω0 dB G 12 dB/oct.− 6 dB/oct.− zω 1 1 1 1 z LPF s G R C s ω + − = − 2 1 z R C ω = Department of Electronic Engineering, NTUT A− 1R iv 2R ov C A → ∞ 40/55
  41. 41. • Third order, type II (integrator plus lead-lag filter) Third-order Type II Lω log mω0 dB G 12 dB/oct.− pω 12 dB/oct.− 6 dB/oct.− zω 1 1 1 1 1 z LPF p s G R C s s ω ω + − = −   +     3 3 1 1 1 z LPF p s G R C s s ω ω + ′ ′− = −   +  ′  ( )2 1 2 1 z R C C ω = + 2 2 1 p R C ω = 4 3 1 z R C ω′ = 5 4 1 p R C ω′ = Department of Electronic Engineering, NTUT A− 1R iv 2R ov 2C 1C A → ∞ A− 3R iv 4R ov′ 3C 5R 4CA → ∞ 41/55
  42. 42. PLL Phase Noise Model Department of Electronic Engineering, NTUT Frequency Divider Xtal ,ref nφ outφ 1 N PFD LPF ,pfd nV ,op nV ,vco nφ ,div nφ VCO + ++ + + ( ) ( ) ( )( ) ( ), , , , , ,out n pfd n op n ref n div n e vco n d T s V V T s H s K φ φ φ φ= + + + + ( ) ( ) ( ) ( )1 G s T s G s H s = + ( ) ( ) ( ) 1 1 eH s G s H s = + ( ) ( ) ( ) ( ) ( ) for for1 c c NG s T s G sG s H s ω ω ω ω << = ≈  >>+  ( ) ( ) ( ) ( ) for1 for1 1 c e c N G sH s G s H s ω ω ω ω  << = ≈  >>+   where Very useful results 42/55
  43. 43. Responses of Noise Transfer Functions cω log mω ( ) ( ) ( )1 G s G s H s+ N ( )G s Transfer function multiplying all sources except VCO cω log mω ( ) ( ) 1 1 G s H s+ 1 ( ) ( ) 1 G s H s Transfer function for VCO cω log mω 20log N ( )dB 0 Department of Electronic Engineering, NTUT Other sources dominate inband noise VCO dominates outband noise 43/55
  44. 44. Typical VCO and PLL Noise Performance Department of Electronic Engineering, NTUT ( )L f∆ (dBc/Hz) VCO PLL ( )log f∆1 kHz 10 kHz 100 kHz 1 MHz 10 MHz −120 −90 −60 −30 dB/dec −20 dB/dec 44/55
  45. 45. Phase-Locked Loop Architectures Department of Electronic Engineering, NTUT • Integer-N PLL • Offset PLL • Fractional-N PLL • DDS offset PLL • Dual loop PLL • Multi-loop PLL 45/55
  46. 46. • Lower division ratio N to reduce inband phase-noise gain • Extend bandwidth with different • Avoid LO pulling Offset PLL out ref offsetf N f f= ⋅ + offsetf Department of Electronic Engineering, NTUT PFD LPF Frequency Divider reff outf /N offsetf ? 46/55
  47. 47. Offset PLL − Dual-Loop PLL (I) Department of Electronic Engineering, NTUT PFD LPF Frequency Divider reff outf /N PFD LPF Frequency Divider offsetf /M 47/55
  48. 48. Offset PLL − Dual-Loop PLL (II) PFD LPF Frequency Divider 1reff outf /N PFD LPF Frequency Divider offsetf /M 2reff 48/55 Department of Electronic Engineering, NTUT
  49. 49. Offset PLL − Hybrid DDS/PLL PFD LPF Frequency Divider reff outf /N DDS offsetf 49/55 Department of Electronic Engineering, NTUT
  50. 50. Multi-Loop PLL Department of Electronic Engineering, NTUT PFD LPF Frequency Divider reff outf /N 1offsetf PFD LPF /M PFD LPF /P 2offsetf 3offsetf 50/55
  51. 51. • Lower division ratio N to reduce inband phase-noise gain • Effectively produce a fractional division value • Generally employee a delta-sigma modulator for division ratio dithering Fractional-N Frequency Synthesis Department of Electronic Engineering, NTUT PFD LPF Dual-modulus Frequency Divider reff outf /N, (N+1) FCW 51/55
  52. 52. • DDS acts a reference source or phase/frequency modulator • A variable reference frequency source can drive a fractional frequency output. DDS-Driven Fractional-N Synthesizer Department of Electronic Engineering, NTUT PFD LPF Frequency Divider reff outf /N Hybrid DDS/PLL FCW DDS 52/55
  53. 53. DDS-feedback Fractional-N Synthesizer 2 out ref offsetn M f f f= ⋅ + Department of Electronic Engineering, NTUT PFD LPF DDS as a Frequency Divider reff outf DDS FCW Hybrid DDS/PLL • DDS acts a frequency divider • DDS output frequency: 53/55
  54. 54. Comparison of Frequency Synthesizers DDS Single-Loop PLL Multi-Loop PLL DDS/DAS DDS Offset PLL DDS Driven PLL BW (output) Narrow < 100MHz Broad > 1GHz Broad > 1GHz Broad > DDS Broad (carefully design) Broad Resolution Extremely Fine < 0.02 Hz Very Course > 250kHz (typical) Medium > 1kHz (typical) Extremely Fine < 0.01 Hz Extremely Fine < 0.01 Hz Extremely Fine < 1Hz Switching Time Very Fast < 100 ns Fast < 100us (typical) Very Slow > 1ms (typical) Very Fast < 1us (limited by RF switch) Fast < 100us (typical) Trade-off vs close-in spurious tones Spurious Noise < 75dBc (limited by DAC) Very Good Good (carefully design) Minimum Close-in Spurious Minimum Close-in Spurious Excellent over Broad Bandwidth Phase Noise Better than clock reference Very Good Very Good Very Good Very Good Good Circuitry Simple Simple Very Complex Moderate Moderate Moderate Department of Electronic Engineering, NTUT54/55
  55. 55. Summary • In this chapter, the basic idea of the phase-locked loops derived from the feedback system was introduced. One may put attention on that the PLL is processing the phase (or frequency) by transforming the phase error into voltage or current signals. • The PLL transient response and the phase noise were also presented in this chapter. We can simply conclude that, as loop bandwidth increases, the locking time is going faster and the VCO inband phase noise can be suppressed with more attenuation while the other components contribute more noise within the loop bandwidth. Department of Electronic Engineering, NTUT55/55

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