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Multiband Transceivers - [Chapter 6] Multi-mode and Multi-band Transceivers


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Multi-mode and Multi-band Transceivers

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Multiband Transceivers - [Chapter 6] Multi-mode and Multi-band Transceivers

  1. 1. 李健榮 助理教授 Department of Electronic Engineering National Taipei University of Technology Multiband RF Transceiver System Chapter 6 Multi-mode and Multi-band Transceivers
  2. 2. Outline • System Level Considerations • Wideband LO Generation • Building Blocks of TX and RX • Isolating Techniques on ICs • RF Power Amplifier • Two multi-mode examples of the transceiver are introduced in this chapter. Department of Electronic Engineering, NTUT2/82
  3. 3. Software-Defined Front-ends • The ultimate dream of every SDR front-end is to deliver an RF transceiver that can be reconfigured into every operating mode. Modes for cellular (2G–2.5G–3G and further), WLAN (802.11a/b/g/n), WPAN (Bluetooth, Zigbee, etc.), broadcasting (DAB, DVB, DMB, etc.), and positioning (GPS, Galileo) functionalities. Obviously, each of them has different center frequency, channel bandwidth, noise levels, interference requirements, transmit spectral mask, and so on. • As a consequence, the performances of all building blocks in the transceiver must be reconfigurable over an extremely wide range. Linearity, filtering, noise, bandwidth, and so on, can be traded for power consumption. Department of Electronic Engineering, NTUT3/82
  4. 4. System Level Considerations • A first choice to be made is the radio architecture to be used. Heterodyne, homodyne, low-IF, and other architectures, which one to choose? • In view of SDR, this question perhaps becomes a little easier to answer. When the characteristics of all possible standards are taken into account, not a single IF can be found that suits them all. Having multiple IFs and the associated (external) filtering stages increases the hardware cost of the SDR, which cannot be tolerated. Thus, the direct-conversion architectures are the right choice for the job. Department of Electronic Engineering, NTUT4/82
  5. 5. Vision of the SDR Transceiver • The transceiver core implemented in CMOS includes a fully reconfigurable direct conversion RX, TX, and two synthesizers (for FDD). The functions that cannot be implemented in CMOS are included on the package substrate. These are related primarily to the interface between the active core and the antenna. They must provide high-Q bandpass filtering or even duplexing, impedance- matching circuits, and power amplification. MEMS switches Tunable matching Tunable filtering Power amplifier DMQ VCO Distr. DMQ NoC controller Frac-N PLL Frac-N PLL MCM substrate CMOS IC Department of Electronic Engineering, NTUT5/82
  6. 6. Hard Works • Determine performance specifications for each block. • The total budget for gain, noise, linearity, and so on, must be divided over all blocks, ensuring that all possible test cases are covered for every standard. • Having very flexible building blocks helps a great deal, but making a smart system analysis is crucial to obtaining an optimal solution. • Gain ranges and signal filtering must be set such that the signal levels are an optimal trade-off between noise and distortion. • With the built-in flexibility, a software-defined radio can achieve state-of-the-art performance very close to that of dedicated single-mode solutions. Department of Electronic Engineering, NTUT6/82
  7. 7. Wideband LO Synthesis • Example: To generate all required LO signals in the range 0.1 to 6 GHz, several frequency generation techniques have been proposed to relax the tuning range specifications of a voltage-controlled oscillator (VCO). They use division, mixing, multiplication, or a combination of these. However, to make these systems efficient in terms of phase noise and power consumption, the VCO tuning range still has to be maximized. Department of Electronic Engineering, NTUT7/82
  8. 8. Frequency Tuning Capacitor • Frequency tuning of LC VCOs is commonly done by changing the capacitance value of the varactors and/or an array of switched capacitors in the tank. Switched or controlled inductor designs remain difficult to cover the desired wideband continuously and to limit the deterioration of the phase noise performance caused by the insertion of these switches. • Instead of using a single large varactor to tune the frequency, a mixed discrete/continuous tuning scheme is usually chosen. A small varactor is used for fine continuous tuning, and larger steps are realized by digitally switching capacitors in and out of the resonant tank. This has two advantages: The VCO gain is lower, allowing easier phase-locked loop (PLL) design, and digitally switched varactors have a higher ratio between the capacitance in the on-state (Con) and the capacitance in the off-state (Coff ). A higher Con/Coff ratio allows a larger VCO frequency tuning range. Department of Electronic Engineering, NTUT8/82
  9. 9. Tank Loss Variations • In the target frequency range (< 5 GHz), the losses in the oscillator tank are usually dominated by the inductor. This simplification is, of course, not completely valid, since extra losses due to the skin effect, for examples will increase the resistance at higher frequencies. • The negative resistance needed to compensate for the inductor losses is given by Gm = RS(ωC)2. • If we want the oscillation frequency to change by a factor of 2: Total capacitance of the resonant tank has to be changed by a factor of 4 Required negative resistance must also change by a factor of 4. The transconductance required for the active core is four times higher at the lower end of the frequency tuning range than at the higher end. Department of Electronic Engineering, NTUT9/82
  10. 10. Wideband VCO Architecture Scale the core biasing current. Change the transistors sizes. Keep parasitics at a minimum (phase noise and the tuning range achievable). Switches to turn transistors on or off. Switches has to avoid degrading the oscillator phase noise as well as to ensure parasitic capacitances are small. For lower frequencies, more and more core units are gradually activated, and the total bias current increases to keep the oscillation amplitude steady and the parasitic capacitance increases, helping the “normal” varactors in their goal to increase the total tank capacitance. M1 M2 SW1 SW1 M3 M4 Dunit Ckvco Vtune 0 1 1 0 Dtune Department of Electronic Engineering, NTUT10/82
  11. 11. Frequency Tuning Sensitivity Variations • Variation in VCO sensitivity for wide-tuning-range VCOs: A change in the control voltage Vtune results in a change C in the analog varactor capacitance Cvar . This causes a change in frequency f : • The VCO with a frequency ratio of 2, tank capacitance has to change by a factor of 4, the VCO frequency sensitivity will then change by a factor 4 √4 = 8. Such a large change in VCO gain presents serious problems for the design of the PLL in which it will be incorporated. It prevents keeping the PLL bandwidth constant and hence endangers the loop stability and an optimal phase noise performance. 1 1 2 4 f f CLC C LCπ π ∆ − = ⇒ = ∆ Department of Electronic Engineering, NTUT11/82
  12. 12. 0.1 to 6-GHz Quadrature Generation • The divide/multiply and quadrature (DMQ) contains several divide-by-2 blocks. They generate I and Q phases down to a division factor of 32. DIV2DIV2DIV2DIV2DIV2 PPF PPF DIV2 BUF 4G 5G:3G 2G 4G:2G 4G 4G 1G 0.5G 250M 1.5G 125M • The DMQ further employs a quadrature-phase single-side band (SSB) mixer to generate 3GHz, 5GHz. • The SDR’s LO frequency can be selected by a multiplexer integrated in the DMQ. Department of Electronic Engineering, NTUT12/82
  13. 13. Receiver Building Blocks • A key aspect for the receiver RF part is its interference robustness. The blocking requirements for simultaneous multi- mode operation imply the need for tunable narrowband circuits at the antenna interface. Either this function can be provided by a multi-band filtering block, in which case the receiver’s input can be a wideband LNA, or part of this burden can be taken up in the LNA design. Department of Electronic Engineering, NTUT13/82
  14. 14. MEMS-Enabled Dual-Band LNA (I) • Using MEMSs switches to build a low-loss reconfigurable antenna filter section on a thin-film substrate. Packaged MEMS switch : connect the LNA to either its 1.8 GHz or 5 GHz matching circuit and antenna filter. The loss of the switch is only 0.2 dB. Cx reduces the gate inductance for the input matching. At 1.8-GHz, a simple matching made up of one or two passive components can fulfill the matching requirement. The bondpad is modeled by a 65-fF cap in series with a 50-Ohm resistance. Each bondwire is modeled by a 1.3-nH inductance. Input Stage Single to Diff. Conv. A BA B Gain Ctrl 5-6 GHz 1.8 GHz Cx 5-6 GHz MEMS SPDT 1.8 GHz C bp Low-band 50 -TL Matching Network Lbond Switchable Matching Network On board On chip High-band Freq. [GHz] 0 1 2 3 4 5 6 7 8 0 -2 -4 -6 -8 -10 -12 -14 -16 -18 -20S11[dB] Department of Electronic Engineering, NTUT14/82
  15. 15. MEMS-Enabled Dual-Band LNA (II) Internally, the LNA has two separate outputs to cover the required frequency range. A resistively loaded output is small in area and wide bandwidth but can only provide enough gain at frequencies up to 2.5 GHz This output is for the 5 to 6-GHz band with an LC-tuned load. A resistor in parallel with this inductor lowers its Q to cover the 1-GHz bandwidth. Gain switching: When the third CG- transistor is activated, which bypasses a certain fraction of the signal current to the power supply so as to reduce the gain. The overall gain is 24 dB. S11 input matching better than −10 dB is achieved in both bands. The simulated LNA NF is around 2 dB, while the IIP3 value is −5 dBm in the low band and 3 dBm in the high band. Input Stage Single to Diff. Conv. A BA B Gain Ctrl 5-6 GHz 1.8 GHz Cx 5-6 GHz MEMS SPDT 1.8 GHz C bp Low-band 50 -TL Matching Network Lbond Switchable Matching Network On board On chip High-band Freq. [GHz] 0 1 2 3 4 5 6 7 8 0 -2 -4 -6 -8 -10 -12 -14 -16 -18 -20 S11[dB] Department of Electronic Engineering, NTUT15/82
  16. 16. Wideband LNAs (I) • Rely on the passives in the antenna interface for RF interference and blocking filtering. This makes the realization of the concept easier, as commercially available (multi-band) filtering blocks can be used in the implementation. • Wideband LNA must now be used that cover an RF frequency range as large as possible for optimal flexibility, but must still achieve state-of-the-art performance with respect to narrowband LNAs. • Covering the full 100 MHz to 6 GHz frequency range is challenging since achieving a low NF at hundreds of MHz requires large transistors with low 1/ f noise On-chip LC-matched common-source (CS) LNAs typically cover a bandwidth from 3 to 10 GHz. Extending the bandwidth down to 100 MHz would require prohibitively large inductors and thus chip area. Department of Electronic Engineering, NTUT16/82
  17. 17. Wideband LNAs (II) • Two LNAs are combined to cover the entire frequency range: An inductor-less feedback LNA with a small form factor covers frequencies from 100 MHz to 2.5 GHz, and a CS LC-matched LNA covers frequencies from 2.5 to 6 GHz. Only one LNA is powered at a time, to save power and provide filtering over half of the bandwidth. Resistive Feedback LNA OUT IN C Bandgap OUT IN VC Bandgap LC-Matched LNA Department of Electronic Engineering, NTUT17/82
  18. 18. Resistive Feedback LNA – 0.1 ~ 2.5 GHz A digitally controlled bank of feedback resistors allows us to switch from high- to intermediate- and low-gain modes. The biasing is done with a 3-bit programmable current source. This allows us to vary the gain in small steps around the different gain modes and to decreasing the power by half when switching from high- to low-gain mode. At a maximum gain of 22 dB, typical simulation results achieve an NF of 2 dB and an IIP3 of −10 dBm at a power consumption of 12 mW. At reduced gain (10 dB), the linearity improves to +3 dBm while the power consumption decreases to 8 mW. OUT IN C Bandgap It employs resistive feedback for wideband matching and noise canceling for low NF over a wide band. It in general has lower gain and a higher noise figure than these of inductively matched narrowband designs, but it offers large savings in area. Department of Electronic Engineering, NTUT18/82
  19. 19. LC-Matched LNA – 2.5 ~ 6 GHz Broadband input-matching is achieved by the inductively degenerated CS-stage into an LC bandpass filter . Input matching from 6 GHz down to 2.4 GHz can be done with inductive elements of reasonable values, but extending that frequency band to lower values is practically not feasible. At the output, a 4- bit programmable capacitor bank provides filtering. A pullup resistor is added to obtain good linearity. Biasing is done with a 3-bit programmable on- chip voltage reference. OUT IN VC Bandgap Simulated values for NF and IIP3 are 2.4 dB and −10 dBm, at a maximum gain of 22 dB, with a power consumption of 12 mW. Gain switching is achieved with a bypass cascode transistor that diverts a part of the signal current to the power supply for lower gain without influencing the input matching Department of Electronic Engineering, NTUT19/82
  20. 20. RF LO B:1 1:B RF+ RF- LO- LO+ LO+ io+ io- Wideband Down-Conversion Mixer • The Gilbert cell is is used for wideband operation up to 6 GHz. An NMOS input pair is used as a transconductance, driving RF signal current into the core switch transistors that form the Gilbert cell The folded switching PMOSs can reduce flicker noise. The extra folding transistors will contribute a certain amount of thermal noise, causing the overall receiver’s NF to deteriorate. The noise contributions in a switching mixer are not easy to understand or analyze but can generally be kept within limits by using large LO signals and reduced dc current through the switching transistors. The switchable gain is achieved by digitally programmable current gain B. The input must be designed carefully, as it will determine both the noise and the linearity performance of the mixer. A considerable biasing current of 5 mA. Department of Electronic Engineering, NTUT20/82
  21. 21. Signal Selection and Dynamic Range • How to handle: Signal Selection and Dynamic Range • Signal Selection Capturing a slice of bandwidth while rejecting adjacent frequencies, which sometimes contain signals of higher power than the signal of interest. • Dynamic Range Defined by the max. and min. signal levels that the receiver can process without distortion that would degrade the SNR to an unacceptable level. • Heterodyne Receivers 1. Convert an RF signal into an IF. The IF is passed through high-Q BPFs to remove undesired signals such as the image and interfering signals. 2. This IF approach works well for systems with defined channel bandwidths. But large banks of fixed filters would be required to cover the broad range of possible channels in SDR applications. This is not a practical solution. Department of Electronic Engineering, NTUT21/82
  22. 22. Variable-bandwidth Problem • One possible solution: switched capacitor circuits. The bandwidth is programmable by varying the capacitor ratio and the clock frequency of the switched capacitor circuit. • Another way to handle varying channel bandwidths is with the use of direct conversion receivers (DCRs). Since no image frequency is produced, thus RF preselect filters can be eliminated. The removal of adjacent channel energy no longer requires high-Q BPFs but can be accomplished with LPFs, which are much easier to integrate. This is a great advantage because it is possible to integrate LPFs with programmable gain and bandwidth in today’s technology. Department of Electronic Engineering, NTUT22/82
  23. 23. Problems of DCRs • Challenges: DC offsets and 1/f noise • DC offsets are caused primarily by mismatch and/or by LO signal coupling into the mixer RF port. The undesired effect is saturation of the following dc-coupled gain stages. • 1/f noise is the dominant source of noise in MOS transistors at frequencies below 100 kHz. In most CMOS processes, PMOS devices have between 2 and 5 times less 1/f noise than do NMOS devices. Where this is true, PMOS devices should be used in parts of the circuit where reducing 1/f noise is critical. Noise has a cumulative effect in a gain lineup, so the gain in the first stage should be as large as possible. Chopper stabilization can greatly reduce the effects of any remaining 1/f noise. Department of Electronic Engineering, NTUT23/82
  24. 24. Transmitter Building Blocks • The pre-power amplifier (PPA) is the final block in the SDR transmit path. The effective output power is not as high, and for many applications the average swing is much lower than the peak. Furthermore, a non-negligible voltage drop across the series resistance of the inductor sets the dc output voltage below the power supply. INN INP OUT The PPA includes extensive programmability of gain settings. The output stage is an inductively loaded CS amplifier with programmable bias current for optimal linearity vs. power trade-off. Department of Electronic Engineering, NTUT24/82
  25. 25. DCFB IN OUT Ref Programmable Gain PPA • The pre-power amplifier provides gain programmability. The core of the amplifier is a CS stage with a PMOS resistive load. 3 additional PMOS transistors are placed in parallel with the main load to control the gain. Their gates can be connected to Vdd , to turn them off and increase the gain, or to ground, to put them in the linear region and decrease the gain. Changing the resistive load has an impact on the dc voltage and so, on its linearity. A dc level feedback circuit (DCFB) controls gate bias (for enhance linearity). The total bias current through the amplifier can be controlled to optimize the power consumption for the linearity required. The performance of this circuit varies widely, of course, over carrier frequency, required output power, bias and gain settings, and so on. Simulation results indicate a total gain range of 50 dB and typical IM3 distortion levels of −35 dB at 0-dBm output power. Department of Electronic Engineering, NTUT25/82
  26. 26. Direct Conversion Transmitters (I) • BB digital process provides complex samples of the encoding intended, including encryption, pulse shaping, and linearization preconditioning. These discrete samples are lowpass-filtered and amplified by the post-baseband (PBB) amplifier. Amp LPF Amp LPF RF Power Amp Differential Quadrature LO I Q BaseBanddigitalprocessing &Digital-to-Analogconverter 0 ° 90 ° Frequency dBmlevel Occupied Signal Bandwith Far Out Noise Level dBc/Hz Carrier frequency Department of Electronic Engineering, NTUT26/82
  27. 27. Direct Conversion Transmitters (II) • The benefit of this direct-launch Cartesian encoded carrier is multi-mode compatibility with baseband frequency bandwidth and mask determination. This enables additional digital processing technology to be applied through the entire transmitter, such as feedforward or predistortion linearization. • Any form of amplitude, angle, frequency, or any combination of modulation formats and bandwidths with no exceptions, including complex non-continuous multi-channel signals. Department of Electronic Engineering, NTUT27/82
  28. 28. Direct Conversion Transmitters (III) • Three main issues: Sideband noise level Local oscillator feedthrough Self-generated interference (LO pulling causes remodulation of the carrier) • Sideband noise level In practice, it is introduced when circuit noise is increased in level by the broadband gain in the transmitter system. • Local oscillator feedthrough The transmission gate ring switching mixer can improve carrier feedthrough. With careful design of the mixer, carrier feedthrough of better than −50 dBc is attainable. • LO pulling Shielding, grounding, .etc. Department of Electronic Engineering, NTUT28/82
  29. 29. Far-out Sideband Noise Contribution • Far-out sideband noise can become an interference to receivers close to the transmission signal. (For GSM, a BPF is often used to reduce the far-out noise significantly outside TX bands.) • The lack of broadband tunable RF bandpass filters results in far-out noise over a very wide range of frequencies and the TX may offend a receiver in close proximity. Amp LPF Amp LPF PADifferential Quadrature LO I Q BaseBanddigitalprocessing &Digital-to-Analogconverter 0 ° 90 ° Gain = G Low Pass Filter Reduces Far Our noise contribution Noise Figure contributes to Input Referenced Added Noise Transmission Gate Switching Mixers Input Referenced Added Noise Carrier VCO Department of Electronic Engineering, NTUT29/82
  30. 30. LO Frequency Pulling • DCTs have an output frequency equal to that of the oscillator signal source frequency. The interference causes perturbations in the VCO that are not corrected by the PLL control loop. This undesirable remodulation of the VCO signal frequency will degrade the quality of the signal transmitted. Amp LPF Amp LPF PA Differential Quadrature LO I Q BaseBanddigitalprocessing &Digital-to-Analogconverter 0 ° 90 ° Transmitter radiated signal coupled input VCO signal source Antenna Network DC supply and ground conducted into VCO signal source network Electromagnetic Shielding Carrier VCO Department of Electronic Engineering, NTUT30/82
  31. 31. Remodulation from Pulling • Overcome Remodulation: Shielding Grounding Decreased VCO sensitivity to electromagnetic signals and the use of subharmonic, higher harmonic, or translated reference signal frequency. Lowering the inductive and capacitive coupling coefficient of the VCO resonant network, use of differential VCO, and integrated implementation with a lower coupling area profile. • These are combined with multiple layers of isolation shielding between the antenna and the VCO to form a remodulation rejection system. Department of Electronic Engineering, NTUT31/82
  32. 32. Required Isolation in ICs Substrate modulator PA LO ωLO paraC PLL synthesizer Isolation > 90~110 dB PA < 30 dBm (1W) LO < 10 dBm In the experiment, LO phase noise degrades when the injection power is as low as −80 dBm. Department of Electronic Engineering, NTUT32/82
  33. 33. Substrate Coupling and Isolation • Dominant Diffusion capacitive coupling Impact ionization Inductive coupling (power grid fluctuations) • Less significant Gate-induced drain leakage (GIDL) Photon-induced reverse current Diode junction leakage current Department of Electronic Engineering, NTUT33/82
  34. 34. Diffusion Capacitive Coupling • SPICE models such elements with "CJ0" and "CJSW" (source/drain-to- substrate capacitance). • Metal-to-metal capacitors: Largest parasitic capacitance to the substrate, hence if these devices are used for implementing large on-chip capacitors, they can act as significant substrate noise- injectors. • As technology feature size reduces, higher doping concentration leads to higher depletion capacitance and hence more coupling effects. ( ) 1 2 1 22 js b q N N C V N N ε ψ   =   + +  Department of Electronic Engineering, NTUT34/82
  35. 35. Impact Ionization • Reduced transistor feature sizes increase the electric field in the channel and therefore impact ionization currents are becoming more significant compared to other injection mechanisms. • In saturation, impact ionization takes place with a high electric field in the depleted region. For a p-type substrate, the generated holes are swept to the substrate generating an effective drain-to-substrate current. Recent experimental evidence suggests that hot-electron induced substrate currents are the dominant cause of substrate noise in NMOSFETs up to at least one hundred megahertz. • Shorter device channel lengths in advanced technologies are likely to increase the impact ionization currents due to increased channel fields and smaller oxide thickness and drain junction depth. Department of Electronic Engineering, NTUT35/82
  36. 36. Inductive Coupling - Power Grid Fluctuations • Due to parasitic effects (mainly bond wire inductance), power supply lines become very noisy because of currents drawn by the switching digital circuits. These currents induce large voltage glitches when they switch (Ldi/dt noise) at substrate and well contacts. In addition, the power grid noise can be also capacitively coupled through metal-to-substrate parasitic capacitance. Department of Electronic Engineering, NTUT36/82
  37. 37. Isolation in Silicon Substrate (Baseline) • The baseline: D = 120 µm Isolation ~ 29 dB As the frequency increases, the isolation is getting more and more worse. In the following slides, some isolation techniques were applied to compare the isolations. Baseline Isolation Department of Electronic Engineering, NTUT37/82
  38. 38. Isolation Techniques – P+ Guard Ring • P+ ring: D = 120 µm W = 3 µm / d = 10 µm Isolation ~ 65 dB • Guard rings sink the noise currents P+ ring w/ low ohmic contact Isolation w/ P+-ring Department of Electronic Engineering, NTUT38/82
  39. 39. Isolation Techniques – N+ Guard Ring • N+ ring: D = 120 µm W = 3 µm / d = 10 µm Isolation ~ 65 dB • Good isolation at low frequencies due to the high capacitive impedance of the p-n junction between n-ring and p-sub. Isolation w/ N+-ring Department of Electronic Engineering, NTUT39/82
  40. 40. Isolation Techniques – Deep N-Well Ring • DNW ring: D = 120 µm W = 3 µm / d = 10 µm Isolation ~ 90 – 70 dB • Good isolation at low freq. • The DNW isolation degrades with increasing the frequency slower than the n-well due to that DNW is lightly doped than the regular n+ well.(p- n junction capacitance with the p substrate is smaller than that of the n+ well) Isolation w/ DNW N+-ring Department of Electronic Engineering, NTUT40/82
  41. 41. Isolation Techniques – Deep Trench • Deep trench: D = 120 µm W = 3 µm / d = 10 µm Isolation ~ 65 dB • A deep trench is a trench in the silicon substrate approximately 10 µm deep that is filled with oxide. • The oxide in the deep trench acts as a high impedance insulator that forces the substrate noise current to dive deep in the substrate. Department of Electronic Engineering, NTUT41/82
  42. 42. Isolation v.s. Distance D – Baseline Department of Electronic Engineering, NTUT42/82
  43. 43. Isolation v.s. D – P+ Guard Ring Department of Electronic Engineering, NTUT43/82
  44. 44. Isolation v.s. Ring Enclosure d Department of Electronic Engineering, NTUT44/82
  45. 45. Isolation v.s. Ring Enclosure Width w Department of Electronic Engineering, NTUT4582
  46. 46. Adopt the Offset PLL • Using an offset PLL eliminates remodulation by increasing the PLL loop bandwidth to include the signal modulation radiated. 0 °90 ° Quad Generator ReferencePhase Frequency Detector Narrow band LPF Synthesizer control Offset VCO Ft or 2*Ft F = Ft or 2*Ft +/− Ft Fr Very Wideband LPF Carrier VCO control Ft or 2*Ft Department of Electronic Engineering, NTUT46/82
  47. 47. Using DDS • An alternative direct-launch architecture that does not have a VCO operating at the transmitter output frequency uses direct digital signal synthesis (DDS). • Briefly stated, a high-frequency clocking signal is used to generate an output signal source without the use of an oscillator operating at the output frequency. DDS can be an all- band signal source for SDR, assuming that it is technically practical for up to 6-GHz operation with acceptable power dissipation. Department of Electronic Engineering, NTUT47/82
  48. 48. Cartesian and Polar Implementation • The advantage of polar processing is direct phase encoding on the signal source and the potential of an efficiency enhancement implementation in the amplitude modulation. First, the bandwidth associated with the magnitude and phase signals are multiples of the Cartesian equivalent signals. Second, the magnitude and phase terms are very different, which causes divergent signal processing effects, including potential time alignment imperfections. These bandwidth and time alignment effects require additional complexity and consideration for each mode of operation. Quadphase - Q Q Magitude = ( ) ( ) 1/22 2 = I t +Q t   InPhase - I Cartesian ( )A t ( ) ( ) ( )Phase= t = arctan Q t /I tθ    ( )v t Q 90° -90° 180° ( )A t ( )tθ ( )v t I 0° Polar Department of Electronic Engineering, NTUT48/82
  49. 49. RF Power Amplification (I) • Power Sources: Portable equipment: < 4 W of peak power (3.6- to 9-V dc battery) Fixed-base station equipment: < 100 W of peak power (110-V ac source) Vehicle mobile class: 10-W or higher (12-V dc battery) • As the number of modes and bands increases, the size, cost, and performance of a bank of single solutions will become prohibitive without technology advancement in the passive frequency-determining elements needed for matching at each port of the power transfer gain stages within the transmitter system. Department of Electronic Engineering, NTUT49/82
  50. 50. RF Power Amplification (II) • Combining bands to expand the RF PA across applications with an acceptable compromise of optimized single-solution performance. • Multi-mode operation would be divided between TDD or FDD. • The TDD switches between transmitter and receiver operation modes, where only one is active at any given time. Amp LPF Amp LPF Differential Quadrature BaseBanddigitalprocessing &Digital-to-Analogconverter I Q 0º 90º PA Tx RxF = F Mulitiplexer Amp LPF RxFDifferential Quadrature LNA 0º 90º Amp LPF I Q BaseBanddigitalprocessing &Analog-to-Digitalconverter TxF Department of Electronic Engineering, NTUT50/82
  51. 51. RF Power Amplification (III) • The FDD has simultaneous transmitter and receiver operation at different frequencies. Another incremental step would use programmable frequency-matching or device- operating conditions to expand the band or mode of application associated with a common RF power amplifier implementation. Amp LPF Amp LPF Differential Quadrature BaseBanddigitalprocessing &Digital-to-Analogconverter I Q 0º 90º PA Tx RxF = F Duplexer Amp LPF RxFDifferential Quadrature LNA 0º 90º Amp LPF I Q BaseBanddigitalprocessing &Analog-to-Digitalconverter TxF Department of Electronic Engineering, NTUT51/82
  52. 52. RF Power Amplification (IV) • The RF signal levels associated with power amplifiers can reach levels greater than the dc voltage source used for program control. • If the RF signal voltage is greater than that of the dc control range of the varactor, its capacitance is a function of the RF signal in combination with the dc control voltage. The capacitance is a function of the RF signal and is no longer constant at all times. This can result in modulation of the programmable element’s impedance value and distortion in the RF signals. Department of Electronic Engineering, NTUT52/82
  53. 53. RF Power Amplification (V) • A single RF PA would be an RF PA with continuous operation across all frequencies and optimized power dissipation at every multi-mode format across all operating power levels. In addition, it would provide acceptable linear performance and harmonic content for any mode or band of operation. • As the bandwidth approaches from 100 MHz to 6 GHz, a cascade of wide bandwidth-matching network gain stages becomes a complex design. Programmable elements could become part of the solution to provide a design that would compete successfully with a band of multi-mode or band- optimized performance solutions. • However, a vector combined distributed implementation (distributed amplifier) can provide wideband performance with a modest increase in complexity. Department of Electronic Engineering, NTUT53/82
  54. 54. Transmitter Efficiency • A linear class B amplifier peak power occurs ideally when the peak output voltage is equal to that of the dc supply voltage: • A modulation format with a peak-to-average ratio of 3 dB would have an efficiency reduction by a factor of 2. • Supply Modulation: Modulating the dc supply voltage as a function of the encoded output power magnitude to provide peak operating efficiency at all power levels. There are a number of supply modulation variations, such as envelope following, envelope elimination and restoration, and polar modulation format. 0.785 4 4 out out in dc P V P V π π η = = = = Department of Electronic Engineering, NTUT54/82
  55. 55. Transmitter Linearization • Error vector magnitude (EVM) and adjacent channel power ratio (ACPR) are used to measure the performance. • ACPR: A measure of the unintended generation of electromagnetic energy from the transmitter network into frequency bands adjacent to the intended operating frequency band. • A transmitter frequency-domain mask is defined by standards to limit transmitter adjacent channel interference signal level. The source of the unwanted adjacent channel energy is nonlinear distortion products within the RF power amplifier. Department of Electronic Engineering, NTUT55/82
  56. 56. Reducing Nonlinear Distortion • Reducing the output power reduces the ACPR at the expense of RF power amplifier efficiency, which is not always a practical solution. • Linearization adds complexity to the amplifier to compensate or cancel generation of the distortion components. Linearization Tech. Advantages Disadvantages Feedback 35 dB improvement Not depend on carrier frequency Narrowband, < 1 MHz Instability potential Dependent on output termination Feed-forward 35 dB improvement Not depend on carrier frequency Wideband < 100 MHz Efficiency performance Complexity Predistortion Reduced hardware Table-based Wideband < 10 MHz Limited improvement (~15 dB) Department of Electronic Engineering, NTUT56/82
  57. 57. Transmitter Stability • When a circuit becomes unstable, it produces output signals at frequencies not represented within the input signal. The most common form of unstable behavior in a circuit is positive feedback, where the output is coupled to the input at a level higher than the original input signal. As the bandwidth of operation increases, these unintended feedback paths become more difficult to avoid. • When an RF power amplifier is working, the operating condition is changing as a function of the input signal level. Therefore, the s-parameters are time-varying functions, making complete stability analysis more difficult to define. Department of Electronic Engineering, NTUT57/82
  58. 58. Broadband LO Generation • Perhaps the greatest challenge for SDRs is the generation of LO oscillator signals over broad, continuous frequency ranges. • Given that direct conversion is the receiver topology of choice, quadrature generation must be part of the solution. This is an area that requires innovation to achieve SDR performance. Department of Electronic Engineering, NTUT58/82
  59. 59. Phase-Locked Loops (I) • PLLs are the most common implementations of frequency synthesizers. A VCO is phase-locked to a stable reference frequency (typically, a xtal) through a feedback path. VCO architectures: Ring oscillators have found applications in some wireless LAN standards and broadband TV tuner applications. However, the LC-tuned oscillator is most commonly used in applications with stringent phase noise requirements. • In addition to the benefits of the PLL integration into CMOS processes, they typically exhibit good performance in terms of phase noise, current drain, area, and spurious performance. The greatest challenge for PLLs is making them tunable over a broad frequency range. Tuning ranges of 20% are typical but can be as high as 30%. Recent results suggest that ranges approaching 50% are possible. Department of Electronic Engineering, NTUT59/82
  60. 60. Phase-Locked Loops (II) • Even a 50% tuning range is not enough, there are a multitude of approaches for extending the frequency range of PLL systems. One approach is to have multiple VCOs, which in most cases can share the same PLL circuitry. • The size of inductors places a practical limit on the number of VCOs and therefore the achievable frequency range. Whether on- or off-chip, cost and manufacturing issues prevent the use of huge banks of VCOs to cover wide tuning ranges. • Another way to get a broad tuning range is to have one or more VCOs and multipliers, dividers, or mixers combinations. Inevitably, the addition of these circuits comes at the cost of higher power consumption for equivalent noise and spurious performance. Department of Electronic Engineering, NTUT60/82
  61. 61. Direct Digital Synthesizer (DDS) • DDSs have a broad frequency tuning range, fine frequency resolution, and very fast switching • ROM consumes a large amount of power. • Synthesized frequency is lower than the reference frequency. Acc. Output ACC n m LPF outF refF DACROM ROM Output DAC Output Acc.Output ROMOutput DACOutput outF Department of Electronic Engineering, NTUT61/82
  62. 62. Digital-to-Time Converter • Use of a digital-to-time converter (DTC) to construct an output frequency from the phase information in the accumulator to lower power consumption. Acc. Output ACC K n m Tap0 TapX Phase Detector Charge Pump LPF vtuneTapped Delay Line Tap0 Tap1 Tap2 …… TapX-1 TapX Tap Selection Logic …… Digital-to-Time Converter outF refF Department of Electronic Engineering, NTUT62/82
  63. 63. Example (I) • A highly reconfigurable low-power transceiver implemented in a 90-nm CMOS process. • The RFIC processes signals of multiple wireless protocols from 100 MHz to 2.5 GHz with −6 dBm and a voltage gain of 48 dB. • The transmitter has better than 40 dB of carrier suppression, 35 dB of sideband suppression, and an EVM of 1% at 800 MHz. • The frequency synthesizer uses direct digital synthesis to achieve instantaneous frequency switching and a phase noise of −115 dBc/Hz at 25 kHz offset for a 500-MHz carrier frequency. Department of Electronic Engineering, NTUT63/82
  64. 64. Rx DDS Reference PLL/VCO PMA VGA BQ DCOC MUX Tx TEST POINTS SPI ButterworthForward pole 2 Input Buffer Tx Forward DDS ReferenceTx Reverse DDSCartesian Rev Amp/Mixer Cartesian BB Forward Reference Example (II) Programmable LPF, 4 kHz ~ 10 MHzDDS DDS DDS PLL DCR DCT Cal. Cal. external connection to an ADC and digital processing. external connection to an DAC and digital processing. Programmable LPF, 4 kHz ~ 10 MHz, 10% BW step 90 dB VGA power control For external LNA Chopping Department of Electronic Engineering, NTUT64/82
  65. 65. Quadradture Mixers • The quadrature mixers on RX inputs 1, 3, and 5 are nonchopped passive mixers built with a quad ring of CMOS transmission gates. • The quadrature mixers on RX inputs 2 and 4 use dynamic matching to improve IP2, flicker noise, and dc offset. The chopping mixers are built with three mixers in, where each mixer is built with a quad ring of CMOS transmission gates. RFp LOm OUTp LOp OUTm RFm RFp RFm CHOPp CHOPm LOp LOm OUTm OUTp Department of Electronic Engineering, NTUT65/82
  66. 66. Baseband Filters (I) • Baseband filters that support multiple bandwidths are implemented along with gain control and dc offset correction. • A filter bandwidth is programmable from 4 kHz to 10 MHz in 6.25% steps or less. Sufficient margin is built into the design to allow for a 20% change in RC tolerance and still maintain the bandwidth range of 4 kHz to 10 MHz. • Bandwidth selection is implemented by adjusting the resistor and capacitor values in the filter design. Department of Electronic Engineering, NTUT66/82
  67. 67. Baseband Filters (II) • Baseband filter gain control is accomplished at three points: A programmable resistor divider at the input of the PMA allows attenuation in four 6-dB steps. The PMA has a maximum gain of 32 dB and a minimum gain of −10 dB. The VGA has a gain range of 8 dB, and the output buffer has a programmable gain control of 0 to 18 dB in 6-dB steps. The entire baseband filter lineup has a maximum gain of 64 dB and a minimum gain of −4 dB. • A notable feature of the baseband filter is the use of chopper stabilization to mitigate the undesirable effects that occur in direct-conversion receivers when designed in a CMOS process. Department of Electronic Engineering, NTUT67/82
  68. 68. Chopper Stabilization • Chopper stabilization is implemented around the first-stage amplifier of each two-stage op-amp (OPA’s input-referred voltage offset and flicker noise performance are heavily dependent on the first stage of the OPA). • The chopper frequency is derived from the crystal input and can be selected as a divide by 1, 2, 4, or 8 of the crystal frequency. Flicker noise is essentially eliminated when chopping is enabled, and thus narrowband protocols will see improvement in receiver sensitivity. Department of Electronic Engineering, NTUT68/82
  69. 69. DC Offset Correction • Dc offset correction circuitry (DCOC) is implemented as a complete control loop that corrects dc offsets automatically at the output of the baseband filter. • DCOC consists of a 1-bit ADC (comparator), control logic, and a 5-bit current-mode DAC that injects current into the feedback resistors of the VGA to adjust the offset voltage. • The control logic implements a successive approximation algorithm that converges on the correct 5-bit word that compensates for the filter’s dc offset. Department of Electronic Engineering, NTUT69/82
  70. 70. Performance of This Transceiver Summary of the Transceiver Performance Frequency range 100 MHz ~ 2.5 GHz RX NF 7 dB RX gain 48 dB RX IIP2 +60 dBm RX IIP3 -6 dBm RX current 40 mA TX output power +6 dBm TX sideband suppression 35 dBc TX current 40~90 mA EVM (pi/4 DQPSK 3.5 Msps) 1%@800 MHz LO phase noise −115 dBc/Hz@25 kHz LO frequency resolution 15 Hz LO current per DDS 80 mA Department of Electronic Engineering, NTUT70/82
  71. 71. Adaptive Multi-mode RF Front-ends • To provide various services from different wireless communication standards with high capacities and high data rates, integrated multi-functional wireless devices are required. • In the current multi-standard scenario, transceivers are mostly implemented by replicating the radio-frequency (RF) front end for each operating standard and by sharing partially the analog baseband circuitry, but with a number of additional switches. Although this approach allows for an optimal performance optimization across the bands, the increase in hardware required to implement such a multifunctional wireless device increases the total silicon area and cost and may reduce the use time compared to single-standard implementations. Department of Electronic Engineering, NTUT71/82
  72. 72. Sharing Building Blocks • By sharing building blocks between different applications and standards, portable wireless devices potentially gain advantage over existing devices: They use a smaller chip area and have a potential for lower overall cost. This requires the development of adaptive circuits and systems that are able to trade off power consumption for performance on the fly. Realization of adaptivity functions requires scaling of current consumption to the demands of the signal processing task. Department of Electronic Engineering, NTUT72/82
  73. 73. Adaptive Multi-mode Low-power Design • Mobile wireless equipment today is shaped by user and application demands and RF microelectronics. • Main drivers for mobile wireless devices are related to cost, which depends on volume of production, size of mobile units, engineering bill of materials, power consumption, and performance. Power consumption depends on available frequency spectrum, functionality, and performance. Performance depends on applications, standards, and protocols. Wireless systems for new applications require an extension of the capabilities for the RF devices of today, creating an opportunity for low-power adaptive and multifunctional RF ICs. Department of Electronic Engineering, NTUT73/82
  74. 74. Low-power and Adaptive RF Circuit • A variety of applications: Supporting the transfer of text, audio, graphic, and video data, maintaining connection with many other devices, position aware, and perhaps wearable. • A combination of multiple functional requirements and limited energy supply from a battery is an argument for the design of both adaptive low-power hardware and software. • An adaptive design approach poses unique challenges: From hardware design to application software, and ultimately throughout all layers of the underlying communication protocol. Department of Electronic Engineering, NTUT74/82
  75. 75. Adaptive Topology adaptive analog RF front-end LNA I-mixer Q-mixer Quadrature generation VCO VGA VGA A/D A/D DSP Memory CPU adaptive analog base-band adaptive digital back-end ( )x t ( )y t ( )I t ( )Q t Setting the performance parameters of an RF front end by means of adaptive circuitry is a way to manage power consumption in the RF path of a receiver. An adaptive LNA, an adaptive mixer, and an adaptive VCO allow more efficient use of scarce battery resources. Furthermore, adaptive analog baseband and digital back-end circuits enable complete hardware adaptivity (monitoring and adjust TRX parameters). Department of Electronic Engineering, NTUT75/82
  76. 76. Multi-mode and Adaptive RF Circuit (I) • GSM, UMTS, Bluetooth, 802.11a/b/g, GPS, and DVB-H are some of the standards likely to be present in the multi-standard, multi-mode, multi-band mobile terminals of the future. • The design of multi-functional wireless devices is accompanied by various challenges: System challenges: Single high-performance, low-power terminal that is cheaper than a compound of separate single-mode terminals. Circuit design challenges: For full integration of multifunctional devices include on-chip image rejection and provision of wide bandwidth and dynamic range. Technological challenges include the integration of low-cost and high- performance-scaled silicon devices on a chip. Department of Electronic Engineering, NTUT76/82
  77. 77. Multi-mode and Adaptive RF Circuit (II) • Multi-standard modules can be implemented in various ways: As stand-alone circuits that are designed for the worst-case condition of the most demanding standard. As multiple circuits (i.e., one per standard). Even though simpler to implement, this approach requires more silicon area. Moreover, when multiple standards operate simultaneously, power consumption increases. As stand-alone adaptive circuits. When different standards do not operate simultaneously, circuit blocks of a multi-mode handset can be beneficiary shared, offering power and area savings compared to other multi-standard receiver implementations, such as multi-standard receivers implemented using circuits designed for the worst-case condition and multi-standard receivers implemented with one receiver circuit per standard. Department of Electronic Engineering, NTUT77/82
  78. 78. Multi-mode and Adaptive RF Circuit (III) • For adaptive LNAs and mixers, power consumption is traded off for dynamic range, whereas adaptive oscillators trade off power consumption for phase noise and oscillation frequency. • After a signal is down-converted to the baseband, it is filtered, amplified, and digitized. To accommodate multiple radio standards with different bandwidths and modulation schemes, multi-mode LPFs need to compromise bandwidth, center frequency, selectivity, and group delay for optimal dynamic range and power consumption. Multi-mode ADCs have to sample signals belonging to different standards, tailoring different sample rates, dynamic range, and linearity requirements. Department of Electronic Engineering, NTUT78/82
  79. 79. Multi-mode Receiver Concept (I) • Wireless devices may use a common receiver if the protocols of the radio standards support intersystem operability. DCS1800 LNA LNA LNA WCDMA WLAN,DECT Bluetooth MMA-QD IC I-IF Q-IF VCO buffer buffer 2-stage Polyphase filter Differential amplifier Differential amplifier I-mixer Q-mixer Impedance matching, packaging, and prefiltering requirements are relaxed and simplified by using multiple LNAs. An RF switch selects the mode of interest. If the VCO and mixer performance is adequate to cover the range of signals anticipated for each application, the quadrature downconverter enables a multi-standard receiver realization with a single circuit block (multi-mode adaptive quadrature down- converter, MMA-QD IC). Department of Electronic Engineering, NTUT79/82
  80. 80. Multi-mode Receiver Concept (II) • Adaptivity to Requirements for Various Standards Referred to LNA Input DCS1800 WCDMA WLAN Bluetooth DECT f0 (GHz) 1.8 2.1 2.4 2.4 2.4 NF (dB) 9 6 10 23 18 IIP3 (dBm) −9 −9 −12 −16 −20 PN@1MHz (dBc/Hz) −123 −110 −110 −110 −100 Receiver Noise and Linearity Specification per Mode of Operation Specification Demanding mode DCS1800/WCDMA Moderate mode WLAN 802.11b Relaxed mode BT/DECT NF (dB) 6 10 18 IIP3 (dBm) −9 −19 −16 PN@1MHz (dBc/Hz) −123 −110 −100 Department of Electronic Engineering, NTUT80/82
  81. 81. Multi-mode Receiver Concept (III) • Referring to the channel spacing of the standards considered, zero-IF and low-IF receiver configurations may be supported. For example, the multi-mode adaptive quadrature down- converter may operate in zero-IF mode for all standards considered except the 200-kHz narrowband GSM (DCS1800 band) standard where low-IF operation would be favored (due to flicker nosie). • The standards considered are chosen to illustrate the feasibility of the adaptivity design concept for multi-mode receivers. Department of Electronic Engineering, NTUT81/82
  82. 82. Multi-mode Receiver Concept (IV) • Given the multi-mode receiver requirements and the specifications for the LNA and baseband circuitry, the noise figure and linearity performance of the quadrature down- converter can be determined for each mode of operation using the cascaded NF and IIP3 formulas. The demanding-mode performance is met with a 0-dB gain of the down-converter with an accompanying NF tuning range (NFTR) of 14.6 dB and an IIP3 tuning range (IIP3TR) of 7.6 dB. Note that by trading off the performance of the LNA and baseband circuitry, a different (more relaxed or more demanding) set of down- converter requirements results. Required Performance for the Multi-mode Quad-Downconverter for 0 dB of Gain Specification Demanding mode DCS1800/WCDMA Moderate mode WLAN 802.11b Relaxed mode BT/DECT NFqd (dB) 12.7 19.75 28.8 IIP3qd (dBm) 6.74 3.35 −0.87 Department of Electronic Engineering, NTUT82/82