The document proposes and analyzes a new circuit topology called a self-oscillating LNA-mixer. The circuit achieves high-performance by reusing the same bias current between the LNA, mixer, and local oscillator functions. It provides amplification, down-conversion, and local oscillator signal generation while sharing a single bias current, resulting in low power consumption. The document describes the evolution of the circuit topology, provides analysis of its gain, noise, linearity and phase noise characteristics, and presents simulation results demonstrating its performance targeting a Galileo satellite navigation application.

1. A Self-Oscillating LNA-Mixer
Tero Koivisto Esa Tiiliharju
University of Turku University of Turku
Microelectronics Laboratory Microelectronics Laboratory
Turku Center for Computer Science, TUCS Turku, Finland
Turku, Finland
Email: tejuko@utu.fi
Vdd Vdd
Abstract—In this paper, a new circuit topology to realize a stacked self-
oscillating LNA-Mixer is proposed. The basic idea has been to recognize
that in a high-performance down-conversion mixer its RF input-stage
gain, linearity, and noise tradeoff is often improved by feeding it with
lo1 lo2 lo1 lo1 lo2 lo1
a bypass current source. This current source could be isolated with an
inductor so as to allow free implementation of the oscillator block on
top of it. Using these guidelines, the presented circuit achieves high- rf1 rf2 rf1 rf2
performance without sacrificing compatibility with modern low-voltage
a) b)
CMOS implementations. To further demonstrate usefulness of the circuit,
an entire single-stage quadrature (IQ) RF front-end using this circuit as Vdd Vdd
a core has been developed. The IQ front-end, targeted for the Galileo
satellite navigation system, has been designed using a 65-nm CMOS VCO
technology, and it achieves NF=4.4 dB, IIP3=-15 dBm and Av =25 dB at
1.575 GHz, while using only 1 mA from the low 1.2-V supply.
lo1 lo2 lo1 lo1 lo2 lo1
I. I NTRODUCTION
The growing market for satellite navigation systems, such as rf1 rf2 rf1 rf2
Galileo and GPS, is considered the killer application for high- c) d)
performance ultra low-power transceivers. For example, it is esti-
mated that EU companies, users and society can make at least 55 Fig. 1. The mixer circuit configurations: a) Gilbert mixer with/without current
bleeding, b) Gilbert mixer with current reused bleed technique, c) Gilbert
billion euros benefits from Galileo over the next 20 years. mixer with current bleeding through differential inductor center tap and d) a
A popular technique to achieve a low-power consumption in RF current reused Gilbert mixer-oscillator.
front-end circuits is reusing the current between different functional
blocks, i.e. the same DC bias current is shared between different
circuit functionalities [1]. A well-known example of this trend is current source supplies part of the RF input stage current, the larger
cascoding the Gilbert mixer on top of the LNA circuit [2], while gm or RL can be utilized with RF and LO transistors operating
less popular is stacking voltage-controlled oscillator (VCO) [3] [4]. safely in saturation. The current source can be realized for example
Recently, an entire RF front-ends merging the functionalities of the using PMOS current sources or passive resistor. Fig. 1b shows a
LNA, mixer and oscillator in a single circuit has been proposed [1] current reused bleeding Gilbert mixer. In this topology, a PMOS
[5]. A widely used technique to realize a high performance low- transistor operates simultaneously as a bleed current source and as
voltage RF CMOS circuits, such as mixer, is current boosting [6]. a part of the RF input amplification. In this case the higher overall
The intention of this idea is to allocate a smaller bias current to the transconduntance reduces the NF [9]. Fig. 1c shows a Gilbert mixer,
switching transistors and to the load resistors than to the RF input wherein the boost current is provided via a differential inductor center
device. This relaxes the trade-offs between linearity, gain and noise tap. The inductor tunes out the parasitic capacitances present in switch
figure (NF) [7]. transistor sources at the LO frequency and lowers the flicker noise
In this paper, a new RF front-end suitable for low-voltage operation of the mixer output. In addition, the boost current source noise is
is presented. The front-end, called self-oscillating LNA-mixer, per- minimized [10] [11].
forms amplification, down-conversion and local oscillator (LO) signal The intention is to use the boost current to realize other functionali-
generation while sharing the same DC bias current between different ties demanded in the receiver. The most power-efficient approach is to
circuit blocks of the RF front-end resulting in a robust operation and use that current to provide LO signals. Fig. 1d shows a Gilbert mixer,
low-power consumption (1.2 mW). wherein we have used the boost current to realize LO signals. The
resulting circuit is a self-oscillating mixer (SOM), wherein the current
II. E VOLUTION TOWARDS THE PROPOSED CIRCUIT allocation between RF, VCO and quad transistors can be selected
The evolution towards the proposed circuit is seen from Fig. 1. A for best performance [4]. Furthermore, the RF signal present at the
CMOS Gilbert mixer is shown in Fig. 1a. In a low-supply voltages, it outputs of the mixer RF transistors is zero at the inductor center tap,
is typically composed of a grounded-source pair, which converts the since it is a virtual ground [10].
RF input voltage to current-mode signal. The current-mode signal
is then fed to the switching quad, which is driven by a large LO III. T HE S ELF -O SCILLATING LNA-M IXER AND CIRCUIT
signal. The output current of the mixer in low-noise applications is ANALYSIS
usually driven to the passive resistor load (RL ). Fig. 1 (dashed lines The proposed self-oscillating LNA-Mixer circuit is shown in Fig. 2,
and box) shows a current boosted Gilbert mixer [8]. Since boost where we have used the VCO topology presented in [12] [13]. The
978-1-4244-8971-8/10$26.00 c 2010 IEEE

2. RF differential pair is now a LNA, a transconductor designed for low B. Noise and Phase-Noise
input noise and 50 ohm input impedance. The chosen LNA topology In active current commutating CMOS mixers, switches contribute
is a widely used inductively degenerated common-source LNA [8]. flicker noise to the mixer output in two different way. In direct
The input impedance match to 50 ohm at the wanted frequency ω mechanism, flicker noise modulates the time instants of mixer switch-
is achieved when the real part of the (1) is ing whereas in indirect mechanism, it induces current in the tail
capacitance Cp , which is commutated to the output. The inductor
Ls gm
Rs = = ωt Ls (1) tunes out the Cp at the LO frequency and suppress the indirect
Cgs
mechanism causing flicker noise. The direct mechanism [8],
and the imaginary part is set to zero [14], which leads to
vn
in = 4Isw (10)
1 ST
ωo = (2)
Cgs Lg + Ls where Isw is the bias current of each switch pair, vn is the
equivalent flicker noise of the switching quad, S is the slope of the
It is seen, that in practice the impedance to 50 ohm is achieved by LO signal at the switching event and T is LO period. It is seen that
controlling the value of the Ls , whereas the Lg sets up the wanted flicker noise caused by this mechanism can be reduced using large
operating frequency by resonating out the parasitic gate capacitance. W/L ratio, reducing the switch current or increasing the slope of
the LO signal [8]. The thermal noise contribution from the VCO is
A. Gain minimized since its noise is common mode at the mixer output due
The effective transconductance of the LNA part of the proposed to boosting through the differential inductor [10]. Therefore, the total
circuit is [14] thermal noise voltage of the mixer output is approximately [16]
2 2RL Isw
Gm = gm Qin (3) Von = 8kT RL (1 + γ + γgm RL ) (11)
πA
Overall, the gain of Gilbert cell mixer is [8] where γ is the FET noise factor and A is the LO signal amplitude.
The three terms in (13) are output noise voltages due to the load
2 resistors, the mixer differential switches and the input transconductor
Av = gm RL (4)
π stage [16]. In the proposed circuit, the Isw is low and LO amplitude
Taking into account the effective transconductance Gm of the LNA is rather high leading to small switch noise contribution. Furthermore,
stage, the gain of the LNA-Mixer is assuming that the Gloss in (11) is ≈1, the total input-referred DSB
white noise spectral density is [16]
2 2 ωo (Lg + Ls )
Av = Gm RL = gm RL (5) γ
π π 2Rs DSBVin ≈ 2π 2 kT
2
(12)
gm Q2
in
However, due to current division between mixer quad and differen-
wherein the assumption is that the input transconductor dominates
tial inductor, part of the RF current flows to the differential inductor
the noise.
[15]. Fig. 3 shows a simplified small-signal model seen at the sources
As far as the oscillator is concerned, the main drawback of the
of mixer quad transistors M3 -M6 . The parasitic capacitance Cp seen
proposed circuit is that the effective supply voltage used in the design
at the source nodes A and B is resonated out with the differential
of the VCO is reduced by the Vds of the input RF transistors. The
inductor at the desired operating frequency. The impedance seen at
phase-noise of the oscillator according to heuristic Leeson formula
the source nodes of each quad transistor is 1/gm . For differential
is [17]
operation, the inductor center tap is a virtual ground and therefore
the effective impedance seen at the source nodes A and B is kT ω 2
SSSB = F (13)
1 1 2Psig Q2 Δω 2
Z= || ||Rp (6)
gm gm where Q is the loaded quality factor of the tank, Δω=2πΔf is
the angular frequency offset, and F is noise factor. It is seen that
where
in order to realize high spectral purity oscillator, it is important to
maximize Q, Psig (or Vsig ) and minimize F. The F is a noise factor
Rp = ωLQ (7) and analyzed in detail in [18]. To maximize Q in (16) the inductor
must be chosen carefully, because the Q, especially at low microwave
and Q is the quality factor of the inductor. Due to current division
frequencies (≤ 5GHz) is mainly determined from the Q of inductor.
the gain of the LNA-Mixer is reduced by to first order by a factor of
Furthermore, in the proposed circuit the VCO is capacitively loaded
gm + gm 2gm enabling a high loaded Q. Overall, the main drawbacks are the lower
Gloss = 1
= 1
(8) effective supply voltage and the presence of the complex biasing
g m + g m + Rp 2gm + Rp
arrangement around the VCO in this circuit.
To minimize the loss, the Rp should be as high as possible. This The achievable spectral purity characteristics of the proposed
is the case when the L and Q are made as large as possible, as can be circuit techniques was determined using Spectre RF simulations. The
seen from (9) [15]. Finally, the gain of the proposed self-oscillating RF transistors M1 and M2 are biased using a current mirror to carry
LNA-Mixer is 0.5 mA each, a total of 1 mA. This bias current is entirely by-passed
to the VCO through differential inductor. The LC-VCO is voltage
2 2 ωo (Lg + Ls ) biased to operate in class-C mode for optimal performance. The mixer
Av = Gm RL Gloss = gm RL Gloss (9)
π π 2Rs quad is voltage biased to carry zero current. The differential 8 nH

3. TABLE I
P HASE NOISE CONTRIBUTION OF EACH NOISE SOURCE
Noise source Contribution Rvco
Drain current 37 %
Inductor 33 %
Voltage bias [V] 20 % Rs Rs
Current bias 5%
Other 5%
1/gm L 1/gm 1/gm L 1/gm
A B
Vdd
Cp Cp
lo11 lo22
Fig. 3. The equivalent model of loading circuit seen at the source nodes of
the switching quad
M8 Vbias M9
by one phase of the LO, and the other pairs on the right by
out1 out2 its quadrature phase. The quadrature LO signal using the circuit
presented in [20]. Furthermore, the NMOS transistors are voltage
biased to operate in class-C mode. The quadrature functionality
lo1 lo2 lo1 is accomplished using linear-region transistors in series with the
M3 M4 M5 M6 gates of the LC-VCO PMOS active transistors. The coupling devices
modulate the negative gm of the switching pairs to accomplish anti-
phase injection-locking. The coupling devices do not consume voltage
Lg Lg
rf1 rf2 headroom and dissipates no power. Furthermore, to first order, the
M1 M2
devices do not contribute to phase-noise [20]. Moreover, only one
differential inductor is needed in this topology, which saves silicon
Ls Ls area.
The voltage conversion gain of the circuit is
LNA
12
Av = gm Qin RL (15)
Fig. 2. The proposed Self-Oscillating LNA-Mixer 2π
where the factor of 1 results from the fact that the LNA drives
2
two mixers in parallel. The noise performance of this topology as
inductor has a high Q of 20. The capacitance consists of a 200 μm compared to that of the pair of Gilbert mixers depends on the relative
NMOS-transistor and a 1 pF MIM-capacitor. The achieved phase- sizes of the RF transistors to the mixer quad transistors. The circuit
noise is -129 dBc/Hz at the 1 MHz offset from 1.6 GHz oscillating has a RF transconductor advantage noise advantage but a mixer quad
frequency. Table I shows the contribution of each noise source to transistor noise disadvantage [19]. Therefore, it is seen that when
the total phase-noise. It is seen that bias circuitry in this design quad transistors are made small, the noise performance of the circuit
contributes 25% of the total noise. The raw high spectral purity relative to a pair of Gilbert mixers improves.
achieved in this self-oscillating LNA-Mixer is due to the class-C
operation of the VCO, high VCO amplitude swing and the high A. Case Study: A Galileo RF Front-End
loaded Q offered by the proposed circuit. The circuit of Fig. 4 has been designed for Galileo application
using 65 nm CMOS technology with an RF option. The LNA is
C. Linearity biased with an current mirror. The bias circuit is isolated from a
The linearity of the circuit is limited in two different points. At signal path using 20 kΩ resistors. The RF transistors M1 and M2 are
the RF (input) port, it is limited by the bias Vef f = Vgs−vt of the biased at the drain current of 0.5 mA each. Then the required source
transistors M1−2 and the voltage gain in the matching circuit [8] inductance to realize 50 Ω input impedance is about 2.5 nH. The
input network includes the total parasitic parallel capacitance (300
IIP 3M1−2
IIP 3 = (14) fF) including the ESD protection diodes and bonding pad structure.
Q2in Finally, in order to series resonate the input impedance at 1.575
GHz, the external gate inductors of 30 nH are used. The Q of these
IV. T HE S ELF -O SCILLATING IQ RF F RONT-E ND external inductors is 30. The circuit is biased such that the current
The LNA in direct conversion receivers must drive two mixers through the Q-VCO is 1 mA and the current through switching quad
to produce quadrature outputs. Essentially, there are two options to is almost zero. The quad is voltage biased in the vicinity of the
accomplish this, either the use of two separate circuits presented in threshold voltage. The load resistance RL is 1 kΩ. The differential 8
Fig. 2 or using a modified version of the circuit in Fig. 2 wherein nH inductor, which isolates the Q-VCO from the other circuitry has
the LNA part of the circuit drives two Gilbert cell switching quads a Q of 20 and resonates out the parasitic capacitance at the 1.575
as shown in Fig. 4 [19]. The two pairs on the left are switched GHz. The Q-VCO consists of a 8 nH differential inductor with a Q

4. Vdd Vdd
V. C ONCLUSION
A new circuit to realize self-oscillating LNA-Mixer has been
Q+ Q- I- I+
proposed. Furthermore, using the proposed circuit it is possible
I+ I- Q+ Q- to realize an entire single-stage RF receiver. The presented circuit
achieves a high-performance with a low-power consumption. The
Vtune Vtune
future work includes the complete analysis of the presented circuit
out1 out2 out3 out4 and possibly fabrication of the circuit.
mp mp
lo1 lo2 lo1 lo3 lo4 lo3
ACKNOWLEDGMENT
s1 s2
The first author would like to thank Jenny and Antti Wihuri
foundation for financial support.
mp
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rf1 rf2
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