This document summarizes measurements and simulations of on-chip microstrip transmission lines in silicon-germanium (SiGe) technology up to 110GHz. Two transmission lines of different lengths were fabricated with and without passivation and measured on wafer. Using de-embedding techniques and a two-line method, the characteristic parameters were extracted and matched well with electromagnetic and parametric simulations up to 110GHz, validating the simulation tools. Passivation increased insertion loss due to a lowered characteristic impedance, as predicted by the models. The accurate transmission line models will enable transmission line component design for millimeter-wave applications in SiGe technology.

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On-Chip SiGe Transmission Line Measurements
and Model Verification up to 110GHz
Thomas Zwick, Member, IEEE, Youri Tretiakov, Member, IEEE, David Goren, Member, IEEE
Abstract— On-chip microstrip transmission lines have been
measured on-wafer from below 1GHz up to 110GHz. Using differ-
ent pad de-embedding techniques as well as a technique based
on two tranmission lines of different length, the characteristic
transmission line parameters have been accurately determined.
The results are compared against simulation results from an
electromagnetic full-wave solution and the parametric IBM model
which is available in the technology’s design kit.
Index Terms— Transmission line interconnect, on-wafer mea-
surements, de-embedding.
I. INTRODUCTION
THe increasing capabilities of silicon germanium (SiGe)
technology [1] have enabled both millimeter-wave
(MMW) transceivers [2] and 40-120Gb/s serializer-deserializer
(serdes) applications [3] in silicon. Due to the small wave-
length on chip, transmission lines can be used for different
functionalities other than just interconnect. For example in
MMW transceivers, transmission lines can be used as stubs,
to realize an inductor or capacitor, and in hybrid couplers [2],
while in high-speed serdes, transmission lines can be used
as distributed peaking elements to enhance clock-distribution
bandwidth [3].
In all cases very accurate models of the transmission lines
are required in the design phase. Therefore a fast parametric
model for standard transmission line types (microstrip and
coplanar in different layers and varying dimensions) has
been developed to be used in the design environment of the
technology. Details about the model and a comparison to initial
measurement results can be found in [4]. In the following, the
model will be referred to as “parametric model”. For more
unusual transmission line structures or other passive elements,
an electromagnetic (EM) simulation tool could be used to
predict the performance. Due to the layered stackup of SiGe’s
back-end-of-the-line (BEOL), the method of moments (MoM)
is a very desirable technique. Here the MoM EM simulator
IE3D from Zeland Software Inc. was used [5].
On-chip transmission lines have been manufactured to ver-
ify both the IE3D simulation tool and the parametric model us-
ing measurements. Since the transmission lines are microstrip,
transition structures to the coplanar probes are required, which
then have to be de-embedded from the measurements to
obtain the “pure” transmission line results. Here, very accurate
measurements of those microstrip transmission lines up to
This work was partially supported by the NASA.
T. Zwick is with the IBM T. J. Watson Research Center, 1101 Kitchawan
Road, Yorktown Heights, NY 10598, USA. Y. Tretiakov and D. Goren are
with IBM Microelectronics, MS863C, 1000 River Road, Essex Junction, VT
05452, USA and the IBM Haifa Research and Development Labs, MATAM,
Haifa 31905, Israel, respectively.
110GHz are presented, which for the first time allow an
accurate verification of the simulations tools and models for
on-chip SiGe transmission line characterization in the MMW
frequency range.
Section II describes the layout and cross section of the mea-
sured transmission lines while the measurement methodology
is given in Sec. III. The results are presented and discussed in
Sec. IV and conclusions are given in Sec.V.
II. TRANSMISSION LINE DESCRIPTION
Two microstrip transmission lines of 1mm and 2mm length
have been fabricated in the standard IBM SiGe BEOL. The
lines have a signal width of 5µm and a ground width of
21µm with fused silica in between. The dielectric constant
of the fused silica was determined to be 4.1 in low frequency
capacitance measurements. Both ends of the tranmission line
are connected to a coplanar pad structure with 150µm pitch for
on-wafer probing. The center signal pad was laid out as small
as possible (80µm×60µm) to minimize its capacitance but still
big enough to guarantee a reliable probe contact. In addition,
the whole pad structure has been shielded below against
the lossy silicon substrate to yield a frequency independant
capacitance and a well defined ground to enable de-embedding
up to MMW frequencies. In addition to the lines, the same
pad structures have been designed with an open and a short
instead of the transmission lines as required for one of the
applied de-embedding methods (see Sec. III). Wafers have
been manufactured with and without passivation (meaning all
layers above the top metal layer which contains the signal line)
to separately investigate the effect of the passivation layers.
III. MEASUREMENT METHODOLOGY
The major problem of accurate on-chip device measure-
ments is that no calibration standards are available on-chip
which would allow an accurate calibration beyond the probe
pads onto the chip. The only possibility is a thru-reflect-
line (TRL) calibration, where the reference impedance of
the measured S-parameters is based on the characteristic
impedance of the transmission line standard used. Since this
impedance is not yet known one has to resort to other
techniques. Based on calibration substrates (usually ceramic)
which are available together with the microwave probes a
very accurate calibration to the probe tips can be performed.
Thereby especially the TRL calibration has been proven to be
very accurate up to MMW frequencies. Here a line-reflect-
reflect-match (LRRM) calibration with automatic load induc-
tance determination [6] was used. This technique performs
very well at MMW frequencies, and can be used down to

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low frequencies with no additional effort. The problem of
de-embedding the probe pads together with the launching
structure to the actual device remains. An overview of the
most commonly used methods for de-embedding is provided
in [7]. Here two methods have been applied which both model
the launching structures by a parallel admittance (closer to
probe pad) and a series impedance. The first method is based
on separate measurements of open and short structures to find
the values for the de-embedding [8]. The second method uses
two transmission lines of different length, where one line is
twice as long as the second line [9]. In the following the two
methods are referred to as “OS de-embedding” and “2L-L de-
embedding”, respecively. From the de-embedded S-parameters
of the transmission line the propagation constant γ and the
characteristic impedance ZL as well as the distributed trans-
mission line parameters R, L, G and C can be obtained [10].
Alternatively, the propagation constant γ of a transmission
line can be directly determined from S-parameter measure-
ments of two transmission lines with identical cross section
but different lengths very accurately without the need to
de-embed any launching structures using the method given
in [11]. This procedure is also part of the TRL calibration
method. In the following this method is referred to as “two-
line method”. For transmission lines on low loss substrates
(note that transmission line substrate here is fused silica) it
has been shown in [12] that G can be neglected (G ωC)
and C can assumed to be constant over frequency. This allows
the determination of ZL from γ by
γ
ZL
≈ jωCmean , (1)
using the mean capacitance Cmean obtained from one of the
above described de-embedding techniques at lower frequencies
where these methods can be expected to be still accurate.
IV. RESULTS
Figure 1 shows G and C extracted from the 2L-L de-
embedded results of the 1mm long line on the wafer with
passivation. The artifacts around 75GHz (around 85GHz for
0 20 40 60 80 100
−50
0
50
100
150
200
G(f)
ω*Cmean
C(f)
Cmean
frequency in GHz
GinS/mandCinpF/m
Fig. 1. G and C extracted from 1mm long transmission line on wafer with
passivation using 2L-L de-embedding.
wafer without passivation) in Fig. 1 are not due to de-
embedding problems but are simply caused by the length of the
line which reaches half a wavelength at this frequency. This
effect could be avoided by measuring smaller transmission
lines which on the other hand would be more sensitive to
probe misplacements. The curves in Fig. 1 clearly show that
C is constant over frequency and G can be neglected as
described in [12] which allows determination of the character-
istic impedance from the propagation constant as described in
Sec. III. Cmean has been determined from the lower frequency
portion of C(f) in Fig. 1.
Figures 2–5 show the propagation constant γ = α+jβ and
the characteristic impedance of both cases, with and without
passivation, obtained from both de-emebedding methods and
the two-line method in comparison with the simulation results.
The very good agreement between the two de-embedding
0 20 40 60 80 100
−2
−1.5
−1
−0.5
0
α
β
frequency in GHz
αindB/mm
0
90
180
270
360
βindegrees/mm
two−line method
OS de−embedding
2L−L de−embedding
IE3D simulation
parametric model
Fig. 2. Propagation constant γ = α + jβ for an on-chip microstrip
transmission line with passivation.
0 20 40 60 80 100
0
10
20
30
40
50
60
70
magnitude of ZL
phase of ZL
frequency in GHz
magnitudeofZL
inΩ
−30
−20
−10
0
10
20
30
40
phaseofZL
indegrees
two−line method
OS de−embedding
2L−L de−embedding
IE3D simulation
parametric model
Fig. 3. Characteristic impedance ZL for an on-chip microstrip transmission
line with passivation.
techniques and the two-line method indicates the validity of
the methods, including the lumped element representation of
the pads, up to 110GHz. Both simulation tools show a very
good agreement with the measurement results except for the
insertion loss α which is slightly underestimated by IE3D and
the parametric model. This could be caused by effects which
are not covered in the simulation tools like metal roughness.
The de-embedded S-parameters also showed a very good
agreement with IE3D and the parametric model. The return
loss (S11, S22) is better than -20dB over the whole frequency

3. ZWICK 3
0 20 40 60 80 100
−2
−1.5
−1
−0.5
0
α
β
frequency in GHz
αindB/mm
0
90
180
270
360
βindegrees/mm
two−line method
OS de−embedding
2L−L de−embedding
IE3D simulation
Fig. 4. Propagation constant γ = α + jβ for an on-chip microstrip
transmission line without passivation.
0 20 40 60 80 100
0
10
20
30
40
50
60
70
magnitude of ZL
phase of ZL
frequency in GHz
magnitudeofZL
inΩ
−30
−20
−10
0
10
20
30
40
phaseofZL
indegrees
two−line method
OS de−embedding
2L−L de−embedding
IE3D simulation
Fig. 5. Characteristic impedance ZL for an on-chip microstrip transmission
line without passivation.
range in all cases (both de-embedding techniques as well
as IE3D and the parametric model). Due to the very good
match of the lines, transmission parameters (S21, S12) are
very similar to the propagation constant results showing the
same slight underestimation of insertion losses as it can be
seen in Figs. 2 and 4. The effective dielectric constant of
the lines found from the measurements was 4.1 for the wafer
with passivation and 3 for the non-passivated wafer which also
matches the simulations.
Besides the characteristic impedance of the transmission
lines which decreases from 55Ω to 46Ω, the insertion loss
of the lines noticably increases with passivation (see Figs. 2
and 4). This can directly be explained by the impedance
change. At high frequencies the condition R ωL is satisfied
which allows one to express the insertion loss by
α ≈
R
2ZL
. (2)
Since the passivation does not effect R, the ratio of the inser-
tion losses equals the ratio of the characteristic impedances
55Ω/46Ω = 1.2 which fits the differences of the results (both
measurements and IE3D simulations) for α in Figs. 2 and 4.
V. CONCLUSIONS
Using different pad de-embedding techniques as well as
the two-line method, the characteristic transmission line pa-
rameters have been determined from 110GHz measurements
and compared to simulation results from IE3D and the para-
metric model. With the optimized pad design used here,
the OS and 2L-L de-embedding methods both show very
good performance up to 110GHz. The method to obtain the
characteristic impedance from the propagation constant [12]
was also shown to be valid. Now that the transmission line
impedance is accurately known, TRL calibration standards
can be manufactured on-chip for future SiGe MMW device
characterization. IE3D and the parametric model were shown
to be reliable up to 110GHz which allows accurate modeling
of transmission lines and other passive structures in the design
process.
ACKNOWLEDGMENT
The authors wish to thank Dr. B. Sheinman and Dr. S.
Shlafman from IBM Haifa Research Labs for providing earlier
measurement results up to 40GHz, Mr. R. Groves from IBM
Microelectronics and Ms. M. Cohen from IBM Haifa Research
Labs for helping with the test site design and the staff at IBM
Microelectronics and IBM research for design automation,
fabrication support and helpful discussions. Big appreciation
also goes to Mr. J. Cacciola from Anritsu Company and Dr.
L. Hayden from Cascade Microtech, Inc. for support with the
110GHz measurement system.
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