1. The document discusses linear wire antennas, specifically infinitesimal dipoles.
2. An infinitesimal dipole is not practical but is used to represent capacitor-plate antennas and build more complex geometries.
3. Expressions are derived for the electric and magnetic fields radiated by an infinitesimal dipole. These fields are valid everywhere except at the source.
The document discusses receiver architecture and design requirements. It covers:
1. The receiver must provide high gain of 100dB while spread across RF, IF, and baseband stages to avoid instability. It must also be sensitive to weak signals down to -110dBm and reject strong adjacent channels.
2. A superheterodyne receiver is most common as it allows for sharper filters at IF to improve selectivity. Downconverting to IF also eases image filtering requirements.
3. Automatic gain control is needed to adjust the receiver gain over a wide range of input signal levels and fit them into the baseband processing range. It helps prevent compression from strong signals exceeding the 1dB compression point.
This document discusses adaptive equalization techniques used in wireless communications. It begins by describing different types of interference such as co-channel, adjacent channel, and inter-symbol interference that affect wireless transmissions. Equalization is introduced as a technique to counter inter-symbol interference by concentrating dispersed symbol energy back into its time interval. Adaptive equalization is specifically discussed as it can track time-varying mobile channel characteristics using algorithms like zero forcing, least mean squares, and recursive least squares. The key components of an adaptive equalizer including its operating modes in training and tracking are also outlined.
LEDs are of interest for fibre optics because of five inherent characteristics..
How it works?
Spectrum of an LED
Modulation of LED
LED Vs. Laser diode
disadvantages of LED
This document discusses various types of antennas and antenna arrays. It begins by describing common antenna types including helical antennas, horn antennas, and parabolic reflector antennas. It then discusses how antenna arrays work, noting that they are composed of multiple similar radiating elements whose spacing and excitation determine the array's properties. Examples of linear and 2D arrays are provided. The document also summarizes different array configurations and beamforming techniques as well as applications such as smart antennas and adaptive arrays. Key benefits of arrays like controlling radiation patterns electronically are highlighted.
Interference limits the capacity of cellular radio systems by creating bottlenecks that reduce performance. The two primary types of interference are co-channel interference, which occurs between cells using the same frequencies, and adjacent channel interference, which occurs between nearby frequency channels. Managing interference is important for cellular system design in order to minimize cross-talk and missed/blocked calls.
Wireless communication systems are impacted by fading effects that cause fluctuations in signal strength. Fading occurs due to multipath propagation which results in multiple versions of the transmitted signal reaching the receiver at different times. This can cause either flat or frequency selective fading depending on the delay spread. Modulation techniques like BPSK can be used to combat fading. Simulation of a Rayleigh fading channel, which occurs when there is no dominant signal path, showed that it significantly impacts the bit error rate of a BPSK modulated signal. Future work could explore additional modulation techniques and integrating the model into a network simulator.
The document discusses receiver architecture and design requirements. It covers:
1. The receiver must provide high gain of 100dB while spread across RF, IF, and baseband stages to avoid instability. It must also be sensitive to weak signals down to -110dBm and reject strong adjacent channels.
2. A superheterodyne receiver is most common as it allows for sharper filters at IF to improve selectivity. Downconverting to IF also eases image filtering requirements.
3. Automatic gain control is needed to adjust the receiver gain over a wide range of input signal levels and fit them into the baseband processing range. It helps prevent compression from strong signals exceeding the 1dB compression point.
This document discusses adaptive equalization techniques used in wireless communications. It begins by describing different types of interference such as co-channel, adjacent channel, and inter-symbol interference that affect wireless transmissions. Equalization is introduced as a technique to counter inter-symbol interference by concentrating dispersed symbol energy back into its time interval. Adaptive equalization is specifically discussed as it can track time-varying mobile channel characteristics using algorithms like zero forcing, least mean squares, and recursive least squares. The key components of an adaptive equalizer including its operating modes in training and tracking are also outlined.
LEDs are of interest for fibre optics because of five inherent characteristics..
How it works?
Spectrum of an LED
Modulation of LED
LED Vs. Laser diode
disadvantages of LED
This document discusses various types of antennas and antenna arrays. It begins by describing common antenna types including helical antennas, horn antennas, and parabolic reflector antennas. It then discusses how antenna arrays work, noting that they are composed of multiple similar radiating elements whose spacing and excitation determine the array's properties. Examples of linear and 2D arrays are provided. The document also summarizes different array configurations and beamforming techniques as well as applications such as smart antennas and adaptive arrays. Key benefits of arrays like controlling radiation patterns electronically are highlighted.
Interference limits the capacity of cellular radio systems by creating bottlenecks that reduce performance. The two primary types of interference are co-channel interference, which occurs between cells using the same frequencies, and adjacent channel interference, which occurs between nearby frequency channels. Managing interference is important for cellular system design in order to minimize cross-talk and missed/blocked calls.
Wireless communication systems are impacted by fading effects that cause fluctuations in signal strength. Fading occurs due to multipath propagation which results in multiple versions of the transmitted signal reaching the receiver at different times. This can cause either flat or frequency selective fading depending on the delay spread. Modulation techniques like BPSK can be used to combat fading. Simulation of a Rayleigh fading channel, which occurs when there is no dominant signal path, showed that it significantly impacts the bit error rate of a BPSK modulated signal. Future work could explore additional modulation techniques and integrating the model into a network simulator.
Smith chart:A graphical representation.amitmeghanani
The document discusses the Smith chart, which is a graphical tool used to solve transmission line problems. Some key points:
- The Smith chart was developed in 1939 and allows tedious transmission line calculations to be done graphically.
- It provides a mapping between the normalized impedance plane and the reflection coefficient plane. Circles of constant resistance and reactance are plotted, along with the reflection coefficient.
- Parameters like impedance, admittance, reflection coefficient, VSWR can all be plotted and derived from locations on the chart.
- Examples are given of using the Smith chart to determine input impedance, reflection coefficient, and stub matching of transmission lines with various termination impedances.
Lecture Notes: EEEC6440315 Communication Systems - Inter Symbol Interference...AIMST University
This document discusses inter-symbol interference (ISI) that occurs when pulses transmitted through a band-limited channel spread into adjacent time slots, and various pulse shaping techniques to eliminate ISI. It explains that rectangular pulses cause ISI in practical band-limited channels, and introduces Nyquist's criterion for zero-ISI transmission. The document also describes raised cosine pulse shaping, which is commonly used when the symbol rate is less than the Nyquist rate, and provides an example of its use in WCDMA cellular systems.
This document discusses dipole and monopole antennas. It notes that dipoles and monopoles are widely used across radio frequencies for applications like mobile communications. An infinitesimal dipole is introduced as a theoretical construct to model antennas like top-loaded designs. The document also provides an example calculation for determining the power density and radiation resistance of a 1 cm Hertzian dipole antenna operating at 100 MHz from a distance of 1 km. Key parameters for dipole antennas like their radiation patterns and the properties of half-wave dipoles are additionally summarized.
Its a good presentation on Antenna topic because every one is know that in electrical engineering antenna is a complete subject & its too much difficult subject of electrical engineering....I hope this ppt slides helpful in your future...Thanks A lot guys.......
KINDLY REGARDS
KHAWAJA SHAHBAZ IQBAL
ELECTRICAL ENGINEER
UNIVERSITY OF CENTRAL PUNJAB ,LAHORE ,PAKISTAN
+923360690272
A phase-locked loop (PLL) is an electronic circuit that compares the phase of an input reference signal with the phase of a signal derived from its output oscillator. It adjusts the oscillator frequency to keep the input and output phases matched. A PLL consists of a phase detector, low-pass filter, and voltage-controlled oscillator (VCO). It is used for synchronization, frequency synthesis, and demodulation in applications like wireless communications, radio transmitters, and signal recovery in noise.
A Gunn diode is a type of diode that uses the Gunn effect to generate microwave frequencies when a voltage above a threshold is applied. It consists of a single piece of N-type semiconductor like gallium arsenide and has a negative differential resistance region in its current-voltage characteristics that allows it to function as an oscillator. Gunn diodes are used to generate microwave signals from 10 GHz to THz and have applications in radar, sensors, and microwave transmission.
This document discusses two approaches for modeling path loss: analytical and empirical. It focuses on two specific path loss models: the log distance path loss model and the log normal shadowing path loss model. The log distance model describes path loss increasing logarithmically with distance, but does not account for environmental clutter. The log normal shadowing model adds a zero-mean Gaussian distributed random variable to the log distance model to account for random variations in path loss caused by environmental clutter. Both models can be used to estimate or predict received signal power probabilities based on distance.
Hello everyone. This is a short presentation on path loss and shadowing. I have not covered all the topics but a brief idea is given on path loss and wireless channel propagation models.
Hope you find it useful.
Thanks
This document discusses Friis transmission formula for free space path loss. It defines key terms like power density, effective aperture, and antenna gain. The Friis formula calculates received power as a function of transmitted power, transmitter and receiver gains, wavelength, and distance. It states that path loss increases with distance and is inversely proportional to the square of the distance. The document also notes some drawbacks of the Friis model and conditions for applying it in the far field region.
MicroStrip Antenna
Introduction .
Micro-Strip Antennas Types .
Micro-Strip Antennas Shapes .
Types of Substrates (Dielectric Media) .
Comparison of various types of flat profile printed antennas .
Advantages & DisAdvantages of MSAs .
Applications of MSAs .
Radiation patterns of MSAs .
How to Optimizing the Substrate Properties for Increased Bandwidth ?
Comparing the different feed techniques .
The document discusses equalization techniques used to mitigate inter-symbol interference (ISI) in digital communication systems. Equalization aims to remove ISI and noise effects from the channel. It is located at the receiver and uses techniques like linear equalizers, decision feedback equalization, and maximum likelihood sequence estimation to estimate the channel response and minimize the error between transmitted and received symbols while balancing noise. As the wireless channel changes over time, adaptive equalization is used where the equalizer periodically trains and tracks the changing channel response.
The document provides an overview of microwave engineering and rectangular waveguides. It defines microwave frequencies as ranging from 1 GHz to 300 GHz. Rectangular waveguides transmit electromagnetic waves through successive reflections from inner walls. Modes in waveguides include transverse electric (TE) and transverse magnetic (TM) modes. The document analyzes the TM and TE modes in rectangular waveguides through solving Maxwell's equations with boundary conditions. Cut-off frequencies above which modes can propagate are determined. Examples demonstrate calculating waveguide parameters and resonant frequencies of cavity resonators.
FellowBuddy.com is an innovative platform that brings students together to share notes, exam papers, study guides, project reports and presentation for upcoming exams.
We connect Students who have an understanding of course material with Students who need help.
Benefits:-
# Students can catch up on notes they missed because of an absence.
# Underachievers can find peer developed notes that break down lecture and study material in a way that they can understand
# Students can earn better grades, save time and study effectively
Our Vision & Mission – Simplifying Students Life
Our Belief – “The great breakthrough in your life comes when you realize it, that you can learn anything you need to learn; to accomplish any goal that you have set for yourself. This means there are no limits on what you can be, have or do.”
Like Us - https://www.facebook.com/FellowBuddycom
This document discusses different types of traveling wave antennas, including long wire antennas and V antennas. It provides definitions of traveling wave antennas as non-resonant antennas where standing waves do not exist along the length. Long wire antennas are classified as having a length between 1-many wavelengths. Their current distribution attenuates along the length due to losses. V antennas consist of two wire antennas arranged horizontally to form a V shape. They can be resonant or non-resonant. Rhombic antennas are formed from two connected V antennas in a diamond shape and are highly directional but require large spaces. The document provides examples of their usage and concludes with designing a rhombic antenna.
This document provides an introduction to basic antenna parameters and concepts. It discusses how antennas convert between guided waves on transmission lines and free-space electromagnetic waves. Antennas radiate energy through accelerated or decelerated charge and currents. Key concepts covered include radiation resistance, patterns, beam area, directivity, gain, apertures, and polarization. The document aims to provide foundational knowledge of antennas and their language and parameters.
This document discusses transmission line theory and analysis. It begins by explaining how power is delivered through wires at low frequencies versus through electric and magnetic fields at microwave frequencies, defining transmission lines. It then lists common types of transmission lines including two-wire, coaxial cable, waveguide, and planar lines. It analyzes the differences between analyzing circuits at low versus high frequencies. Finally, it provides details on metallic cable transmission media, including balanced vs unbalanced lines, equivalent circuits, wave propagation, losses, phasors, and characteristic impedance.
Design & Study of Microstrip Patch Antenna.The project here provides a detailed study of how to design a probe-fed Square Micro-strip Patch Antenna using HFSS, v11.0 software and study the effect of antenna dimensions Length (L), and substrate parameters relative Dielectric constant (εr), substrate thickness (t) on the Radiation parameters of Bandwidth and Beam-width.
The document discusses different types of microwave phase shifters. It describes that a phase shifter is a two-port device that provides a fixed or variable phase shift of an RF signal with minimal attenuation. It then focuses on ferrite phase shifters, which use ferrite materials to provide a variable phase shift by changing the bias field of the ferrite. The document also discusses distributed phase shifters, active vs. passive phase shifters, analog vs. digital phase shifters, and fixed vs. variable phase shifters.
Microstrip transmission lines are used extensively in microwave integrated circuits. They consist of a conducting strip separated from a ground plane by a dielectric substrate and support a quasi-TEM wave. Microstrip lines can be easily fabricated using printed circuit board technology. Their characteristic impedance depends on the strip width, thickness, distance to the ground plane, and dielectric constant of the substrate material. Microstrip lines are used for interconnecting high-speed circuits due to their uniform signal paths and ability to be fabricated automatically, though they have higher radiation losses than other transmission line types.
The radiation pattern of an antenna is defined as a mathematical function or graphical representation of the radiation properties of the antenna as a function of space coordinates. It is used to determine the antenna's radiation properties including radiation intensity, power density, and directivity.
The radiation pattern is usually plotted on a logarithmic scale using decibels (dB), which can accentuate very low values. It typically represents the magnitude of the electric or magnetic field, or the square of the magnitude (power pattern), as a function of angular space.
The main components of a radiation pattern are the major lobe containing the maximum radiation, minor lobes of lesser radiation, and side and back lobes representing undesired
Here are the key steps to test the logic gates:
1. Connect the power supply (5V) and ground to the IC.
2. Apply different combinations of logic inputs (0V, 5V) to the input pins of each gate.
3. Observe the output pin of each gate using an LED or logic probe for the expected output based on the gate's truth table.
4. Record the observed outputs and compare them to the expected outputs based on the gate's function.
5. Test all the gates in the given ICs (AND, OR, NOT, NAND, NOR) following the same procedure.
6. Note any discrepancies between observed and expected outputs.
Smith chart:A graphical representation.amitmeghanani
The document discusses the Smith chart, which is a graphical tool used to solve transmission line problems. Some key points:
- The Smith chart was developed in 1939 and allows tedious transmission line calculations to be done graphically.
- It provides a mapping between the normalized impedance plane and the reflection coefficient plane. Circles of constant resistance and reactance are plotted, along with the reflection coefficient.
- Parameters like impedance, admittance, reflection coefficient, VSWR can all be plotted and derived from locations on the chart.
- Examples are given of using the Smith chart to determine input impedance, reflection coefficient, and stub matching of transmission lines with various termination impedances.
Lecture Notes: EEEC6440315 Communication Systems - Inter Symbol Interference...AIMST University
This document discusses inter-symbol interference (ISI) that occurs when pulses transmitted through a band-limited channel spread into adjacent time slots, and various pulse shaping techniques to eliminate ISI. It explains that rectangular pulses cause ISI in practical band-limited channels, and introduces Nyquist's criterion for zero-ISI transmission. The document also describes raised cosine pulse shaping, which is commonly used when the symbol rate is less than the Nyquist rate, and provides an example of its use in WCDMA cellular systems.
This document discusses dipole and monopole antennas. It notes that dipoles and monopoles are widely used across radio frequencies for applications like mobile communications. An infinitesimal dipole is introduced as a theoretical construct to model antennas like top-loaded designs. The document also provides an example calculation for determining the power density and radiation resistance of a 1 cm Hertzian dipole antenna operating at 100 MHz from a distance of 1 km. Key parameters for dipole antennas like their radiation patterns and the properties of half-wave dipoles are additionally summarized.
Its a good presentation on Antenna topic because every one is know that in electrical engineering antenna is a complete subject & its too much difficult subject of electrical engineering....I hope this ppt slides helpful in your future...Thanks A lot guys.......
KINDLY REGARDS
KHAWAJA SHAHBAZ IQBAL
ELECTRICAL ENGINEER
UNIVERSITY OF CENTRAL PUNJAB ,LAHORE ,PAKISTAN
+923360690272
A phase-locked loop (PLL) is an electronic circuit that compares the phase of an input reference signal with the phase of a signal derived from its output oscillator. It adjusts the oscillator frequency to keep the input and output phases matched. A PLL consists of a phase detector, low-pass filter, and voltage-controlled oscillator (VCO). It is used for synchronization, frequency synthesis, and demodulation in applications like wireless communications, radio transmitters, and signal recovery in noise.
A Gunn diode is a type of diode that uses the Gunn effect to generate microwave frequencies when a voltage above a threshold is applied. It consists of a single piece of N-type semiconductor like gallium arsenide and has a negative differential resistance region in its current-voltage characteristics that allows it to function as an oscillator. Gunn diodes are used to generate microwave signals from 10 GHz to THz and have applications in radar, sensors, and microwave transmission.
This document discusses two approaches for modeling path loss: analytical and empirical. It focuses on two specific path loss models: the log distance path loss model and the log normal shadowing path loss model. The log distance model describes path loss increasing logarithmically with distance, but does not account for environmental clutter. The log normal shadowing model adds a zero-mean Gaussian distributed random variable to the log distance model to account for random variations in path loss caused by environmental clutter. Both models can be used to estimate or predict received signal power probabilities based on distance.
Hello everyone. This is a short presentation on path loss and shadowing. I have not covered all the topics but a brief idea is given on path loss and wireless channel propagation models.
Hope you find it useful.
Thanks
This document discusses Friis transmission formula for free space path loss. It defines key terms like power density, effective aperture, and antenna gain. The Friis formula calculates received power as a function of transmitted power, transmitter and receiver gains, wavelength, and distance. It states that path loss increases with distance and is inversely proportional to the square of the distance. The document also notes some drawbacks of the Friis model and conditions for applying it in the far field region.
MicroStrip Antenna
Introduction .
Micro-Strip Antennas Types .
Micro-Strip Antennas Shapes .
Types of Substrates (Dielectric Media) .
Comparison of various types of flat profile printed antennas .
Advantages & DisAdvantages of MSAs .
Applications of MSAs .
Radiation patterns of MSAs .
How to Optimizing the Substrate Properties for Increased Bandwidth ?
Comparing the different feed techniques .
The document discusses equalization techniques used to mitigate inter-symbol interference (ISI) in digital communication systems. Equalization aims to remove ISI and noise effects from the channel. It is located at the receiver and uses techniques like linear equalizers, decision feedback equalization, and maximum likelihood sequence estimation to estimate the channel response and minimize the error between transmitted and received symbols while balancing noise. As the wireless channel changes over time, adaptive equalization is used where the equalizer periodically trains and tracks the changing channel response.
The document provides an overview of microwave engineering and rectangular waveguides. It defines microwave frequencies as ranging from 1 GHz to 300 GHz. Rectangular waveguides transmit electromagnetic waves through successive reflections from inner walls. Modes in waveguides include transverse electric (TE) and transverse magnetic (TM) modes. The document analyzes the TM and TE modes in rectangular waveguides through solving Maxwell's equations with boundary conditions. Cut-off frequencies above which modes can propagate are determined. Examples demonstrate calculating waveguide parameters and resonant frequencies of cavity resonators.
FellowBuddy.com is an innovative platform that brings students together to share notes, exam papers, study guides, project reports and presentation for upcoming exams.
We connect Students who have an understanding of course material with Students who need help.
Benefits:-
# Students can catch up on notes they missed because of an absence.
# Underachievers can find peer developed notes that break down lecture and study material in a way that they can understand
# Students can earn better grades, save time and study effectively
Our Vision & Mission – Simplifying Students Life
Our Belief – “The great breakthrough in your life comes when you realize it, that you can learn anything you need to learn; to accomplish any goal that you have set for yourself. This means there are no limits on what you can be, have or do.”
Like Us - https://www.facebook.com/FellowBuddycom
This document discusses different types of traveling wave antennas, including long wire antennas and V antennas. It provides definitions of traveling wave antennas as non-resonant antennas where standing waves do not exist along the length. Long wire antennas are classified as having a length between 1-many wavelengths. Their current distribution attenuates along the length due to losses. V antennas consist of two wire antennas arranged horizontally to form a V shape. They can be resonant or non-resonant. Rhombic antennas are formed from two connected V antennas in a diamond shape and are highly directional but require large spaces. The document provides examples of their usage and concludes with designing a rhombic antenna.
This document provides an introduction to basic antenna parameters and concepts. It discusses how antennas convert between guided waves on transmission lines and free-space electromagnetic waves. Antennas radiate energy through accelerated or decelerated charge and currents. Key concepts covered include radiation resistance, patterns, beam area, directivity, gain, apertures, and polarization. The document aims to provide foundational knowledge of antennas and their language and parameters.
This document discusses transmission line theory and analysis. It begins by explaining how power is delivered through wires at low frequencies versus through electric and magnetic fields at microwave frequencies, defining transmission lines. It then lists common types of transmission lines including two-wire, coaxial cable, waveguide, and planar lines. It analyzes the differences between analyzing circuits at low versus high frequencies. Finally, it provides details on metallic cable transmission media, including balanced vs unbalanced lines, equivalent circuits, wave propagation, losses, phasors, and characteristic impedance.
Design & Study of Microstrip Patch Antenna.The project here provides a detailed study of how to design a probe-fed Square Micro-strip Patch Antenna using HFSS, v11.0 software and study the effect of antenna dimensions Length (L), and substrate parameters relative Dielectric constant (εr), substrate thickness (t) on the Radiation parameters of Bandwidth and Beam-width.
The document discusses different types of microwave phase shifters. It describes that a phase shifter is a two-port device that provides a fixed or variable phase shift of an RF signal with minimal attenuation. It then focuses on ferrite phase shifters, which use ferrite materials to provide a variable phase shift by changing the bias field of the ferrite. The document also discusses distributed phase shifters, active vs. passive phase shifters, analog vs. digital phase shifters, and fixed vs. variable phase shifters.
Microstrip transmission lines are used extensively in microwave integrated circuits. They consist of a conducting strip separated from a ground plane by a dielectric substrate and support a quasi-TEM wave. Microstrip lines can be easily fabricated using printed circuit board technology. Their characteristic impedance depends on the strip width, thickness, distance to the ground plane, and dielectric constant of the substrate material. Microstrip lines are used for interconnecting high-speed circuits due to their uniform signal paths and ability to be fabricated automatically, though they have higher radiation losses than other transmission line types.
The radiation pattern of an antenna is defined as a mathematical function or graphical representation of the radiation properties of the antenna as a function of space coordinates. It is used to determine the antenna's radiation properties including radiation intensity, power density, and directivity.
The radiation pattern is usually plotted on a logarithmic scale using decibels (dB), which can accentuate very low values. It typically represents the magnitude of the electric or magnetic field, or the square of the magnitude (power pattern), as a function of angular space.
The main components of a radiation pattern are the major lobe containing the maximum radiation, minor lobes of lesser radiation, and side and back lobes representing undesired
Here are the key steps to test the logic gates:
1. Connect the power supply (5V) and ground to the IC.
2. Apply different combinations of logic inputs (0V, 5V) to the input pins of each gate.
3. Observe the output pin of each gate using an LED or logic probe for the expected output based on the gate's truth table.
4. Record the observed outputs and compare them to the expected outputs based on the gate's function.
5. Test all the gates in the given ICs (AND, OR, NOT, NAND, NOR) following the same procedure.
6. Note any discrepancies between observed and expected outputs.
These lecture notes cover microwave engineering topics such as transmission line analysis, microwave networks, impedance matching, power dividers and couplers, noise and active components, and microwave amplifier design. The notes are based on the textbook Microwave Engineering by David M. Pozar and contain 7 main sections that describe key microwave engineering concepts and analysis methods. Contact information is provided for the author, Dr. Serkan Aksoy, for future versions or proposals related to the material.
This document describes the design and simulation of a dual band patch antenna for WLAN/WiMax applications. Three miniaturized dual band u-slot patch antenna designs are investigated that can operate in both the 2.4 GHz and 5 GHz bands. The second resonant frequency band can be shifted by varying the width of the u-slot. The antennas are simulated using Ansoft HFSS and Antenna Magus software. The antenna parameters such as input impedance, return loss, and polarization are obtained and optimized to meet design requirements.
This document discusses the design of high performance printed circuit boards. It covers topics such as PCB design flow, mechanical design considerations, electrical considerations like impedance and noise, capacitor advantages and disadvantages, microstrip and signal trace configurations, cross-talk control techniques, termination methods, and guidelines for layout and EMI reduction. The goal is to minimize delays, noise, and reliability issues through careful mechanical and electrical design.
Iwashita - Laminated conductor structure for rf in normal conducting casethinfilmsworkshop
http://www.surfacetreatments.it/thinfilms
Laminated Conductor Structure for RF in normal conducting case (Yoshihisa Iwashita - 20')
Speaker: Yoshihisa Iwashita - Kyoto University | Duration: 20 min.
Abstract
Laminated conductor structure for RF was proposed by A.M. Clogston in 1951.The motivation was to reduce the skin-effect loss caused by Joule heating.When the currents are well distributed to conductor foils that is thinner than the skin depth, the current density can be reduced and the power dissipation in the conductor can be reduced.This structure, however, has not been practically used, maybe because of some restrictions to apply. When we apply the similar layered structure for superconducting (sc.) surfaces, similar restriction may hit.Some thoughts that may be useful for sc. from the study on normal conducting case will be discussed.
This document is a thesis written in German that discusses active switching in long distance quantum state teleportation. It begins by providing background on principles of quantum information theory, including elements of classical information theory like Shannon entropy and noisy channel coding theorem. It then discusses quantum entanglement and nonlocality, including entangled states, the EPR paradox, Bell's inequalities, and loopholes. The thesis goes on to describe quantum state teleportation and Bennett's scheme. It discusses the general setup, components, and implementation of long distance quantum state teleportation, including Alice and Bob's laboratories, the classical and quantum channels, and hardware for actively triggered unitary transformations using Pockels cells.
This document discusses coherence and optical fibers. It defines temporal and spatial coherence, which refer to the ability of waves to interfere with themselves or other waves at different times or positions, respectively. Coherence is necessary for interference. Optical fibers transmit light via total internal reflection within a core surrounded by cladding. Fibers can be single-mode or multi-mode depending on the number of propagation modes supported. Numerical aperture specifies the range of angles at which light can enter and propagate within the fiber.
This thesis examines weak localization effects in disordered graphene. The document outlines the fabrication process and experimental setup used. Chapter 1 provides background on graphene's band structure, transport properties, and weak localization effects. Chapter 2 describes the device fabrication process, including cleaning the silicon substrate, exfoliating graphene, identifying samples with Raman spectroscopy, electron beam lithography, metal deposition, and lift-off. Electrical characterization of the fabricated devices is discussed in Chapter 4, focusing on measurements of conductivity, mobility, temperature dependence, and magnetic field effects. The goal is to use weak localization to characterize charge density fluctuations in graphene resulting from defects and trapped charges.
Wireless Mobile Charging Using MicrowavesJishid Km
It is a hectic task to carry everywhere the charger of mobile phones or any electronic gadget while travelling, or it is very cruel when your mobile phone getting off by the time you urgently need it. It is the major problem in today’s electronic gadgets. Though the world is leading with the developments in technology, but this technology is still incomplete because of these limitations. Today’s world requires the complete technology and for this purpose here we are proposing the wireless charging of batteries using Microwaves.
Now in the recent days we come across some solutions for this problem by using the Witricity (Wireless Transmission of Electricity). Recently Nokia has launched Nokia Lumia 920 smart phones whose special feature is its wireless charging. But this is possible only when the device is placed on to the plate given for the wireless charging. So it is also somewhat difficult to travel with those charging plates. There may chance has forgetting the charging plates, and then we require something which can charge our electronic gadgets whenever they get used
The proposed method gives the solution for this problem. Once think that how it will be when your electronic gadget gets charged on using it? Then the label will come as “CHARGE ON USE”. This wireless charging method works on the principle of MICROWAVE OVEN. As the things when placed in microwave oven gets heated, in the same way these batteries should work using microwaves which are the medium of communication from long back. We are getting our network in terms of microwaves and it is proved that the total radiation coming from the cellular mobile communication is not been using and the remaining radiation is creating hazardous problem for human beings. So here we are working on the concept that why can’t we use those remaining radiations in order to charge our batteries? This will be the best solution to reduce the effect of radiation.
This document contains lecture notes on microwave engineering. It covers topics such as electromagnetic theory, transmission line theory, transmission lines and waveguides, microwave network analysis, impedance matching and tuning, power dividers and directional couplers, and electromagnetic compatibility and interference. Some key concepts discussed include Maxwell's equations, time-harmonic fields, plane wave propagation, transmission line models, Smith charts, impedance matching techniques, and microwave network parameters.
This document is Francesco Volpe's 2003 PhD thesis from Ernst Moritz Arndt University in Greifswald, Germany. The thesis presents a novel diagnostic technique for measuring electron temperature profiles in the W7-AS stellarator plasma above the electron cyclotron emission cutoff density. The technique uses electron Bernstein waves, which are confined within the upper hybrid layers and converted to ordinary mode waves that propagate out of the plasma. An antenna and transmission line were designed and optimized using ray tracing calculations to maximize the conversion efficiency. Experimental results demonstrated measurements of electron temperature profiles, edge localized modes, confinement transitions, and radiative collapses using this diagnostic up to densities of 3.8×1020 m-3. The technique was also
This document is a project report submitted to fulfill the requirements for a Bachelor of Science degree in Physics. It provides a PowerPoint presentation on solid state devices. The presentation introduces solid state devices and covers topics such as P and N type materials, PN junctions, diodes and their types including Zener diodes, transistors and their types, integrated circuits, and the advantages of solid state devices. It describes how silicon and germanium are used to make diodes and how dopants create P and N type materials. It also explains how PN junctions work and the differences between forward and reverse bias. Different types of diodes like LEDs and photodiodes are outlined along with their uses. The presentation concludes by
Design methodology for undersea umbilical cablesthinknice
This document describes a methodology for designing umbilical cables used in undersea applications. It presents a design nomograph and simple formulas as preliminary design aids. The methodology involves specifying operational parameters, conducting a preliminary design, defining the cable cross-section, analyzing the cable suspension, using computer-aided design, and prototype testing. Key steps include determining armor area, helical wire coverage, torque and stress balance, and using linear cable equilibrium equations. The goal is to satisfy strength, deformation, and handling requirements through an iterative design process.
This document provides a seminar report on Cellonics technology, which allows for modem speeds 1,000 times faster than current technologies. It discusses the principles of Cellonics, which are inspired by how biological cells communicate through nonlinear dynamical systems. Cellonics circuits mimic this behavior to encode and transmit digital data. Applications to telecommunications are explored, demonstrating carrier-rate decoding that transmits one symbol per radio frequency cycle, vastly increasing data rates. Proof-of-concept systems are presented, including wired and wireless narrowband communication systems transmitting at 5.7 Mbps and 26.7 Mbps respectively, showcasing the speed and robustness of Cellonics technology.
The document describes an electronics lab course that focuses on antenna design and implementation of digital communication systems using MATLAB/Simulink. The course covers topics like using a slotted line to determine unknown frequency and reflection coefficients, investigating yagi antennas by examining driven elements, reflectors and directors, and studying the effects of conductor thickness on bandwidth. It also includes implementing digital modulation and coding techniques like PCM, PSK and FSK using MATLAB. Students use antenna hardware and software to study properties of dipole antennas and parasitic elements in yagi antenna systems.
The document describes an electronics lab course that focuses on antenna design and implementation of digital communication systems using MATLAB/Simulink. The course covers topics like using a slotted line to determine unknown frequency and reflection coefficients, investigating yagi antennas by examining driven elements and parasitic directors and reflectors, and implementing modulation/demodulation techniques like PCM, PPM, and various digital coding formats in software simulations. Students will explore how antenna performance is affected by parameters like element thickness and stacking/baying configurations. The course aims to help students understand fundamental antenna properties and simulate key communication systems.
This document provides an overview of X-ray fluorescence (XRF) spectroscopy. It discusses XRF theory, instrumentation, hardware, and applications. XRF uses X-rays to excite a sample, and a detector then measures the fluorescent X-rays emitted from the sample that are characteristic of its elemental composition. The document compares wavelength dispersive XRF and energy dispersive XRF, and describes the components of XRF systems including X-ray sources, detectors, filters, and electronics. It provides examples of XRF applications in qualitative and quantitative elemental analysis across various industries.
1. CHAPTER4 Linear Wire Antennas
4.1 INTRODUCTION .............................................................................................................................................................................................................................. 2
.
4.2 INFINITESIMAL DIPOLE ................................................................................................................................................................................................................... 2
4.2.1 Radiated Fields ..................................................................................................................................................................................................................... 3
4.2.2 Power Density and Radiation Resistance ............................................................................................................................................................................ 7
4.2.3 Near‐Field ( ) Region .............................................................................................................................................................................................. 13
4.2.5 Intermediate‐Field (kr > 1) Region ..................................................................................................................................................................................... 15
4.2.6 Far‐Field (kr >> 1) Region ................................................................................................................................................................................................... 17
4.2.7 Directivity ........................................................................................................................................................................................................................... 19
4.3 SMALL DIPOLE ....................................................................................................................................................................................................................... 21
4.4 REGION SEPARATION .............................................................................................................................................................................................................. 25
4.4.1 Far‐Field (Fraunhofer) Region ............................................................................................................................................................................................ 27
4.4.2 Radiating Near‐Field (Fresnel) Region ............................................................................................................................................................................... 30
4.4.3 Reactive Near‐Field Region ................................................................................................................................................................................................ 32
4.5 FINITE LENGTH DIPOLE ................................................................................................................................................................................................................. 33
4.5.1 Current Distribution ........................................................................................................................................................................................................... 33
4.5.2 Radiated Fields: Element Factor, Space Factor, and Pattern Multiplication ..................................................................................................................... 35
4.5.3 Power Density, Radiation Intensity, and Radiation Resistance ......................................................................................................................................... 37
4.5.4 Directivity ........................................................................................................................................................................................................................... 41
4.5.5 Input Resistance ................................................................................................................................................................................................................ 42
.
4.6 HALF‐WAVELENGTH DIPOLE ......................................................................................................................................................................................................... 45
4.7 LINEAR ELEMENTS NEAR OR ON INFINITE PERFECT CONDUCTORS ............................................................................................................................................. 49
4.7.1 Image Theory ..................................................................................................................................................................................................................... 50
4.7.2 Vertical Electric Dipole ....................................................................................................................................................................................................... 53
1. Radiation pattern ................................................................................................................................................................................................................ 54
2. Radiation power and directivity ......................................................................................................................................................................................... 57
.
3. monopole ............................................................................................................................................................................................................................ 61
4.7.4 Antennas for Mobile Communication Systems ................................................................................................................................................................. 63
4.7.5 Horizontal Electric Dipole .................................................................................................................................................................................................. 67
PROBLEMS .......................................................................................................................................................................................................................................... 74
2.
4.1
1 INTROD
DUCTION
Wire antennas,
a , linear or curved, are some e of the o
oldest, sim
mplest, cheapest,
an
nd the mo ile for many applica
ost versati ations.
4.2
2 INFINITESIMAL D
DIPOLE
Infinitesimal dipol
les are no
ot practica
al, they ar
re used to
o represen
nt capacit
tor‐plate
antennas
s.
In additio
on, they a
are utilized as build
ding more
e complex
x geometr
ries.
The end pl lates are used to
e
provide c e loading to mainta
capacitive ain
the current on the dippole neaarly
uniform.
3. The plates are very small, their radiation is usually negligible. The wire, in
addition to being very small (l <<), is very thin ( ). The spatial variation of
the current is assumed to be constant
′ ; = constant (4‐1)
4.2.1 Radiated Fields
To find the fields radiated by the current element, it will be required to
determine first and and then find the and .
1. Calculation of
Since the source only carries an electric current , therefore and the
potential function are zero. To find we write
, , ′ ′ (4‐2)
x, y, z : the observation point ; x’, y’, z’ : the source coordinates
: the distance from any point on the source to the observation point
path C : is along the length of the source
4.
Fo
or the problem of F
Figure 4.1
, , 4‐3
0 (infinite
esimal dip
pole)
′
so
o we can w
write (4‐2) as
/
, , /
(4‐4)
5. 2. Calculation of and
To calculate and , it is simpler to transform (4‐4) from rectangular to
spherical components.
(4‐5)
0
For this problem, 0, so (4‐5) using (4‐4) reduces to
(4‐6)
0
⟹ (4‐7)
Substituting (4‐6) into (4‐7) reduces it to
0
(4‐8)
1
6.
Th
he electric
c field E ca
an now be
e found. T
That is,
∙ (4‐9)
1
1 (4‐10)
0
The and ‐field components
are valid ev
e verywher except on the
re, t
so
ource itself, and th are sketched
hey s
in Figure 4
4.1(b) on the surfa of a
ace
sp
phere of ra adius .
7.
4.2.2 Power Density and Radiation Resistance
The input impedance of an antenna consists of real and imaginary parts. For a
lossless antenna, the real part of the input impedance was radiation resistance.
To find the input resistance for a lossless antenna, the following procedure is
taken.
For the infinitesimal dipole, the complex Poynting vector can be written using
(4‐8a)–(4‐8b) and (4‐10a)–(4‐10c) as
1 ∗
1 ∗
2 2
∗ ∗
(4‐11)
8. 1
⟹ (4‐12)
| |
1
Since is imaginary, it will not contribute to real radiated power. The
reactive power density, which is most dominant for small values of , has both
radial and transverse components. It merely changes between outward and inward
directions to form a standing wave at a rate of twice per cycle. It also moves in the
transverse direction.
The complex power moving in the radial direction is obtained by integrating
(4‐11)–(4‐12b) over a closed sphere of radius r. Thus it can be written as
∯ ∙ ∙ 4‐13
⟹ 1 (4‐14)
Equation (4‐13), which gives the real and imaginary power that is moving
outwardly, can also be written as
9. ∗
∙ 1 P j2ω W W (4‐15)
Where: P power in radial direction ; Prad time‐average power radiated
W time‐average magnetic energy density in radial direction
W time‐average electric energy density in radial direction
2 W W time‐average imaginary reactive power
From (4‐14)
P ; 2ω W W (4‐16, 17)
It is clear from (4‐17) that When kr ∞, the reactive power diminishes
and vanishes.
10.
1. radiation resistance of the infinitesimal dipole
Since the antenna radiates its real power through the radiation resistance, for
the infinitesimal dipole it is found by equating (4‐16) to
| | ⇒ 80 (4‐18, 19)
For a wire antenna to be classified as an infinitesimal dipole, its overall length
must be very small (usually ).
11.
Example 4.1
Find the radiation resistance of an infinitesimal dipole whose overall length is
/50.
Solution:
Using (4‐19)
1
80 80 0.316
50
Since the radiation resistance of an infinitesimal dipole is about 0.3 ohms, it
will present a very large mismatch when connected to practical transmission lines,
many of which have characteristic impedances of 50 or 75 ohms. The reflection
efficiency ( ) and hence the overall efficiency ( ) will be very small.
12.
2. The reactance of an infinitesimal dipole is capacitive.
This can be illustrated by considering the dipole as a flared open‐circuited
transmission line. Since the input impedance of an open‐circuited transmission line
a distance from its open end is given by
2
where is its characteristic impedance, it will always be negative (capacitive) for
≪ .
13.
4.2.3 Near‐Field ( ) Region
An inspection of (4‐8) and (4‐10) reveals that for / 2 they
can be approximated by
(4‐8a, 10c) (4‐20c)
(4‐8b) (4‐20d)
(4‐10a) (4‐20a)
(4‐10b) (4‐20b)
The E‐field components, and are in time‐phase;
They are in time‐phase quadrature with the H‐field component ;
Therefore there is no time‐average power flow associated with them. This is
demonstrated by forming the time‐average power density as
∗ ∗
W Re E H∗ Re
14. | |
⟹W Re 0 (4‐22)
Equations (4‐20a) and (4‐20b) are similar to those of a static electric dipole and
(4‐20d) to that of a static current element. Thus we usually refer to (4‐20a)–(4‐20d)
as the quasi‐stationary fields.
15.
4.2.5 Intermediate‐Field (kr > 1) Region
As the values of begin to increase and become greater than unity, the
terms that were dominant for ≪ 1 become smaller and eventually vanish.
1 (4‐8b) (4‐23d)
1 (4‐10a) (4‐23a)
1 (4‐10b) (4‐23b)
For moderate values of :
The E‐field components lose their in‐phase condition and approach
time‐phase quadrature.
Their magnitude is not the same, they form a rotating vector whose
extremity traces an ellipse. This is analogous to the polarization problem
except that the vector rotates in a plane parallel to the direction of
propagation and is usually referred to as the cross field.
16. At these intermediate values of , the and components
approach time‐phase, which is an indication of the formation of
time‐average power flow in the outward direction.
(4‐8a, 10c) (4‐23c)
(4‐8b) (4‐23d)
(4‐10a) (4‐23a)
(4‐10b) (4‐23b)
The total electric field is given by
(4‐24)
17.
4.2.6 Far‐Field (kr >> 1) Region
In a region where ≫ 1 , (4‐23a) – (4‐23d) can be simplified and
approximated by
(4‐8a, 10c) (4‐26b)
(4‐8b) (4‐26c)
(4‐10a) (4‐26b)
(4‐10b) (4‐26a)
The ratio of to is equal to
Z (4‐27)
The E‐ and H‐ field components are perpendicular to each other, transverse to
the radial direction of propagation. The fields form a Transverse ElectroMagnetic
(TEM) wave,its wave impedance is the intrinsic impedance of the medium.
18.
Example 4.2
For an infinitesimal dipole determine and interpret the vector effective length.
At what incidence angle does the open‐circuit maximum voltage occurs at the
output terminals of the dipole if the electric‐field intensity of the incident wave is
10 mV/m? The length of the dipole is 10 cm.
Solution:
Using (4‐26a) and the effective length as defined by (2‐92), we can write that
4 26a
⟹
2 92
The maximum value occurs at 90 and it is equal to . The open‐circuit
maximum voltage is equal to
| ∙ | 10 10 ∙ | 10 volts
19.
4.2.7 Directivity
The real power P radiated by the dipole was found in Section 4.2.2, as
given by (4‐16). The same expression can be obtained by first forming the average
power density, using (4‐26a)–(4‐26c). That is,
∗
Re | | (4‐28)
Integrating (4‐28) over a closed sphere of radius r reduces it to (4‐16).
Associated with the average power density of (4‐28) is a radiation intensity U which
is given by
| | ⟹ (4‐29, 30)
Using (4‐16) and (4‐30), the directivity reduces to
4 (4‐31)
and the maximum effective aperture to
(4‐32)
20.
21.
4.3 SMALL DIPOLE
3 E
The creation of th he current distribution on a
a thin wiree was disscussed in
n Section
1.4, and it was illu
ustrated w
with somee examplees in Figure 1.16.
The radiaation propperties of
f an infinit
tesimal di
ipole were
e discusseed in the previous
section. I
Its curren
nt distribution was assumed to be con nstant.
A consttant curre
ent distrib
bution is n
not realizable. A be
etter approximatioon of the
cu
urrent disttribution o ntennas, (/50
of wire an ( /10) is the tr
riangular v
variation,
whhich is sho
own in Fig
gure 4.4(bb)
1 , 0
, ,
1 , 0
(4
4‐33)
22.
Th
he vector potential can be w
written using (4‐33) as
/
/
1 1 (4‐34)
Becausse the lenngth of th
he dipole is very sm
mall /10 , for diff
ferent ’
alo wire are not much different from . T
ong the w Thus c
can be ap
pproximatted by
throughoutt the integ
gration pa
ath.
The maximum phase error in (4 4‐34) by allowing will be /2
g
/
/10 18 ffor /10. Thiss amount of phase error hass very litt
tle effect
on rall radiation characteristics. Then, (4
n the over 4‐34) redu
uces to (4‐
‐35)
23. (4‐35)
which is one‐half of that for the infinitesimal dipole.
/
Ref: , , /
(4‐4)
The potential function (4‐35) becomes a more accurate approximation as kr → ∞.
Since the potential function for the triangular distribution is one‐half of the
corresponding one for the constant (uniform) current distribution, the
corresponding fields of the former are one‐half of the latter. Thus we can write the
E‐ and H‐fields radiated by a small dipole as
(4‐26b) (4‐36b)
(4‐26a) (4‐36a)
(4‐26c) (4‐36c)
24. Since the directivity of an antenna is controlled by the relative shape of the
field or power pattern, the directivity, and maximum effective area of this antenna
are the same as the ones with the constant current distribution given by (4‐31) and
(4‐32), respectively.
Using the procedure established for the infinitesimal dipole, the radiation
resistance for the small dipole is
80 (4‐18) | |
20 (4‐37)
The small dipole its radiated power is of (4‐18). Thus the radiation
resistance of the antenna is strongly dependent upon the current distribution.
25.
4.4 REGION SEPARATION
Before solving the fields radiated by a finite dipole of any length, it is desirable
to discuss the separation of the space surrounding an antenna into three regions
The reactive near‐field
The radiating near‐field
The far‐field
To solve for the fields efficiently, approximations can be made to simplify the
formulation. The difficulties in obtaining closed form solutions that are valid
everywhere for any practical antenna stem from the inability to perform the
integration of
, , ′ ′ (4‐2, 38)
where
(4‐38a)
In the calculations for infinitesimal dipole and small dipole. The major
simplification of (4‐38) will be in the approximation of R.
26. The Fig
gure showws a very tthin dipol
le of finite
e length l
l symmetrically pos
sitioned.
Be
ecause the
e wire is v (x’ y’ 0), we
very thin ( e can writte (4‐38) a
as
(4‐39)
wh
hich can b
be written
n as
2
2 ′ (4‐40
0)
Us
sing the binomial e
expansion, we can w
write (4‐4
40) in a se
eries as
⋯ (4‐41)
wh
hose higher order t
terms bec
come less
s significan
nt provide
ed r >> z’.
.
27.
4.4.1 Far‐Field (Fraunhofer) Region
The most convenient simplification of (4‐41) is to approximate it by
≃ ′ (4‐42)
To maintain the maximum phase error of an antenna equal to or less than /8
rad (22.5 ), the observation distance r must equal or be greater than 2 /.
2 / (4‐45)
The usual simplification for the far‐field region is
≃ for phase terms
(4‐46)
≃ for amplitude terms 1/
Ref: , , ′ ′ (4‐38)
For any other antenna whose maximum dimension is , the approximation of
(4‐46) is valid provided
r 2D /λ (4‐47)
For an aperture antenna the maximum dimension is taken to be its diagonal.
28.
It wou
uld seem that thee approxiimation o R in (4‐46) fo
of or the am
mplitude
is more sev
vere than that fo
or the pha
ase.
Ex
xample 44.3
For an antenn with an overall lengt
n na th 5, the o
observations are
made at 60. Find the
e errors in phase and ampplitude ussing (4‐4
46).
So
olution:
For 90 , z’
, 2
2.5, and r
d 6
60, (4‐40
0) reduce
es to
29. 2 2 ′ (4‐40)
60 2.5 60.052
≃ for phase terms
With (4‐46)
≃ for amplitude terms 1/
r 60
Therefore the phase difference is
2
∆ ∆ 0.327 18.74 22.5
The difference of the inverse values of R is
1 1 1 1 1 1.44 10
60 60.052
which should always be a very small value in amplitude.
30.
4.4.2 Radiating Near‐Field (Fresnel) Region
If the observation point is chosen to be smaller than 2 / , the maximum
phase error by the approximation of (4‐46) is greater than /8 rad (22.5o).
≃ for phase terms
(4‐46)
≃ for amplitude terms 1/
If it is necessary to choose observation distances smaller than 2 / , another
term (the third) in the series solution of (4‐41) must be retained to maintain a
maximum phase error of /8 rad (22.5o).
⋯ (4‐41)
Doing this, the infinite series of (4‐41) can be approximated by
(4‐48)
A value of greater than that of (4‐52a) will lead to an error less than /8 rad
(22.5o).
31. 0.385 or 0.62 / (4‐52, 4‐52a)
√ √
The region where the first three terms of (4‐41) are significant, and the
omission of the fourth introduces a maximum phase error of /8 rad (22.5o), is
defined by
/
2 0.62 / (4‐53)
This region is designated as radiating near field because
The radiating power density is greater than the reactive power density
The field pattern is a function of the radial distance r.
This region is also called the Fresnel region because the field expressions in
this region reduce to Fresnel integrals.
32.
4.4.3 Reactive Near‐Field Region
If the distance of observation is smaller than the inner boundary of the Fresnel
region, this region is usually designated as reactive near‐field with inner and outer
boundaries defined by
0.62 / > 0 (4‐54)
In summary, the space surrounding an antenna is divided into three regions
whose boundaries are determined by
Reactive near‐field 0.62 / > 0 (4‐55a)
/
Radiating near‐field (fresnel) 2 0.62 / (4‐55b)
/
Far‐field (fraunhofer) 2 0.62 / (4‐55c)
33.
4.5 FINITE LENGTH DIPOLE
The techniques developed previously can be used to analyze the radiation
characteristics of a linear dipole of any length. To reduce the mathematical
complexities, it will be assumed that the dipole has a negligible diameter.
4.5.1 Current Distribution
For a very thin dipole (ideally zero diameter), the current distribution can be
written, to a good approximation, as
, 0
0, 0, (4‐56)
, 0
This distribution assumes that the antenna is
center‐fed
the current vanishes at the end points.
Experiments have verified that the current in a center‐fed wire antenna has
sinusoidal form with nulls at the end points.
34. For /2 an /2
nd t the current distrib
bution of (4‐56) is shown
f s
plo
otted in F
Figures 1.1
16(b) and
d (c), respeectively. T
The geommetry of th
he antenn na is that
shown in Figure 4.5.
35.
4.5.2 Radiated Fields: Element Factor, Space Factor, and Pattern
Multiplication
Since closed form solutions, which are valid everywhere, cannot be obtained
for many antennas, the observations will be restricted to the far‐field region.
The finite dipole antenna is subdivided into a number of infinitesimal dipoles
of length ’. For an infinitesimal dipole of length dz’ positioned along the z‐axis
at z’, the electric and magnetic field components in the far field are given as
, ,
(4‐26a) ′ (4‐57a)
(4‐26b) (4‐57b)
, ,
(4‐26b) ′
(4‐57c)
where R is given by (4‐39) or (4‐40).
Using the far‐field approximations given by (4‐46), (4‐57a) can be written as
36. , ,
′ (4‐58)
Summing the contributions from all the infinitesimal elements to integration. Thus
/ /
/ /
, , ′ (4‐58a)
The factor outside the brackets is designated as the element factor
And that within the brackets as the space factor.
For this antenna, the element factor is equal to the field of a unit length
infinitesimal dipole located at a reference point. The total field of the antenna is
equal to the product of the element and space factors.
For the current distribution of (4‐56), (4‐58a) can be written as
′
4 / 2
/
′ ′ (4‐60)
⇒ (4‐62a)
37. The total component can be written as
(4‐62b)
4.5.3 Power Density, Radiation Intensity, and Radiation Resistance
For the dipole, the average Poynting vector can be written as
∗
∗ ∗
| |
| | (4‐63)
and the radiation intensity as
| |
(4‐64)
The normalized elevation power patterns, for /4, /2, 3/4, and are
shown in Figure 4.6. The current distribution of each is given by (4‐56). The power
patterns for an infinitesimal dipole ≪ is also included for
comparison.
38. It is found that the 3‐dB 0
0 330 30
beamwidth of each is equal to
-10
≪ : 3dB beamwidth 90
300 60
-20
/4: 3dB beamwidth 87
/2: 3dB beamwidth 78 -30
3/4: 3dB beamwidth 64 -40 270 90
: 3dB beamwidth 47.8 -30
As the length of the antenna -20
240 120
increases, the beam becomes -10
narrower. Because of that, the 0 210 150
180
directivity should also increase with
1/50 1/4 1/2
length. 3/4 1
As the dipole’s length increases beyond one wavelength , the number
of lobes begin to increase. The normalized power pattern for a dipole with
1.25 is shown in Figure 4.7.
39.
Figure 4.7(a) is the
e three‐di
imensiona
al pattern
n
Figure 4.7(b) is the
e two‐dim
mensional pattern
The cuurrent di istribution for th dipole with
n he es
/4, /2, , 3/ and 2, as given by (4
/2, 4‐56), is
shown in Figure 4.8.
0
0 330 30
-1
10
300 60
-2
20
-3
30
-4 270
40 90
-3
30
-2
20
240 120
-1
10
0 210 150
180 Figure 4.8 Current dist
tributions
Fig
gure 4.7 Thr
ree‐ and twoo‐dimensionnal amplitudde patterns f ength of a li
for a thin along the le inear wire
and sinuso
dipole of l = 1.25 t distribution.
oidal current antenna.
To find the total power radiated the average Po
d r d, oynting ve
ector of (4‐63) is
int
tegrated o
over a sphhere of ra
adius r. Th
hus
40. ∯ ∙ ∮ ∙
| |
∮ (4‐66)
After some extensive mathematical manipulations, it can be reduced to
| | 1
2 2
4 2
/2 2 2 (4‐68)
where C 0.5772 (Euler’s constant) and Ci x and Si x are the cosine and
sine integrals given by
; 4 68a, b
The radiation resistance can be obtained using (4‐18) and (4‐68)
2 1
2 2
| | 2 2
/2 2 2 (4‐70)
41.
4.5.4 Directivity
The directivity was defined mathematically by (2‐22), or
, |
4 (4‐71)
,
where F , is related to the radiation intensity U by (2‐19), or
, (4‐72)
From (4‐64), the dipole antenna of length has
| |
F θ, ϕ F θ , B η (4‐73,73b)
Because the pattern is not a function of , (4‐71) reduces to
|
(4‐74)
,
The corresponding values of the maximum effective aperture are related to
the directivity by
(4‐76)
42.
4.5.5 Inpu
ut Resista
ance
The inp
put imped
dance was defined
d as“the ratio of t
the voltag
ge to curr
rent at a
pa of term
air minals or the ratio of the appropri
r iate comp
ponents of the ele
o ectric to
ma
agnetic fie
elds at a p
point.”
The reaal part of the input
t impedan nce was deefined as the input
t resistanc
ce which
for a lossles
ss antenna reduces adiation resistance.
s to the ra
Th radiati
he ion resist
tance of a dipole of leng l with
e gth
sin current distribution
nusoidal c n is expres
ssed by (4
4‐70).
2
| |
1
2 2
2 2
/2 2 2 (4‐70
0)
43. By the definition
n, the rad
diation resistance i
is referred
d to the m
maximum
m current
whhich for so /4, 3/4, , etc.) do
ome lengths (l = / he input terminals
oes not occur at th
of the antennna.
To refe
er the radiation res
sistance to
o the inpu ut terminals of
the antenna, the ant tenna is f
first assum
med to be e lossless (RL =
0). Then th power at the in
he nput term
minals is e
equated to the
o
po
ower at th he currennt maximu um. Refer rring to Figure 4.10 0, we
can write
| | | |
⟹ (4‐77)
Figure 4.10 Current
here
wh distribution, m
maximum
does not occcur at the
R rad
diation re
esistance a
at input (f
feed) term
minals
minals.
input term
R = ra
adiation resistance
e at curren
nt maximu
umEq. (4‐
‐70)
I = cu
urrent maximum
I = cu
urrent at input term
minals
dipole of length l, the curre
For a d e input terminals (I ) is re
ent at the elated to
44.
the current maximum
m (I ) ref
ferring to Figure 4.10, by
(4‐78)
he input ra
Thus th adiation r
resistance a) can be written a
e of (4‐77a as
(4‐79)
gure 4.9 R
Fig Radiation resistanc
ce, input r
resistance
e and directivity of a thin dip
pole with
sinu
usoidal cu
urrent distribution.
.
45.
4.6 HALF‐WAVELENGTH DIPOLE
One of the most commonly used antennas is the half‐wavelength (l = /2)
dipole. Because
Its radiation resistance is 73 ohms very near the 50/75‐ohm characteristic
impedances of some transmission lines,
Its matching to the line is simplified especially at resonance.
The electric and magnetic field components of a half‐wavelength dipole can be
obtained from (4‐62a) and (4‐62b) by letting l = /2.
, (4‐84, 85)
The time‐average power density and radiation intensity can be written,
respectively, as
| | | |
(4‐86)
| | | |
(4‐86)
46. Figure 4
4.6 and 4.11 show the two‐ and the t
three‐ dim
mensional
l pattern.
0
0 330 30
-10
0
300 60
-20
0
-30
0
-40
-40 270
0 90
-30
0
-20
0
240 120
-10
0
0 210 150
180
Th
he total po
ower radiated can be obtain
ned as a special cas
se of (4‐67
7)
| |
(4‐88)
| | | |
2 (4‐89)
By
y (4‐69)
2 0.577
72
ln 2 2
2 0.5
5772 1.838 0.02 2
2.435
(4‐90)
47. Using (4‐87), (4‐89) and (4‐90), the maximum directivity of the half‐wavelength
dipole reduces to
| /
4 4 1.643 (4‐91)
.
The corresponding maximum effective area is equal to
1.643 0.13 (4‐92)
and the radiation resistance, for a free‐space medium ( 120), is
| |
2 30 2.435 73 (4‐93)
The radiation resistance of (4‐93) is also the radiation resistance at the input
terminals (input resistance) since the current maximum for a dipole of /2
occurs at the input terminals. As it will be shown later, the imaginary part
associated with the input impedance of a dipole is a function of its length (for
/2, it is equal to j42.5). Thus the total input impedance for /2 is equal
to
73 42.5 (4‐93a)
48. To reduce the imaginary part of the input impedance to zero, the antenna is
matched or reduced in length until the reactance vanishes. The latter is most
commonly used in practice for half‐wavelength dipoles.
Depending on the radius of the wire, the length of the dipole for first
resonance is about 0.47 to 0.48; the thinner the wire, the closer
the length is to 0.48.
For thicker wires, a larger segment of the wire has to be removed from
/2 to achieve resonance.
49.
4.7 LINEAR ELEMENTS NEAR OR ON INFINITE PERFECT CONDUCTORS
The presence of obstacles, especially when it is near the radiating element, can
significantly alter the overall radiation properties.
The most common obstacle is the ground. Any energy from the radiating
element directed toward the ground undergoes a reflection. The amount of
reflected energy and its direction are controlled by the ground.
The ground is a lossy medium ( 0) whose effective conductivity increases
with frequency. Therefore it should be expected to act as a good conductor above
a certain frequency, depending primarily upon its composition and moisture
content. To simplify the analysis,
First assuming the ground is a perfect electric conductor, flat, and infinite.
The same procedure can also be used to investigate the characteristics of any
radiating element near any other infinite, flat, perfect electric conductor.
The effects that finite dimensions have on the radiation properties of a
radiating element can be accounted for by the use of the Geometrical Theory of
Diffraction and/or the Moment Method.
50.
4.7.1 Imag ge Theor ry
To analyze the performaance of an antenna near an infinite plane conductor,
n a n
vir
rtual sour duced to account for the reflections, which
rces (images) will be introd r
whhen comb bined with the real sources, form an n equivale
ent system
m. The eq quivalent
system give the same radiated field on and a
es above the conduc
ctor as th actual
he
system itself. Below the condu uctor, thee field is zero.
(a) Vertical electric dip
pole (b
b) Field com
mponents at point of ref
flection
Figure 4
4.12 Vertical electric dipole abo
ove an infin
nite, flat, p
perfect elec
ctric condu
uctor
51. The amount of reflection is generally determined by the respective
constitutive parameters of the media below and above the interface.
For a perfect electric conductor below the interface, the incident
wave is completely reflected and the field below the boundary is zero.
Vertical polarization
The tangential components of the electric field must vanish on the interface.
Thus for an incident electric field with vertical polarization, the polarization of the
reflected waves must be as indicated in the figure. To excite the polarization of the
reflected waves, the virtual source must also be vertical and with a polarity in the
same direction as that of the actual source (thus a reflection coefficient of 1).
Horizontal polarization
Another orientation of the source will be to have the radiating element in a
horizontal position, the virtual source (image) is also placed a distance h below the
interface but with a 180 polarity difference relative to the actual source (thus a
reflection coefficient of 1).
52. In addi
ition to electric so
e ources, artificial equivalent
t“magne
etic”sour
rces and
ma
agnetic co
onductors
s have been introduced.
Figure 4
4.13(a) displays th source and their imag for a electric plane
he es ges an
conducto The d
or. direction of the arrow id
dentifies the polarity. Sinc many
ce
problems s can be s
solved usiing duality
y.
Figure 4.
.13(b) illu
ustrates th source and their image when the obstacle is an
he es es t
flat, perfe “magnetic” conducto
infinite, f ect or.
(a) E
Electric con
nductor (b) Magnetic conductor
r
Figure 4
4.13 Electr gnetic sources and th
ric and mag heir images near elec
ctric (PEC) and
magnetic (PMC) cond
m ductors.
53.
4.7.2 Vertiical Electtric Dipoole
Assumi ing a vertical electr
ric dipole is placed a distancce above an infinite, flat,
pe
erfect elec
ctric cond
ductor as sshown in Figure 4.1 12(a).
For an ob
bservation point P1, there is
s a
rect wave
dir e.
On the interface, the incid
dent wave e is comp
pletely ref
flected annd the field below
the bounda ary is zero
o. The tanngential c
componen nts of thee electric field mus
st vanish
n the inter
on rface.
54.
1. Radiation pattern
(1) Direct component
The far‐zone direct component of the electric field of the infinitesimal dipole
of length , constant current , and observation point P is given according to
(4‐26a) by
(4‐94)
(2) The reflected component
The reflected component can be accounted for by the introduction of the
virtual source (image), as shown in Figure 4.14(a), and it can be written as
(4‐95, 4‐95a)
(3) The total field
The total field above the interface (z≥0) is equal to the sum of the direct and
reflected components as given by (4‐94) and (4‐95a). In general, we can write that
/ /
2 , 2 (4‐96a, b)
55. bservation r ≫ h , (4‐96a and (4‐96b) reduce us
For far‐field ob ns a) sing the
bin
nomial exxpansion tto
, (
(4‐97a,b)
(4‐98)
2 cos z 0
(4‐99)
0 0
56. The shaape and a amplitude e of the field is not t only con
ntrolled b
by the field of the
sin
ngle elem
ment but also by th positio
a he oning of t eleme relativ to the ground.
the ent ve
Th normalized pow patte
he wer erns for 0, /8, /4, 3 /8, /2, and have been
a
plo
otted in F
Figure 4.155..
0 5
15 0 15
0 30 30
-10
0 45 45
-20
0 60 60
-30
0
h=0 5
75 75
-40
0
h=1/8
8 h=3/8
-50
0 h=1/4
4 90
9 h=1/2 90
h=1
-40
0
10
05 105
-30
0
-20
0 120 120
-10
0 135 135
0 150 150
180 16
65 180 165
For h λ/4 more minor lobes, in n addition
n to the m
major one
es, are for
rmed. As
h attains v reater than λ, an even greater nu
values gr n umber of minor lobes is
int
troduced.
.
57. 0 15
0 30
-10 45
These are shown in Figure 4.16 for -20 60
h 2λ and 5λ . In general, the total -30
75
-40
number of lobes is equal to the integer that -50 h=2 90
h=5
is closest to -40
105
-30
2 120
number of lobes 1 -20
-10 135
0 150
180 165
2. Radiation power and directivity
The total radiated power over the upper hemisphere of radius r using
/
1
∙ | |
2
/
| | (4‐101)
which simplifies, with the aid of (4‐99), to
58.
(4‐102)
As kh → ∞ the radiated power, as given by (4‐102), is equal to that of an
isolated element.
As kh → 0, it can be shown that the power is twice that of an isolated element.
The radiation intensity can be written as
| | (4‐103)
(4‐103)
The directivity can be written as
4
(4‐104)
The maximum value occurs when kh 2.881 h 0.4585 , and it is equal to
6.566 which is greater than four times that of an isolated element (1.5). The
pattern for h 0.4585 is shown plotted in Figure 4.17 while the directivity, as
given by (4‐104), is displayed in Figure 4.18 for 0 h 5.
59.
Figure 4.17 Elevation plane amplitu
ude pattern
n of a vertica
al infinitesim
mal electric d
dipole at a h
height of
0.4585 ab
bove an infin
nite perfect electric connductor.
Us
sing (4‐10
02), the radiation re
esistance can be written as
| |
2 (4‐105)
(4‐19)
Th radiation resista
he 4.18 for 0 h
ance is plotted in Figure 4
p 0 5 when = /50
5
an
nd the element is ra
adiating innto free‐s
space (η 120).
60.
Figure 4.18 Directivity and radiation
i D n n resistance of a vertical infinitesimal electric d
dipole as a fu
unction of
its height above an infinite perfectt electric conductor
61.
3. monopo
ole
In prac
ctice, a w
wide use has been made o a quar
n of rter‐wavelength m monopole
( λ/4) m
mounted above a g
a ground plane, and fed by a coaxial line, as s
d a shown in
Fig
gure 4.199(a). For analysis purposes a λ/4 image is introduc
s, ced and it forms
the λ/2 eq quivalent of Figur 4.19(b). It should be e
re emphasize that the λ/2
ed
eqquivalent of Figure 4.19(b) g gives the correct field value
es for the
e actual sy ystem of
gure 4.19(a) only above the interface
Fig e (z 0, 0 θ /2).
gure 4.19 Quarter‐
Fig ‐waveleng
gth monopole on a
an infinite perfect e
electric co
onductor
62. Thus, the far‐zone electric and magnetic fields for the λ/4 monopole above
the ground plane are given, respectively, by (4‐84) and (4‐85).
, (4‐84, 4‐85)
The input impedance of a λ/4 monopole above a ground plane is equal to
one‐half that of an isolated λ/2 dipole. Thus, referred to the current maximum,
the input impedance Z is given by
Z monopole Z dipole 73 j42.5 36.5 j21.25 (4‐106)
63.
4.7.4 Antennas for Mobile Communication Systems
The dipole and monopole are two of the most widely used antennas for
wireless mobile communication systems.
An array of dipole elements is extensively used as an antenna at the base
station of a land mobile system while the monopole, because of its broadband
characteristics and simple construction, is perhaps to most common antenna
element for portable equipment, such as cellular telephones, cordless
telephones, automobiles, trains, etc.
An alternative to the monopole for the handheld unit is the loop. Other
elements include the inverted F, planar inverted F antenna (PIFA), microstrip
(patch), spiral, and others.
The variations of the input impedance, real and imaginary parts, of a vertical
monopole antenna mounted on an experimental unit are shown in Figure 4.21.
64.
65.
Figure 4.21 Input impedance, real and i
imaginary parts, of a ve
ertical mono
opole mount
ted on an
expe
erimental ce hone device
ellular teleph e.
It is a
apparent that the first reso
onance, around 1,0 MHz, is slowly varying
000 y
values of immpedance versus frequency and of desirable magnitude, for practical
e y, f
66.
im
mplementa
ation.
Above the first
t resonance, the im
mpedance
e is induct
tive. The s
second re
esonance
rapid changes in th values of the impedance. These values and variation of
he s e
im
mpedance are usually undesirable for practical implementation.
67.
4.7
7.5 Horizo
ontal Elec
ctric Dipo
ole
When the line eleme
n ear ent
is placed ho
orizontally
y relative
e to
the infinitte electric grou und
plaane, as sh
hown in Fiigure 4.24
4.
Fig
gure 4.24 Ho
orizontal eleectric dipole,, and its associated
imaage, above a
an infinite, f
flat, perfect nductor
t electric con
The aanalysis p
procedure of this is identica to the one of th vertica dipole.
e s al he al
Int
troducing an imag and assuming far field observat
g ge a tions, as shown in Figure
4.2
25(a, b),
68.
(a) Horizontal electric
c dipole abo
ove ground p
plane (b) Far‐
‐field observ
vations
gure 4.25 Ho
Fig orizontal ele
ectric dipole
e above an infinite perfe
ect electric conductor
oefficient is equal to R
Since the reflection co 1, The direct and the
ref
flect components c can be wr ritten as
(4‐111)
⟹ (4‐112)
ind the angle ψ , which is measu
To fi ured from the y‐
m ‐axis tow
ward the
ob
bservation
n point, w
we first for
rm
69. ∙ ∙ (4‐113)
⟹ 1 1 (4‐114)
Since for far‐field observations
for phase variations (4‐115a)
for amplitude variations (4‐115b)
the total field, which is valid only above the ground plane (z≥h; 0≤θ≤/2, 0≤
≤2), can be written as
E 1 sin sin 2 sin cos (4‐116)
Equation (4‐116) again consists of the product of the field of a single isolated
element placed symmetrically at the origin and a factor (within the brackets)
known as the array factor.