1. The document describes a new Motorola TMOS V power MOSFET transistor with improved performance over previous designs.
2. Key features include an on-resistance area product about half that of standard MOSFETs, allowing for faster switching speeds. It is designed for low voltage, high-speed switching applications like power supplies and motor controls.
3. The document provides detailed electrical and thermal specifications, characteristics curves, and application information for the transistor.
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ANALYSIS, DESIGN AND IMPLEMENTATION OF ZERO-CURRENT-SWITCHING RESONANT CONVER...hunypink
This paper presents a Buck type circuit structure, the designing of ZCS resonant Buck converter and analysis of the working principles involved. The designed buck converter uses ZCS technique and the function is realized so that the power form is converted from 12V DC to 5V DC (1A). A detailed analysis of zero current switching buck converters is performed and a mathematical analysis of the mode of operation is also presented. In order to reduce the switching losses in associated with conventional converters; resonant inductor and resonant capacitor (LC resonant circuit) is applied which helps to turn on-off the switch at zero current. The dc-dc buck converter receives the energy from the input source, when the switch is turned-on. If the switch is turned-off the LC resonant circuit pumps the energy by ensuring that the current does not come to zero. During the hardware implementation Ton, Toff, duty cycle & operating frequency values were determined and thoroughly tuned through the NE555 IC circuit. As a result of this various waveforms across capacitors,inductors and load resistor were observed. A simulation study was carried out and the effectiveness of the designed converter is verified by PSpice simulation results.
A novel single switch resonant power converterSameer Kasba
This deals with the novel single-switch resonant power converter for renewable energy generation applications. This circuit topology integrates a novel single switch resonant inverter with zero-voltage-switching (ZVS) with an energyblocking diode with zero-current-switching (ZCS).
Enhancing the Design of VRM for Testing Magnetic ComponentsIJERA Editor
The aim of this work is to design, build and test a voltage regulator module circuit (VRM) that can be used to
compare the performance of different magnetic component designs. The VRM will be used to convert the input
voltage (typically 12V) to a lower level which will supply a microprocessor load e.g. the Intel Pentium. The
work will include review of VRM circuit topologies for VRM 10.1 specification. Circuit design will be
performed for available controller IC. Simulation and analysis of the circuit in PSPICE and characterization
under transient conditions, a circuit will be designed for simulating a transient load change in PSPICE.
Finally all required components will be ordered and the circuit will be built and can be used for testing of
inductors and transformers
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ANALYSIS, DESIGN AND IMPLEMENTATION OF ZERO-CURRENT-SWITCHING RESONANT CONVER...hunypink
This paper presents a Buck type circuit structure, the designing of ZCS resonant Buck converter and analysis of the working principles involved. The designed buck converter uses ZCS technique and the function is realized so that the power form is converted from 12V DC to 5V DC (1A). A detailed analysis of zero current switching buck converters is performed and a mathematical analysis of the mode of operation is also presented. In order to reduce the switching losses in associated with conventional converters; resonant inductor and resonant capacitor (LC resonant circuit) is applied which helps to turn on-off the switch at zero current. The dc-dc buck converter receives the energy from the input source, when the switch is turned-on. If the switch is turned-off the LC resonant circuit pumps the energy by ensuring that the current does not come to zero. During the hardware implementation Ton, Toff, duty cycle & operating frequency values were determined and thoroughly tuned through the NE555 IC circuit. As a result of this various waveforms across capacitors,inductors and load resistor were observed. A simulation study was carried out and the effectiveness of the designed converter is verified by PSpice simulation results.
A novel single switch resonant power converterSameer Kasba
This deals with the novel single-switch resonant power converter for renewable energy generation applications. This circuit topology integrates a novel single switch resonant inverter with zero-voltage-switching (ZVS) with an energyblocking diode with zero-current-switching (ZCS).
Enhancing the Design of VRM for Testing Magnetic ComponentsIJERA Editor
The aim of this work is to design, build and test a voltage regulator module circuit (VRM) that can be used to
compare the performance of different magnetic component designs. The VRM will be used to convert the input
voltage (typically 12V) to a lower level which will supply a microprocessor load e.g. the Intel Pentium. The
work will include review of VRM circuit topologies for VRM 10.1 specification. Circuit design will be
performed for available controller IC. Simulation and analysis of the circuit in PSPICE and characterization
under transient conditions, a circuit will be designed for simulating a transient load change in PSPICE.
Finally all required components will be ordered and the circuit will be built and can be used for testing of
inductors and transformers
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Hierarchical Digital Twin of a Naval Power SystemKerry Sado
A hierarchical digital twin of a Naval DC power system has been developed and experimentally verified. Similar to other state-of-the-art digital twins, this technology creates a digital replica of the physical system executed in real-time or faster, which can modify hardware controls. However, its advantage stems from distributing computational efforts by utilizing a hierarchical structure composed of lower-level digital twin blocks and a higher-level system digital twin. Each digital twin block is associated with a physical subsystem of the hardware and communicates with a singular system digital twin, which creates a system-level response. By extracting information from each level of the hierarchy, power system controls of the hardware were reconfigured autonomously. This hierarchical digital twin development offers several advantages over other digital twins, particularly in the field of naval power systems. The hierarchical structure allows for greater computational efficiency and scalability while the ability to autonomously reconfigure hardware controls offers increased flexibility and responsiveness. The hierarchical decomposition and models utilized were well aligned with the physical twin, as indicated by the maximum deviations between the developed digital twin hierarchy and the hardware.
Sachpazis:Terzaghi Bearing Capacity Estimation in simple terms with Calculati...Dr.Costas Sachpazis
Terzaghi's soil bearing capacity theory, developed by Karl Terzaghi, is a fundamental principle in geotechnical engineering used to determine the bearing capacity of shallow foundations. This theory provides a method to calculate the ultimate bearing capacity of soil, which is the maximum load per unit area that the soil can support without undergoing shear failure. The Calculation HTML Code included.
Cosmetic shop management system project report.pdfKamal Acharya
Buying new cosmetic products is difficult. It can even be scary for those who have sensitive skin and are prone to skin trouble. The information needed to alleviate this problem is on the back of each product, but it's thought to interpret those ingredient lists unless you have a background in chemistry.
Instead of buying and hoping for the best, we can use data science to help us predict which products may be good fits for us. It includes various function programs to do the above mentioned tasks.
Data file handling has been effectively used in the program.
The automated cosmetic shop management system should deal with the automation of general workflow and administration process of the shop. The main processes of the system focus on customer's request where the system is able to search the most appropriate products and deliver it to the customers. It should help the employees to quickly identify the list of cosmetic product that have reached the minimum quantity and also keep a track of expired date for each cosmetic product. It should help the employees to find the rack number in which the product is placed.It is also Faster and more efficient way.
Welcome to WIPAC Monthly the magazine brought to you by the LinkedIn Group Water Industry Process Automation & Control.
In this month's edition, along with this month's industry news to celebrate the 13 years since the group was created we have articles including
A case study of the used of Advanced Process Control at the Wastewater Treatment works at Lleida in Spain
A look back on an article on smart wastewater networks in order to see how the industry has measured up in the interim around the adoption of Digital Transformation in the Water Industry.
Explore the innovative world of trenchless pipe repair with our comprehensive guide, "The Benefits and Techniques of Trenchless Pipe Repair." This document delves into the modern methods of repairing underground pipes without the need for extensive excavation, highlighting the numerous advantages and the latest techniques used in the industry.
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Ideal for homeowners, contractors, engineers, and anyone interested in modern plumbing solutions, this guide provides valuable insights into why trenchless pipe repair is becoming the preferred choice for pipe rehabilitation. Stay informed about the latest advancements and best practices in the field.
2. MTP3055V
2 Motorola TMOS Power MOSFET Transistor Device Data
ELECTRICAL CHARACTERISTICS (TJ = 25°C unless otherwise noted)
Characteristic Symbol Min Typ Max Unit
OFF CHARACTERISTICS
Drain–Source Breakdown Voltage
(VGS = 0 Vdc, ID = 250 µAdc)
Temperature Coefficient (Positive)
V(BR)DSS
60
—
—
65
—
—
Vdc
mV/°C
Zero Gate Voltage Drain Current
(VDS = 60 Vdc, VGS = 0 Vdc)
(VDS = 60 Vdc, VGS = 0 Vdc, TJ = 150°C)
IDSS
—
—
—
—
10
100
µAdc
Gate–Body Leakage Current (VGS = ± 20 Vdc, VDS = 0) IGSS — — 100 nAdc
ON CHARACTERISTICS (1)
Gate Threshold Voltage
(VDS = VGS, ID = 250 µAdc)
Temperature Coefficient (Negative)
VGS(th)
2.0
—
2.7
5.4
4.0
—
Vdc
mV/°C
Static Drain–Source On–Resistance (VGS = 10 Vdc, ID = 6.0 Adc) RDS(on) — 0.10 0.15 Ohm
Drain–Source On–Voltage (VGS = 10 Vdc)
(ID = 12 Adc)
(ID = 6.0 Adc, TJ = 150°C)
VDS(on)
—
—
1.3
—
2.2
1.9
Vdc
Forward Transconductance (VDS = 7.0 Vdc, ID = 6.0 Adc) gFS 4.0 5.0 — mhos
DYNAMIC CHARACTERISTICS
Input Capacitance
(VDS = 25 Vdc, VGS = 0 Vdc,
f = 1.0 MHz)
Ciss — 410 500 pF
Output Capacitance (VDS = 25 Vdc, VGS = 0 Vdc,
f = 1.0 MHz)
Coss — 130 180
Reverse Transfer Capacitance
f = 1.0 MHz)
Crss — 25 50
SWITCHING CHARACTERISTICS (2)
Turn–On Delay Time
(VDD = 30 Vdc, ID = 12 Adc,
VGS = 10 Vdc,
RG = 9.1 Ω)
td(on) — 7.0 10 ns
Rise Time (VDD = 30 Vdc, ID = 12 Adc,
VGS = 10 Vdc,
RG = 9.1 Ω)
tr — 34 60
Turn–Off Delay Time
VGS = 10 Vdc,
RG = 9.1 Ω) td(off) — 17 30
Fall Time
G = 9.1 Ω)
tf — 18 50
Gate Charge
(See Figure 8)
(VDS = 48 Vdc, ID = 12 Adc,
VGS = 10 Vdc)
QT — 12.2 17 nC
(See Figure 8)
(VDS = 48 Vdc, ID = 12 Adc,
VGS = 10 Vdc)
Q1 — 3.2 —(VDS = 48 Vdc, ID = 12 Adc,
VGS = 10 Vdc) Q2 — 5.2 —
Q3 — 5.5 —
SOURCE–DRAIN DIODE CHARACTERISTICS
Forward On–Voltage (1)
(IS = 12 Adc, VGS = 0 Vdc)
(IS = 12 Adc, VGS = 0 Vdc, TJ = 150°C)
VSD
—
—
1.0
0.91
1.6
—
Vdc
Reverse Recovery Time
(See Figure 15)
(IS = 12 Adc, VGS = 0 Vdc,
dIS/dt = 100 A/µs)
trr — 56 — ns
(See Figure 15)
(IS = 12 Adc, VGS = 0 Vdc,
dIS/dt = 100 A/µs)
ta — 40 —(IS = 12 Adc, VGS = 0 Vdc,
dIS/dt = 100 A/µs) tb — 16 —
Reverse Recovery Stored Charge QRR — 0.128 — µC
INTERNAL PACKAGE INDUCTANCE
Internal Drain Inductance
(Measured from contact screw on tab to center of die)
(Measured from the drain lead 0.25″ from package to center of die)
LD
— 3.5
4.5
—
nH
Internal Source Inductance
(Measured from the source lead 0.25″ from package to source bond pad)
LS — 7.5 — nH
(1) Pulse Test: Pulse Width ≤ 300 µs, Duty Cycle ≤ 2%.
(2) Switching characteristics are independent of operating junction temperature.
3. MTP3055V
3Motorola TMOS Power MOSFET Transistor Device Data
TYPICAL ELECTRICAL CHARACTERISTICS
RDS(on),DRAIN–TO–SOURCERESISTANCE(OHMS)
RDS(on),DRAIN–TO–SOURCERESISTANCE
(NORMALIZED)
0 1 2 3 4 5
0
8
16
24
VDS, DRAIN–TO–SOURCE VOLTAGE (VOLTS)
Figure 1. On–Region Characteristics
ID,DRAINCURRENT(AMPS)
2 4 6 8 10
0
8
16
24
ID,DRAINCURRENT(AMPS)
VGS, GATE–TO–SOURCE VOLTAGE (VOLTS)
Figure 2. Transfer Characteristics
0 4 8 16 24
0
0.10
0.20
0.30
RDS(on),DRAIN–TO–SOURCERESISTANCE(OHMS)
0 8 20 24
0.08
0.09
0.13
0.15
ID, DRAIN CURRENT (AMPS)
Figure 3. On–Resistance versus Drain Current
and Temperature
ID, DRAIN CURRENT (AMPS)
Figure 4. On–Resistance versus Drain Current
and Gate Voltage
– 50
0.6
0.8
1.2
1.6
0 20 50 60
1
10
100
TJ, JUNCTION TEMPERATURE (°C)
Figure 5. On–Resistance Variation with
Temperature
VDS, DRAIN–TO–SOURCE VOLTAGE (VOLTS)
Figure 6. Drain–To–Source Leakage
Current versus Voltage
IDSS,LEAKAGE(nA)
– 25 0 25 50 75 100 125 150
TJ = 25°C VDS ≥ 10 V TJ = – 55°C
25°C
100°C
TJ = 25°C
VGS = 0 V
VGS = 10 V
VGS = 10 V
ID = 6 A
9 V
8 V
6 V
5 V
4 V
7 V
4
12
20
3 5 7 9
4
12
20
VGS = 10 V
TJ = 100°C
25°C
– 55°C
12 20 4 12 16
10 30 40
0.05
0.15
0.25
0.10
0.12
0.14
0.11
1.0
1.4
TJ = 125°C
VGS = 10 V
15 V
175
4. MTP3055V
4 Motorola TMOS Power MOSFET Transistor Device Data
POWER MOSFET SWITCHING
Switching behavior is most easily modeled and predicted
by recognizing that the power MOSFET is charge controlled.
The lengths of various switching intervals (∆t) are deter-
mined by how fast the FET input capacitance can be charged
by current from the generator.
The published capacitance data is difficult to use for calculat-
ing rise and fall because drain–gate capacitance varies
greatly with applied voltage. Accordingly, gate charge data is
used. In most cases, a satisfactory estimate of average input
current (IG(AV)) can be made from a rudimentary analysis of
the drive circuit so that
t = Q/IG(AV)
During the rise and fall time interval when switching a resis-
tive load, VGS remains virtually constant at a level known as
the plateau voltage, VSGP. Therefore, rise and fall times may
be approximated by the following:
tr = Q2 x RG/(VGG – VGSP)
tf = Q2 x RG/VGSP
where
VGG = the gate drive voltage, which varies from zero to VGG
RG = the gate drive resistance
and Q2 and VGSP are read from the gate charge curve.
During the turn–on and turn–off delay times, gate current is
not constant. The simplest calculation uses appropriate val-
ues from the capacitance curves in a standard equation for
voltage change in an RC network. The equations are:
td(on) = RG Ciss In [VGG/(VGG – VGSP)]
td(off) = RG Ciss In (VGG/VGSP)
The capacitance (Ciss) is read from the capacitance curve at
a voltage corresponding to the off–state condition when cal-
culating td(on) and is read at a voltage corresponding to the
on–state when calculating td(off).
At high switching speeds, parasitic circuit elements com-
plicate the analysis. The inductance of the MOSFET source
lead, inside the package and in the circuit wiring which is
common to both the drain and gate current paths, produces a
voltage at the source which reduces the gate drive current.
The voltage is determined by Ldi/dt, but since di/dt is a func-
tion of drain current, the mathematical solution is complex.
The MOSFET output capacitance also complicates the
mathematics. And finally, MOSFETs have finite internal gate
resistance which effectively adds to the resistance of the
driving source, but the internal resistance is difficult to mea-
sure and, consequently, is not specified.
The resistive switching time variation versus gate resis-
tance (Figure 9) shows how typical switching performance is
affected by the parasitic circuit elements. If the parasitics
were not present, the slope of the curves would maintain a
value of unity regardless of the switching speed. The circuit
used to obtain the data is constructed to minimize common
inductance in the drain and gate circuit loops and is believed
readily achievable with board mounted components. Most
power electronic loads are inductive; the data in the figure is
taken with a resistive load, which approximates an optimally
snubbed inductive load. Power MOSFETs may be safely op-
erated into an inductive load; however, snubbing reduces
switching losses.
10 0 10 15 25
GATE–TO–SOURCE OR DRAIN–TO–SOURCE VOLTAGE (VOLTS)
C,CAPACITANCE(pF)
Figure 7. Capacitance Variation
VGS VDS
TJ = 25°CVDS = 0 V VGS = 0 V
1200
1000
800
600
400
200
0
20
Ciss
Coss
Crss
5 5
Ciss
Crss
5. MTP3055V
5Motorola TMOS Power MOSFET Transistor Device Data
VDS,DRAIN–TO–SOURCEVOLTAGE(VOLTS)
VGS,GATE–TO–SOURCEVOLTAGE(VOLTS)
DRAIN–TO–SOURCE DIODE CHARACTERISTICS
0.50 0.60 0.70 0.80 1.0
VSD, SOURCE–TO–DRAIN VOLTAGE (VOLTS)
Figure 8. Gate–To–Source and Drain–To–Source
Voltage versus Total Charge
IS,SOURCECURRENT(AMPS)
Figure 9. Resistive Switching Time
Variation versus Gate Resistance
RG, GATE RESISTANCE (OHMS)
1 10 100
t,TIME(ns)
VDD = 30 V
ID = 12 A
VGS = 10 V
TJ = 25°C
tf
td(off)
VGS = 0 V
TJ = 25°C
Figure 10. Stored Charge
0
QT, TOTAL CHARGE (nC)
2 4 6 8 13
ID = 12 A
TJ = 25°C
VGS
0
6
8
10
12
1000
100
10
1
10
6
2
0
12
8
4
60
50
40
30
20
10
0
VDS
1 3 5 7 9
4
0.55 0.65 0.75 0.85 0.90
2
0.95
QT
Q1 Q2
Q3
1110 12
td(on)
tr
0 4 12
IS, SOURCE CURRENT (AMPS)
QRR,STOREDCHARGE(C)
dIS/dt = 100 A/µs
VDD = 25 V
TJ = 25°C
0.08
0.10
0.11
0.12
0.13
0.09
2 6 8 10
Figure 11. Diode Forward Voltage versus Current
µ
SAFE OPERATING AREA
The Forward Biased Safe Operating Area curves define
the maximum simultaneous drain–to–source voltage and
drain current that a transistor can handle safely when it is for-
ward biased. Curves are based upon maximum peak junc-
tion temperature and a case temperature (TC) of 25°C. Peak
repetitive pulsed power limits are determined by using the
thermal response data in conjunction with the procedures
discussed in AN569, “Transient Thermal Resistance–General
Data and Its Use.”
Switching between the off–state and the on–state may tra-
verse any load line provided neither rated peak current (IDM)
nor rated voltage (VDSS) is exceeded and the transition time
(tr,tf) do not exceed 10 µs. In addition the total power aver-
aged over a complete switching cycle must not exceed
(TJ(MAX) – TC)/(RθJC).
A Power MOSFET designated E–FET can be safely used
in switching circuits with unclamped inductive loads. For reli-
able operation, the stored energy from circuit inductance dis-
sipated in the transistor while in avalanche must be less than
the rated limit and adjusted for operating conditions differing
from those specified. Although industry practice is to rate in
terms of energy, avalanche energy capability is not a
constant. The energy rating decreases non–linearly with an
increase of peak current in avalanche and peak junction tem-
perature.
Although many E–FETs can withstand the stress of drain–
to–source avalanche at currents up to rated pulsed current
(IDM), the energy rating is specified at rated continuous cur-
rent (ID), in accordance with industry custom. The energy rat-
ing must be derated for temperature as shown in the
accompanying graph (Figure 13). Maximum energy at cur-
rents below rated continuous ID can safely be assumed to
equal the values indicated.
6. MTP3055V
6 Motorola TMOS Power MOSFET Transistor Device Data
SAFE OPERATING AREA
TJ, STARTING JUNCTION TEMPERATURE (°C)
EAS,SINGLEPULSEDRAIN–TO–SOURCE
Figure 12. Maximum Rated Forward Biased
Safe Operating Area
0.1 10 100
VDS, DRAIN–TO–SOURCE VOLTAGE (VOLTS)
Figure 13. Maximum Avalanche Energy versus
Starting Junction Temperature
AVALANCHEENERGY(mJ)
ID,DRAINCURRENT(AMPS)
25 50 75 100 125
VGS = 20 V
SINGLE PULSE
TC = 25°C
ID = 12 A
1.0 150
t, TIME (s)
Figure 14. Thermal Response
r(t),NORMALIZEDEFFECTIVE
TRANSIENTTHERMALRESISTANCE
RθJC(t) = r(t) RθJC
D CURVES APPLY FOR POWER
PULSE TRAIN SHOWN
READ TIME AT t1
TJ(pk) – TC = P(pk) RθJC(t)
P(pk)
t1
t2
DUTY CYCLE, D = t1/t2
Figure 15. Diode Reverse Recovery Waveform
di/dt
trr
ta
tp
IS
0.25 IS
TIME
IS
tb
1.0
100
0.1 0
75
25
10
1.0
0.1
0.01
0.2
D = 0.5
0.05
0.01
SINGLE PULSE
0.1
1.0E–05 1.0E–04 1.0E–03 1.0E–02 1.0E–01 1.0E+00 1.0E+01
dc
100 µs
1 ms
10 ms
10 µs
RDS(on) LIMIT
THERMAL LIMIT
PACKAGE LIMIT
50
0.02
175
7. MTP3055V
7Motorola TMOS Power MOSFET Transistor Device Data
PACKAGE DIMENSIONS
CASE 221A–06
ISSUE Y
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION Z DEFINES A ZONE WHERE ALL
BODY AND LEAD IRREGULARITIES ARE
ALLOWED.
STYLE 5:
PIN 1. GATE
2. DRAIN
3. SOURCE
4. DRAIN
DIM MIN MAX MIN MAX
MILLIMETERSINCHES
A 0.570 0.620 14.48 15.75
B 0.380 0.405 9.66 10.28
C 0.160 0.190 4.07 4.82
D 0.025 0.035 0.64 0.88
F 0.142 0.147 3.61 3.73
G 0.095 0.105 2.42 2.66
H 0.110 0.155 2.80 3.93
J 0.018 0.025 0.46 0.64
K 0.500 0.562 12.70 14.27
L 0.045 0.060 1.15 1.52
N 0.190 0.210 4.83 5.33
Q 0.100 0.120 2.54 3.04
R 0.080 0.110 2.04 2.79
S 0.045 0.055 1.15 1.39
T 0.235 0.255 5.97 6.47
U 0.000 0.050 0.00 1.27
V 0.045 ––– 1.15 –––
Z ––– 0.080 ––– 2.04
B
Q
H
Z
L
V
G
N
A
K
F
1 2 3
4
D
SEATING
PLANE–T–
C
ST
U
R
J
8. MTP3055V
8 Motorola TMOS Power MOSFET Transistor Device Data
Motorolareserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit,
and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters can and do vary in different
applications. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola does
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