a switched capacitor bidirectional converter with high gain

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 33, NO. 11, NOVEMBER 2018 9459
A Switched-Capacitor Bidirectional DC–DC
Converter With Wide Voltage Gain Range for Electric
Vehicles With Hybrid Energy Sources
Yun Zhang , Member, IEEE, Yongping Gao, Lei Zhou , and Mark Sumner , Senior Member, IEEE
Abstract—A switched-capacitor bidirectional dc–dc converter
with a high step-up/step-down voltage gain is proposed for elec-
tric vehicles with a hybrid energy source system. The converter
presented has the advantages of being a simple circuit, a reduced
number of components, a wide voltage-gain range, a low voltage
stress, and a common ground. In addition, the synchronous rec-
tifiers allow zero voltage switching turn-on and turn-off without
requiring any extra hardware, and the efficiency of the converter
is improved. A 300 W prototype has been developed, which vali-
dates the wide voltage-gain range of this converter using a variable
low-voltage side (40–100 V) and to give a constant high-voltage
side (300 V). The maximum efficiency of the converter is 94.45%
in step-down mode and 94.39% in step-up mode. The experimen-
tal results also validate the feasibility and the effectiveness of the
proposed topology.
Index Terms—Bidirectional dc–dc converter (BDC), electric
vehicles (EVs), hybrid energy source system (HESS), switched-
capacitor, synchronous rectification, wide voltage-gain range.
I. INTRODUCTION
TO ADDRESS the challenges of fossil fuels as the primary
energy source for transport (including reducing stockpiles
and polluting emissions) [1], [2], electric vehicles (EVs) pow-
ered by battery systems with low or zero polluting emissions,
are increasing in popularity. Although the developed advance-
ment of batteries can provide higher population performance for
EVs, the unlimited charging or discharging current (i.e., inrush
current) from batteries will result in shorter battery cycle life,
as well as reducing the efficiency [3]. The combination of a
battery and supercapacitors as a hybrid energy source system
(HESS) for EVs is considered as a good way to improve overall
vehicle efficiency and battery life [4]. Supercapacitors have ad-
vantages of high power density, high-cycle life, and very good
Manuscript received November 8, 2017; accepted December 22, 2017. Date
of publication January 1, 2018; date of current version August 7, 2018. This
work was supported in part by the National Natural Science Foundation of
China under Grant 51577130, and in part by the Research Program of Ap-
plication Foundation and Advanced Technology of Tianjin China under Grant
15JCQNJC03900. Recommended for publication by Associate Editor Dr. Babak
Fahimi. (Corresponding author: Yun Zhang.)
Y. Zhang, Y. Gao, and L. Zhou are with the School of Electrical and In-
formation Engineering, Tianjin University, Nankai 300072, China (e-mail:
zhangy@tju.edu.cn; gaoyongping@tju.edu.cn; Luxuszl@163.com).
M. Sumner is with the Department of Electrical and Electronic Engineering,
University of Nottingham, Nottingham NG9 2QJ, U.K (e-mail: mark.sumner
@nottingham.ac.uk).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2017.2788436
charge/discharge efficiency. They can also provide a large tran-
sient power virtually instantaneously and are, therefore, suitable
for meeting sudden EV power changes such as acceleration or
meeting an incline. The HESS can make full use of the per-
formance of batteries and supercapacitors: the supercapacitors
supply power for acceleration and regenerative braking with the
battery meeting the requirement of high energy storage density
for long range operation [5]. A challenge for the HESS is that
the terminal voltage of supercapacitors is low, and varies over
a wide range as they are charged or discharged. Therefore, a
bidirectional dc–dc converter with a wide voltage-gain range is
desired for the HESS to connect low-voltage supercapacitors
with a high-voltage dc bus.
There are two broad classifications for bidirectional dc–dc
converters (BDCs), namely isolated converters and nonisolated
converters. Isolated converters, such as half-bridge and full-
bridge topologies are implemented using a transformer [6]–[8].
In addition, the half-bridge converter in [6] needs a center-
tapped transformer, which results in a complex structure, and
the full-bridge converters in [7] and [8] require a higher num-
ber of semiconductor devices. High-frequency transformers and
coupled inductors can be used in isolated converters to obtain
high step-up and step-down ratios [9]–[11]. However, in [9], the
realization of bidirectional power flow requires ten power semi-
conductors and two inductors. The converter in [10] requires two
inductors in addition to the transformer, and three inductors are
used for the converter in [11]. The structure of these converters
is complex, the cost is high, and it is difficult to standardize the
design. When the turns ratio of the high-frequency transformer
increases, the number of winding turns increase correspondingly
and the leakage inductance of the transformer may result in high
voltage spikes across the main semiconductors during switching
transitions. In order to reduce the voltage stress caused by the
leakage inductance, a BDC with an active clamp circuit in [12]
and a full-bridge BDC with a Flyback snubber circuit in [13]
were proposed. Besides, the dual active bridge converter in [14]
and the phase-shift full-bridge converter in [15] also utilized the
leakage inductance to achieve the soft-switching, and the en-
ergies stored in the leakage inductance were transferred to the
load. When the input and output voltages do not match the turns
ratio of the transformer, the power switch losses will increase
dramatically [16], which reduces the efficiency of the converter.
For nonisolated topologies, such as Cuk and Sepic/zeta con-
verters, their efficiencies are low [17], [18] as they use cascaded
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9460 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 33, NO. 11, NOVEMBER 2018
configurations of two power stages. Conventional buck–boost
converters are good candidates for low-voltage applications due
to their high efficiency and low cost. Unfortunately, the draw-
backs of narrow voltage conversion range, high voltage stress
and extreme duty cycle for the semiconductors make them un-
suitable for application to EV HESS. The voltage gain of the
BDC in [19] is greatly improved, but the voltage stress across
the power semiconductors is still equal to that of the high-
voltage side. The voltage stress across the power semiconduc-
tors of the bidirectional three-level dc–dc converters in [20] and
[21] is half that of conventional buck–boost converters, but its
voltage-gain range is still small. In addition, the low-voltage and
high-voltage side grounds of this converter are connected by a
power semiconductor, and therefore, the potential difference
between the two grounds is a high-frequency pulse width mod-
ulation (PWM) voltage, which may result in extra maintenance
issues and electromagnetic interference (EMI) problems. The
low-voltage and high-voltage sides of the bidirectional three-
level dc–dc converter in [22] share a common ground, but the
voltage-gain of this converter is still limited. In addition, this
converter requires complicated control scheme to balance the
flying-capacitor voltage. A high bidirectional voltage conver-
sion ratio with lower voltage stresses across the power semi-
conductors can be achieved by the converter of [23] with a
reasonable duty ratio, but the converter still has many problems
such as a large number of components, and a high-frequency
PWM voltage between the low-voltage and high-voltage sides.
The multilevel converter in [24] can achieve a high voltage
gain with low voltage stress across the power semiconductors.
However, this converter needs a higher number of power semi-
conductors, which leads to increased losses and higher cost.
Switched-capacitor converter structures and control strategies
are simple and easy to expand. They use different charging and
discharging paths for the capacitors to transfer energy to either
the low-voltage or the high-voltage side to achieve a high volt-
age gain. Thus, the switched-capacitor converter is considered to
be an effective solution to interface the supercapacitors with the
high-voltage dc bus. Single capacitor bidirectional switched-
capacitor converters were proposed in [25] and [26], but the
converter’s efficiency is low. The efficiency of the converter in
[27] has been improved through soft-switching technology, but
it required many extra components. Peng et al. [28] proposed a
multilevel bidirectional converter with very low-voltage stress
across the power semiconductors, but twelve semiconductors
are needed, and the drawbacks of low voltage gain, complex
control, and structure limit its application. The high voltage
gain BDCs in [29] and [30] need only four semiconductors.
However, the maximum voltage stress of the converter in [29] is
that of the high-voltage side, and the maximum voltage stress of
the converter in [30] is higher than that of the high-voltage side,
which will increase switching losses and reduce the conver-
sion efficiency of these converters. The bidirectional converter
in [31] only requires three semiconductors, but its voltage-gain
range is still small. In addition, the low-voltage and high-voltage
side grounds of this converter are connected by an inductor,
which will also generate extra EMI problems. Finally, the con-
verter in [32] has improved the conversion efficiency greatly,
Fig. 1. Proposed topology of the switched-capacitor BDC.
but it needs three inductors and a higher number of power semi-
conductors, which increases the conduction losses and makes
the design more challenging. Although exponential switched-
capacitor converters have high step-up capabilities, they operate
relatively poorly with respect to the switch and capacitor volt-
age stresses, as they involve several different higher voltage
levels [33].
To meet the requirements for the bidirectional converter for
the supercapacitor in an EV HESS, a high-ratio BDC, which uses
synchronous rectification, is proposed in this paper, as shown in
Fig. 1. The main contribution of the proposed converter lies in
the integrated advantage of having a wide voltage-gain range,
in the case of requiring less number of components with the
reduced voltage stress. In addition, the synchronous rectifiers
allow ZVS turn-on and turn-off without requiring any extra
hardware. The efficiency of the power conversion is, therefore,
improved, as well as the utilization of the power switches. Al-
though the proposed converter has a high voltage gain, it is built
without the magnetic coupling, and it can simplify the converter
design due to eliminating the need for coupled-inductor. Finally,
the proposed converter is suitable for EV applications because
its input inductor can provide a continuous current, and the
switched-capacitors can also be taken advantage of efficiently
with the dynamic balanced switched-capacitor voltages.
The paper is organized as follows. In Section II, the topology
of the switched-capacitor BDC is presented. In Section III, the
operating principles of the proposed converter are analyzed.
The steady-state characteristics of the converter are analyzed in
Section IV and experimental results are presented in Section V.
II. PROPOSED CONVERTER
Fig. 1 shows the proposed switched-capacitor BDC, which
is composed of four power semiconductors Q1 − Q4, four ca-
pacitors and one inductor L. Clow, and Chigh are the energy
storage/filter capacitors of the low-voltage and high-voltage
sides, and C1, C2 are the switched capacitors. L is an en-
ergy storage/filter inductor. In addition, power semiconductors
Q2 − Q4, and C1, C2, Chigh form the switched-capacitor net-
work, including switched-capacitor units C1 − Q2, C2 − Q3,
and Chigh − Q4. ilow, ihigh are the currents through the low-
voltage and high-voltage sides, Ulow, UC 1, UC 2, Uhigh are the
voltages across Clow, C1, C2 and Chigh, respectively.

ZHANG et al.: SWITCHED-CAPACITOR BDC WITH WIDE VOLTAGE GAIN RANGE FOR ELECTRIC VEHICLES WITH HYBRID ENERGY SOURCES 9461
Fig. 2. Typical waveforms of the proposed converter in step-up mode.
III. OPERATING PRINCIPLES
To simplify the steady-state analysis of the proposed con-
verter, the operating conditions are assumed to be as follows:
1) All the power semiconductors and energy storage components
of the converter are treated as ideal, and the converter operates
in the continuous conduction mode. 2) All the capacitances are
large enough that each capacitor voltage is considered constant
over each switching period.
A. Step-Up Mode
When the energy flows from the low-voltage side to the high-
voltage side, the output voltage Uhigh is stepped up from Ulow by
controlling the power semiconductor Q1, and the antiparallel
diodes of Q2, Q3 and Q4. UQ1, UQ2, UQ3, and UQ4 are the
voltage stresses across the corresponding power switches in
step-up mode. d1 = dBoost is the duty cycle of Q1. Fig. 2 shows
the typical waveforms in the step-up mode, and Fig. 3 shows
the current-flow paths of the proposed converter.
Mode I: Power semiconductor Q1 is turned ON. The antipar-
allel diode of Q3 turns ON, while the antiparallel diodes of Q2
and Q4 turn OFF. The current-flow paths of the proposed con-
verter are shown in Fig. 2(a). The energy of the dc source Ulow
is transferred to inductor L. Meanwhile, C1 is being charged by
capacitor C2. Chigh provides energy for the load.
Mode II: Power semiconductor Q1 and the antiparallel diode
of Q3 are OFF, while the antiparallel diodes of Q2 and Q4 are
ON. The current-flow paths of the proposed converter are shown
in Fig. 2(b). C2 charges from inductor L. Meanwhile, C1 is
discharging and Chigh is charging. The dc source Ulow, L and C1
provide energy for the load.
As shown in Figs. 2 and 3, when the proposed switched-
capacitor bidirectional converter operates in the step-up mode,
the currents flow into the corresponding antiparallel diodes. This
will result in lower efficiency, as well as lower utilization of
the power semiconductors. Therefore, a high step-up/step-down
ratio switched-capacitor BDC with synchronous rectification is
proposed further in this paper.
Fig. 4 shows the principle of operation of the synchronous
rectification for the proposed switched-capacitor BDC in the
step-up mode. The power semiconductor Q1 switches according
to the gate signal S1 shown in Fig. 4(a). During the dead time td,
the current must flow in the corresponding antiparallel diodes
Fig. 3. Current-flow paths of the proposed converter in the step-up mode.
(a) Mode I S1 = 1. (b) Mode II S1 = 0.
Fig. 4. Synchronous rectification operating principle for the proposed bidi-
rectional converter. (a) Gate signals and dead time in the step-up mode.
(b) Current-flow paths in the step-up mode.
of Q2, Q3 and Q4, as shown in Fig. 4(b). Otherwise, the current
will flow in the controlled power semiconductors Q2, Q3, and
Q4 due to their lower on-state resistance and on-state voltage
drop using the gate signals S2, S3, and S4 shown in Fig. 4(a).
In addition, when Q2, Q3, and Q4 are operating in synchronous
rectification, their gate signals will be turned OFF in advance by
the dead-time td. During the dead-time td, the currents flow in

Fig. 5. Typical waveforms of the proposed converter in step-down mode.
the corresponding antiparallel diodes of Q2, Q3, and Q4, and
their voltage stress across them are close to zero due to the
forward voltage drops of the antiparallel diodes, as shown in
Fig. 4(b). As a result, the controlled MOSFETS of Q2, Q3, and
Q4 are turned OFF with the ZVS. Similarly, the gate signals of
Q2, Q3 and Q4 will be turned ON by delaying the dead-time
td. The currents flow in the corresponding antiparallel diodes of
Q2, Q3, and Q4 during the dead-time td, and then flow in the
controlled MOSFETS of Q2, Q3, and Q4 due to their lower on-
state resistance, as shown in Fig. 4(b). As a result, the controlled
MOSFETS of Q2, Q3, and Q4 are also turned ON with the ZVS.
Thus, the efficiency of the converter can be further improved.
B. Step-Down Mode
When energy flows from the high-voltage side to the low-
voltage side, the output voltage Ulow is stepped down from Uhigh
by controlling the power semiconductors Q2, Q3, and Q4, and
the antiparallel diode of Q1. UQ1, UQ2, UQ3, and UQ4 are the
voltage stresses across the corresponding power switches in
step-down mode. The relationship between d2 and d4 can be
written as d2 = d4 = dBuck, where d2 and d4 are the duty cycles
of Q2 and Q4, respectively. Fig. 5 shows the typical waveforms
in the step-down mode, and Fig. 6 shows the current-flow paths
of the proposed converter.
Mode I: Power semiconductors Q2 and Q4 are turned ON.
Power semiconductor Q3 and the antiparallel diode of Q1 are
turned OFF. The current-flow paths of the proposed converter are
shown in Fig. 6(a). L is charging from capacitor C2. Meanwhile,
C1 is charging from Chigh and Uhigh. The dc source Uhigh, L and
C2 provide energy for the load.
Mode II: Power semiconductor Q3 and the antiparallel diode
of Q1 turned ON, while power semiconductors Q2 and Q4 turned
OFF. The current-flow paths of the proposed converter are shown
in Fig. 6(b). L is discharging. Meanwhile, C2 is charging from
capacitor C1, and Chigh is charging from Uhigh. L provides energy
for the load.
Fig. 7 shows the synchronous rectification operating prin-
ciple for the proposed switched-capacitor BDC in the step-
down mode. The power semiconductors Q2, Q3, and Q4 switch
Fig. 6. Current-flow paths of the proposed converter in the step-down mode.
(a) Mode I S2 S3 S4 = 101. (b) Mode II S2 S3 S4 = 010.
according to gate signals S2, S3, and S4, shown in Fig. 7(a). Dur-
ing the dead time td, the current must flow in the corresponding
antiparallel diodes of Q1, as shown in Fig. 7(b). Otherwise, the
current can flow in the controlled power semiconductors Q1 due
to its lower on-state resistance and on-state voltage drop using
the gate signal S1, shown in Fig. 7(a). As a result, the controlled
MOSFET of the synchronous rectifier Q1 is also turned ON and
turned OFF with ZVS.
C. Control Strategy of Bidirectional Power Flow
Based on the operating principles previously described, the
bidirectional power flow control strategy can be illustrated as
shown in Fig. 8. The block diagram representation of the ex-
perimental configuration is shown in Fig. 8(a). The voltages
Uhigh and Ulow, and the current ilow are obtained by sampling
the sensors, and the converter voltage and current loops are
implemented on a TMS320F28335 DSP controller.
As shown in Fig. 8(b), the proposed BDC switches be-
tween the step-up and the step-down modes, according to
the power flow control signal Uc, which is calculated by the
TMS320F28335 DSP controller. It operates in the step-up mode
when Uc = 0, the voltage Uhigh is controlled by the voltage con-
troller with the reference voltage Uref-Boost in the voltage-loop.
Meanwhile, the feedback current ilow is controlled by the current
controller using the reference current Iref-Boost in the current-
loop. The corresponding PWM schemes as shown in Figs. 2
and 4(a) are selected to generate the gate signals S1 − S4 in the
step-up mode.
In a similarly way, the converter operates in the step-down
mode when Uc = 1: the voltage Ulow is controlled by the voltage

Fig. 7. Synchronous rectification operation principle of the proposed bidi-
rectional converter. (a) Gate signals and dead time in the step-down mode.
(b) Current-flow paths in the step-down mode.
Fig. 8. Control strategy for bidirectional power flow. (a) Block diagram repre-
sentation of experimental configuration. (b) Realization of double closed-loop
control strategy.
controller with the reference voltage Uref-Buck, and the feedback
current ilow is controlled by the current controller with the ref-
erence current Iref-Buck, (which has the opposite polarity to the
reference current Iref-Boost). The corresponding PWM schemes
as shown in Figs. 5 and 7(b) are also selected to generate the
gate signals S1 − S4 in the step-down mode.
IV. ANALYSIS OF STEADY-STATE CHARACTERISTICS
A. Voltage-Gain in Steady-State
1) Voltage-Gain in Step-Up Mode: As shown in Figs. 2 and
3(a), C1 and C2 are connected in parallel when S1 = 1, so
that the voltages across C1 and C2 are equal. According to
Fig. 3(a) and (b), and the volt-second balance principle on L,
the following equations can be obtained:
⎧
⎪⎨
⎪⎩
dBoost × Ulow = (1 − dBoost) × (UC 2 − Ulow)
UC 1 + UC 2 = Uhigh
UC 1 = UC 2
. (1)
Therefore, by simplifying (1), the following equation can be
written:
UC 1 = UC 2 = 1
1−dBoost
Ulow
Uhigh = 2
1−dBoost
Ulow
. (2)
Based on the law of energy conservation, Ilow × Ulow =
Uhigh × Ihigh. Therefore
Ilow =
2
1 − dBoost
Ihigh (3)
where Ilow and Ihigh are the average currents of ilow and
ihigh, respectively, in the step-up mode. According to (2), the
voltage-gain of the proposed converter in the step-up mode is
2/(1 − dBoost), which is twice as large as the voltage-gain of
the conventional buck–boost converter. In addition, the voltage
stress of C1 and C2 can be reduced to half of the output voltage
Uhigh.
2) Voltage-Gain in the Step-Down Mode: As shown in
Figs. 5 and 6(b), C1 and C2 are connected in parallel when
S2S3S4 = 010, so that the voltages of C1 and C2 are equal.
According to Fig. 6(a) and (b), and the volt-second balance
principle on L, the following equation can be obtained:
⎧
⎪⎨
⎪⎩
dBuck × (UC 2 − Ulow) = (1 − dBuck) × Ulow
UC 1 + UC 2 = Uhigh
UC 1 = UC 2
. (4)
Therefore, by simplifying (4), the following equation can be
written:
UC 1 = UC 2 = 1
2 Uhigh
Ulow = dBuck
2 Uhigh
. (5)
By substituting Ilow × Ulow = Uhigh × Ihigh in (5)
Ihigh =
dBuck
2
Ilow (6)
where Ilow and Ihigh are the average currents of ilow and ihigh, re-
spectively, in the step-down mode. According to (5), the voltage-
gain of the proposed converter in the step-down mode is dBuck/2,
which is half of the voltage-gain of the conventional buck–boost
converter. In addition, the voltage stress of C1 and C2 are still
half of the input voltage Uhigh.
B. Voltage and Current Stresses of Power Semiconductors
1) Voltage Stress: As shown in Fig. 3(a) in the step-up mode
and Fig. 6(b) in the step-down mode, Q1 is turned ON and Q2
is turned OFF, so that Q2 and C2 are connected in parallel.
Therefore, the voltages across Q2 and C2 are equal. Similarly,
the voltages across the other power semiconductors can also be

obtained. According to (2) in the step-up mode and (5) in the
step-down mode, the voltage stress for the power semiconduc-
tors can be written as
⎧
⎪⎪⎨
⎪⎪⎩
UQ1 = UC 2 =
Uhigh
2
UQ2 = UQ3 = UC 1 =
Uhigh
2
UQ4 = Uhigh − UC 2 =
Uhigh
2
. (7)
Based on (7), all the voltage stresses of the power semi-
conductors and switched capacitors C1 and C2 are half of the
voltage Uhigh.
2) Current Stress: According to Fig. 3 and (3), the current
stress of the power semiconductors in the step-up mode can be
obtained by applying the ampere-second balance principle on
C1, C2, and Chigh as follows.
⎧
⎪⎪⎪⎪⎨
⎪⎪⎪⎪⎩
IQ1 = 2
1−dBoost
+ 1
dBoost
Ihigh
IQ2 = IQ4 = 1
1−dBoost
Ihigh
IQ3 = 1
dBoost
Ihigh
. (8)
In a similar way, according to Fig. 6 and (6), the current
stress of the power semiconductors in the step-down mode can
be obtained as (9)
⎧
⎪⎪⎪⎪⎨
⎪⎪⎪⎪⎩
IQ1 = 1 + dBuck
2(1−dBuck) Ilow
IQ2 = IQ4 = 1
2 Ilow
IQ3 = dBuck
2(1−dBuck) Ilow
. (9)
Based on (8) and (9), it can be seen that the current stress of
Q1 is slightly higher than that of the power semiconductors of
a conventional buck–boost converter operating under the same
conditions. However, it is easier (and cheaper) to choose a MOS-
FET with a higher rated current than the one with a higher rated
voltage. Furthermore, the proposed switched-capacitor bidirec-
tional converter can obtain a high voltage gain while the duty
cycle is in the range 0.5 < dBoost < 1 in the step-up mode or
0 < dBuck < 0.5 in the step-down mode. In addition, the volt-
age stress of all the power semiconductors is half of the high
side voltage Uhigh, and the current stress of Q2, Q3, and Q4 is
significantly lower than that of Q1 in both step-up and step-
down modes. Therefore, the difference of the current stress is
limited, and it will not affect the selection of the power semi-
conductors. Using these deductions comparisons can be drawn
between the proposed topology and the other bidirectional so-
lutions as shown in Table I.
The conventional buck–boost and the BDC in [22] need one
inductor, respectively, but their ideal voltage-gain 1/(1 − d) is
limited to a lower value due to the effects of parasitic resistance
and extreme duty cycles, and the lowest efficiency is less than
90%. It is noted that the voltage stress across the four semi-
conductors in the converter in [22] can be reduced by a half
compared with that of the conventional converter, due to the
use of two additional semiconductors and one flying capacitor.
The high voltage-gain BDCs in [29] and [30] need two induc-
tors, respectively. In addition, in [29], the maximum voltage
TABLE I
COMPARISONS BETWEEN PROPOSED AND OTHER BIDIRECTIONAL SOLUTIONS
Bidirectional
Solution
Voltage
Gain
Number of
Switches
Number of
Inductors
Voltage Stress
Buck/Boost
converter
1
1−d 2 1 Uhigh
Converter
in [22]
1
1−d 4 1 Uhigh/2
Converter
in [29]
2
1−d 4 2 Uhigh/2, Uhigh
Converter
in [30]
1
(1−d)2 4 2 Uhigh(1 − d), Uhigh,
Uhigh + Uhigh(1 − d)
Proposed
Converter
2
1−d 4 1 Uhigh/2
Fig. 9. Experimental prototype of the proposed switched-capacitor BDC.
stress across the semiconductors is the high side voltage Uhigh,
and in [30], the maximum voltage stress across the semicon-
ductors is Uhigh + Uhigh(1 − d). The converters in [29] and [30]
both have semiconductors with a voltage stress higher than or
equal to the high side voltage Uhigh, rather than Uhigh/2. For
the converter proposed in this paper, the number of main com-
ponents is between those of the converters described in [22]
and [30], the voltage stress across all the semiconductors is
Uhigh/2, and its voltage gain is higher than that of [22]. When
the step-up voltage gain is 6.25, the efficiency of the converter
in [30] is approximately equal to 91.2%, while the proposed
converter’s conversion efficiency is 91.9% with the same volt-
age gain. Moreover, the efficiency of the converter in [22] is
nearly equal to 90% when Ulow = 220 V,Uhigh = 340 V, and
Pn = 300 W, while the proposed converter’s efficiency reaches
94.39% when Ulow = 100 V,Uhigh = 300 V, and Pn = 300 W.
V. EXPERIMENTAL RESULTS AND ANALYSIS
In order to validate the theoretical analysis, a 300 W experi-
mental prototype for the proposed switched-capacitor BDC was
developed, as shown in Fig. 9. The parameters of the experiment
rig are shown in Table II.
A. Experimental Results in the Step-Up Mode
In order to build the initial voltages across the switched capac-
itors and eliminate the inrush current when the converter starts
up, a soft-starting circuit is adopted between the battery and the
input side of the proposed converter in this paper. Then, the low
voltage battery and the high-voltage dc bus are interfaced by
the proposed BDC, and the experimental results are shown in

TABLE II
EXPERIMENT PARAMETERS
Parameters Values
Rated power Pn 300 W
Storage/filter capacitors Clow and Chigh 520 µF
Switched-capacitors C1 and C2 520 µF
Storage/filter inductor L 353 µH
High side voltage Uhigh 300 V
Low side voltage Ulow 40–100 V
Switching frequency fs 20 kHz
Power semiconductors Q1 − Q4 IXTK 88N30P
Fig. 10. Experimental results of the soft start-up. (a) Input voltage Ulow and
the output voltage Uhigh. (b) Voltages across C1 and C2 .
Fig. 10. In Fig. 10(a), when the converter starts up, the input
voltage Ulow rises from 0 to 50 V gradually over 2 s, due to the
soft-starting circuits. Accordingly, the output voltage rises from
0 to 300 V (i.e., the reference voltage) gradually with a voltage
control loop. It is noticed that the output voltage Uhigh arrives
at the reference voltage (300 V) before the input voltage Ulow
reaches the battery voltage (50 V), because the voltage control
loop gets rid of the duty cycle limitation, and obtains the static
state when the input voltage Ulow rises to 40 V approximately.
In addition, as shown in Fig. 10(b), the switched capacitor volt-
ages UC 1 and UC 2 rise according to the output voltage Uhigh. It
is also noticed that switched capacitor voltages UC 1 and UC 2
still keep at half of the output voltage Uhigh due to the voltage
balance characteristic, especially in the soft start-up stage.
Fig. 11. PWM voltages of power semiconductors Q1 and Q2 .
Fig. 12. Voltages across C1 and Chigh under Ulow = 40 V and Uhigh = 300 V.
The voltage stress across the semiconductors and the capaci-
tors in the step-up mode for Ulow = 40 V and Uhigh = 300 V are
shown in Figs. 11 and 12. It can be seen in Fig. 11, that the duty
cycle of the active power semiconductor Q1 is dBoost = 0.73,
when the voltage-gain is 7.5. In addition, the PWM blocking
voltage of each power semiconductor is 150 V, namely half
of the high side voltage Uhigh, which validates the analysis in
Section IV. The voltages across C1 and Chigh are shown in
Fig. 12. The voltage stress of C1 is 150 V, which is also half of
the high-side voltage Uhigh. Therefore, the switched-capacitor
BDC can perform with a high voltage gain and a low voltage
stress across the semiconductors and the capacitors.
The voltage waveforms of the synchronous rectifiers of the
proposed converter in the step-up operating mode are shown
in Fig. 13. The current flows through the antiparallel diodes of
Q2, Q3, and Q4 during the dead time, and the blocking voltages
of Q2, Q3, and Q4 are around zero. Otherwise, the controlled
MOSFETS Q2, Q3, and Q4 are turned ON and turned OFF with
ZVS by synchronous rectification. The gate signal S3 and the
voltage stress of Q3 are shown in Fig. 13.
In the step-up mode, the output voltage stays constant around
the reference voltage 300 V by the action of the voltage control
loop. Fig. 14 illustrates the dynamic state of the output voltage
when the input voltage is changed from 100 to 40 V over a
period of 10 s. According to Fig. 14, when the input voltage
Ulow varies from 100 to 40 V, the output voltage remains at

Fig. 13. Gate signal and voltage stress of synchronous rectification power
semiconductor Q3 .
Fig. 14. Output voltage and the wide-range changed input voltage from
100 to 40 V in the step-up mode.
Fig. 15. PWM voltages of power semiconductors Q3 and Q4 .
300 V, which means the proposed converter can obtain a wide
voltage-gain range varying from 3 to 7.5.
B. Experimental Results in the Step-Down Mode
The voltage stress of the semiconductors and the capacitors
in the step-down mode for Ulow = 40 V and Uhigh = 300 V are
shown in Figs. 15 and 16. It can be seen in Fig. 15 that the duty
cycle of the active power semiconductor Q4 is dBoost = 0.27,
when the voltage-gain is 1/7.5. In addition, the PWM blocking
voltage of each power semiconductor is 150 V. The voltages
Fig. 16. Voltages across C2 and Chigh under Ulow = 40 V and Uhigh = 300 V.
Fig. 17. Gate signal and voltage stress of synchronous rectification power
semiconductor Q1 .
across C2 and Chigh are shown in Fig. 16. The voltage stress of
C2 is also 150 V. Obviously, it can be concluded that the voltage
stress of the semiconductors and the capacitors are also half of
the high-side voltage Uhigh in the step-down mode.
Fig. 17 shows the voltage waveforms of the synchronous
rectifier of the proposed converter in the step-down operating
mode. The current flows through the antiparallel diode of Q1
during the dead time, and the blocking voltage of Q1 is also
close to zero. Otherwise, the controlled MOSFETS Q1 is turned
ON and turned OFF with ZVS by synchronous rectification, as
shown in Fig. 17.
Fig. 18 illustrates the dynamic state of the output voltage
Ulow and the input voltage Uhigh when the output voltage is
controlled from 40 to 100 V and the input voltage is kept at
300 V. According to Fig. 18, under the control of the voltage
loop, when the input voltage stays at 300 V, the output voltage
Ulow can be controlled continuously over 8 s from 40 to 100 V,
which means the proposed converter can obtain a wide voltage-
gain range varying from 1/7.5 to 1/3.
C. Experiment Results for Bidirectional Power Flow Control
Fig. 19 shows the EV HESS, where the supercapacitor bank
is made up of CSDWELL model MODWJ001PM031Z2 su-
percapacitors. The battery in the HESS is a lithium iron phos-
phate battery, and a resistive load Pload is used to simulate the

Fig. 18. Input voltage and the wide-range output voltage from 40 to 100 V in
the step-down mode.
Fig. 19. Hybrid energy sources system of EVs.
electric vehicle load. In the HESS shown in Fig. 19, Ubat, Ibat,
and Pbat are the output voltage, output current and output power
of the battery, Usc, Isc, and Psc are the output voltage, output
current, and output power of the supercapacitor. In this exper-
iment, the output voltages of the battery and the supercapaci-
tors are 50 V and 40 V, respectively, and the electric vehicle’s
power varies with step changes between 400 W and 650 W (the
power difference 250 W is provided by the supercapacitors in-
stantaneously through the proposed converter). The proposed
switched-capacitor BDC in this paper is applied as the power
interface between the supercapacitor and the dc bus, and it op-
erates according to the control strategy shown in Fig. 8. In addi-
tion, filter control is adopted to determine the power distribution
between the battery and supercapacitors.
The experimental results of the bidirectional power flow con-
trol are shown in Fig. 20. Fig. 20(a) shows Ibat and Isc when the
proposed BDC is operating (i.e., the dc bus is powered by the
HESS). Fig. 20(b) shows Ibat and Isc when the proposed BDC
is not operating (i.e., the dc bus is just powered by the battery).
It can be seen from Fig. 20(a) that, when the dc bus power
demand is changed from 400 to 650 W with a step change,
the control system sets the control signal Uc = 0. At the same
time, the proposed switched-capacitor bidirectional converter
responds quickly and operates in the step-up mode. The current
Isc quickly goes to 6 A, and the instantaneous power provided by
the supercapacitor is nearly equal to the required power change
of the dc bus, avoiding any step change in current from the
battery. Following this process, the current of the battery rises
from 8 to 13 A gradually, and the current of the supercapacitor
falls to zero from Isc = 6 A to match the increase of the battery
Fig. 20. Experimental results of bidirectional power flow control. (a) Su-
percapacitors are taken into operation. (b) Supercapacitors are not taken into
operation.
current. Similarly, when the dc bus demand power is changed
from 650 to 400 W with a step change, the control system sets the
control signal Uc = 1. The proposed switched-capacitor bidirec-
tional converter responds quickly and operates in the step-down
mode. The current Isc quickly goes up to 6 A with the opposite
polarity. As a result, the current from the battery falls from 13
to 8 A gradually, and the current of the supercapacitor falls to
zero from Isc = −6 A.
If the proposed BDC is not operating, the battery has to supply
all the load demands by itself. It can be seen from Fig. 20(b)
that, when the dc bus demand power is changed from 400 to
650 W with a step change, the current Ibat needs to suddenly
increase from 8 to 13 A with a step change. When the dc bus
demand power is changed from 650 to 400 W with a step change,
the current Ibat suddenly decreases from 13 to 8 A with a step
change. Therefore, when the load power changes with a step, the
output current of the battery also has to change instantaneously.
This has a detrimental impact on the battery itself during the
electric vehicle’s acceleration and deceleration, as it shortens
the battery’s service life.
Comparing the experimental results of Fig. 20(a) and (b), it is
seen that when the dc bus demand power suddenly increases or
decreases, the proposed switched-capacitor bidirectional con-
verter can respond quickly according to the control signal Uc,
and the supercapacitor can compensate (take in or send out) the
power difference between the battery and the dc bus side to en-
sure that the current from the battery changes slowly. Therefore,
the overall aim of improving the battery life can be achieved.

Fig. 21. Efficiencies of the proposed switched-capacitor bidirectional con-
verter in step-up and step-down modes (Uhigh = 300 V,Ulow = 40–100 V,
Pn = 300 W).
Fig. 22. Calculated power loss distributions for the experiment when Ulow =
40 V, Uhigh = 300 V, and Pn = 300 W. (a) In step-up mode. (b) In step-down
mode.
The efficiencies of the proposed BDC in the step-up and step-
down modes were measured using a YOKOGAWA/WT3000
power analyzer and are shown in Fig. 21, when the high-side
voltage Uhigh is 300 V and the low-side voltage Ulow varies from
40 to 100 V or 100 to 40 V continuously. According to Fig. 21,
the measured efficiencies range from 90.08 to 94.39% in the
step-up mode, and from 90.86% to 94.45% in the step-down
mode. The efficiencies are improved when the low-side voltage
Ulow increases (due to the lower voltage-gain), and the efficiency
in the step-down mode is slightly higher than that in the step-
up mode. Moreover, the maximum efficiencies are 94.39% and
94.45% for step-up and step-down modes, respectively, when
the low-voltage side Ulow is 100 V.
The calculated power loss distributions for the experiment
when Ulow = 40 V,Uhigh = 300 V, and Pn = 300 W are shown
in Fig. 14. In step-up mode, the total losses of the converter are
13.548 W, and the loss distribution is shown in Fig. 22(a). By an-
alyzing the power loss distributions, it can be concluded that the
major loss comes from the inductor, namely the copper and core
losses of the inductor account for 38.566% of the total losses.
The capacitor losses account for 22.018% of the total losses.
The conduction and switching (turning ON and OFF) losses
of the semiconductors account for 19.922% and 19.494%, re-
spectively. In step-down mode, the total losses of the converter
are 12.508 W, and Fig. 22(b) shows the power loss distributions.
The largest power losses are also the copper and core losses of
the inductor, which account for 41.774% of the total losses. The
conduction losses and the switching (turning ON and OFF) losses
of the semiconductors account for 39.422%, and the remain-
ing 18.804% of the total losses is occupied by the capacitor
losses.
VI. CONCLUSION
A switched-capacitor BDC has been proposed. The
topology has a high step-up/step-down ratio and a wide
voltage-gain range, in the case of requiring less number of
components with the reduced voltage stress. The synchronous
rectifiers can be turned ON and turned OFF using ZVS, and the
efficiency is improved. The proposed BDC, which interfaces
the low voltage supercapacitor and the high-voltage dc bus, can
rapidly output or absorb the power difference due to a load step
change. It can satisfy the requirements of a complex dynamic re-
sponse, and effectively protect the battery from providing a step
change in current. Thus, the proposed BDC is suitable for the
power interface between the low-voltage supercapacitors and the
high-voltage dc bus of a HESS for EVs.
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Yun Zhang (M’13) was born in Jiangsu, China, in
1980. He received the B.S. and M.S. degrees in elec-
trical engineering from Harbin University of Science
and Technology, Harbin, China, in 2003 and 2006,
respectively, and the Ph.D. degree in electrical engi-
neering from Harbin Institute of Technology, Harbin,
China, in 2010.
In 2010, he joined the Tianjin University,
Tianjin, China, as a Lecturer with the School of Elec-
trical and Information Engineering, where he is cur-
rently an Associate Professor. From December 2016
to December 2017, he was an Academic Visitor with the Power Electronics,
Machines and Control Group, University of Nottingham, Nottingham, U.K. His
current research interests include topologies, modulation, and control strategies
of power converters for electric vehicles and microgrids.
Dr. Zhang is an Associate Editor of the Journal Of Power Electronics.
Yongping Gao was born in Shanxi, China. He
received the B.S. degree in electrical engineering
from China University of Mining and Technology,
Xuzhou, China, in 2015. He is currently working to-
ward the M.S. degree in electrical engineering from
Tianjin University, Tianjin, China, in 2015.
His current research interests include power elec-
tronics converters, and energy management.
Lei Zhou was born in Ningxia, China. He received
the B.S. degree in electrical engineering from Tianjin
University, Tianjin, China, in 2015. He is currently
working toward the M.S. degree in electrical engi-
neering from Tianjin University, Tianjin, China, in
2015.
His current research interests include dc–dc con-
verters, modeling, and analysis of dc–dc converters.
Mark Sumner (SM’05) received the B.Eng. degree
in electrical and electronic engineering from Leeds
University, Leeds, U.K., in 1986, and the Ph.D. de-
gree in induction motor drives from the University of
Nottingham, Nottingham, U.K., in 1992.
He was with Rolls Royce, Ltd., Ansty, U.K. He
was a Research Assistant with the University of
Nottingham, where he became a Lecturer in October
1992, and is currently a Professor of electrical en-
ergy systems. His research interests include control
of power electronic systems including sensorlessmo-
tor drives, diagnostics and prognostics for drive systems, power electronics for
enhanced power quality, and novel power system fault location strategies.

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