DoA Estimation

1. Int. J. Electron. Commun. (AEÜ) 68 (2014) 1016–1021
Contents lists available at ScienceDirect
International Journal of Electronics and
Communications (AEÜ)
journal homepage: www.elsevier.com/locate/aeue
A novel miniaturized power amplifier with nth harmonic suppression
M. Hayatia,b,∗
, S. Roshania
a
Electrical Engineering Department, Faculty of Engineering, Razi University, Tagh-E-Bostan, Kermanshah 67149, Iran
b
Computational Intelligence Research Centre, Razi University, Tagh-E-Bostan, Kermanshah 67149, Iran
a r t i c l e i n f o
Article history:
Received 16 November 2013
Accepted 17 May 2014
Keywords:
Power amplifier (PA)
Harmonic suppression
Low pass filter (LPF)
Matching networks
a b s t r a c t
In this paper, a novel miniaturized power amplifier (PA) matched by two proposed low pass filters (LPFs)
with nth harmonics suppression is presented. In the proposed PA, the LPFs are employed as an output
and input impedance transformer networks, which transform 50 to the desired impedances. In the
proposed PA the conventional output and input matching networks are eliminated, which results in
52% size reduction and 6% power added efficiency (PAE) improvement compared with the conventional
PA. Moreover, using the LPFs at the output and input impressively suppress the unwanted harmonics
(2nd–6th) with high level of attenuation. The proposed PA works at the 2.6 GHz, which is suitable for
long term evolution (LTE) applications. The measured and simulated results are in the good agreement,
which confirm the validity of the proposed method.
© 2014 Elsevier GmbH. All rights reserved.
1. Introduction
With the demand of high-speed data communication, the 4th
generation (4G), LTE system has attracted a lot of attentions.
For LTE applications, power amplifiers (PAs) should be low cost,
high integrated and light weight [1]. Nowadays, the growing
demand for higher data rates and the increasing number of wire-
less communications users have resulted into rapidly rising power
consumption. The energy efficiency of base stations should be con-
stantly improved in order to reduce the power loss. Significant
energy saving can be achieved by improving the efficiency of the
power amplifier (PA) of RF transmitters used in the base station
[2]. Different techniques have been proposed to increase the effi-
ciency of power amplifiers. Envelope elimination and restoration
(EER) [3], envelope tracking (ET) [4], Doherty amplifiers [5,6] and
varactor-based dynamic load modulation [7] are the most common
techniques being proposed to enhance the efficiency of PAs.
Power amplifier performances significantly depend on
their output matching networks (OMNs). Nowadays, exten-
sive researches have been performed on the OMNs to improve
harmonic suppression [8,9], PA’s efficiency improvement [10],
bandwidth enhancement [11], multiband capability, etc. The OMN
and input matching network (IMN) in the conventional PA occupy
∗ Corresponding author at: Electrical Engineering Department, Faculty of Engi-
neering, Razi University, Tagh-E-Bostan, Kermanshah 67149, Iran.
Tel.: +98 9188312041.
E-mail addresses: mohsen hayati@yahoo.com, hayati@razi.ac.ir (M. Hayati).
a large size, and they have several discontinuities between the
narrow and wide lines, which excite high order modes [1].
In [8,9], to enhance PAE and suppress unwanted harmonics, fil-
ters have been used in the PA structure, but in these works, there is
not any size reduction and OMN is still an indispensable part. In [1]
BPF is used as the OMN, which results in 21% size reduction com-
pared to the conventional PA and suppresses the 2nd harmonic, but
the miniaturization and harmonic suppression in this work are not
so prominent.
In the proposed structure, a miniaturized power amplifier inte-
grated by two low pass filters (LPFs) at input and output is proposed.
The applied LPF in this structure is based on a proposed filter in
[12]. It works as an IMN and OMN, therefore, the conventional out-
put and input matching networks are eliminated, which results in
ultra size reduction and high harmonics suppression. Compared
with reported works, the proposed PA demonstrates better size
reduction (52%) and suppresses the 2nd–6th harmonics with high
level of attenuation.
2. Design process
The schematic structures of the conventional PA, is shown in
Fig. 1. In the conventional PA, the output and input matching
networks occupied a large size, and they have several discontinu-
ities between the narrow and wide lines. The OMN and IMN are
used to transform 50 to the desired impedances, which shown
as ZL and ZS in Fig. 1.
The proposed PA is designed using a MW6S004N LDMOS tran-
sistor. The applied transistor is designed for class A or class AB base
http://dx.doi.org/10.1016/j.aeue.2014.05.003
1434-8411/© 2014 Elsevier GmbH. All rights reserved.

2. M. Hayati, S. Roshani / Int. J. Electron. Commun. (AEÜ) 68 (2014) 1016–1021 1017
Fig. 1. Schematic diagram of the conventional power amplifier, with conventional IMN, OMN.
Fig. 2. Transfer characteristics of the applied device.
station applications with frequency up to 2 GHz. Typical output
power is 4 W, IDQ = 50 mA and Vds = 28 V.
The transfer characteristics of the applied LDMOS transistor
MW6S004N, which performed with the Advanced Design System
(ADS) software (Agilent Technologies, Santa Clara, CA) is shown
in Fig. 2. The plot represents the range of gate voltages and the
corresponding mode of operation with a drain voltage of 28 V. The
amplifier is biased in class AB (quiescent drain current of 60 mA and
gate voltage of 2.8 V). Increasing of the quiescent current resulted
in better performances, but caused extra heating, so in the pro-
posed design the current did not increase very much and there is a
good trade of between heating and performances. Fig. 3 shows the
output characteristics of the applied device.
3. Load/source pull
The first step to design the PA is to perform load/source pull
simulation to find the optimum load/source impedances at the
desired frequency (2.6 GHz). The load/source-pull simulations for
the PA are performed with the Advanced Design System (ADS)
Fig. 3. Output characteristics of the applied device.
Fig. 4. Simulated optimum source impedances (ZS) of the main device at
2.5–2.7 GHz frequency range.
Fig. 5. Simulated optimum load impedances (ZL) of the main device at 2.5–2.7 GHz
frequency range.
software (Agilent Technologies, Santa Clara, CA) and resulting opti-
mum load/source impedances at 2.5–2.7 GHz frequency range are
shown in Figs. 4 and 5, while these values of the optimum load
and source impedances are listed in Table 1. The optimum load and
source impedances at the 2.6 GHz (desired frequency) are 12–14j
(ZL) and 7.5–12j (ZS), respectively.
The PA parameters have been inferred from the dc I–V curves
of the device (Figs. 2 and 3), and load/source pull analysis
(Figs. 4 and 5). The values of the design parameters and their sym-
bols are listed in Table 2.
4. Proposed PA design and implementation
The schematic diagram of the proposed PA is shown in Fig. 6. As
mentioned above the LPFs have a significant role in the proposed
Table 1
The optimum impedance values from load/source pull simulation in desired
frequency.
Frequency 2.5 GHz 2.6 GHz 2.7 GHz
Optimum load impedance (ZL) 11–12j 12–14j 11–16j
Optimum source impedance (ZS) 8–13j 7.5–12j 7–11j

3. 1018 M. Hayati, S. Roshani / Int. J. Electron. Commun. (AEÜ) 68 (2014) 1016–1021
Table 2
Power amplifier DC parameters.
Parameter Symbol Value
DC bias current IDC 60 mA
DC supply voltage VS 28 V
DC gate voltage VGG 2.8 V
Optimum load impedance ZL 12–14j
Optimum source impedance ZS 7.5–12j
structure, which used as an impedance transformer. Therefore,
based on the applied transistor it should transfer the 50 to the
desired impedance.
4.1. Low pass filter
The applied LPF is based on a stepped impedance resonator,
which is used in [12]. The LPF is fabricated on the RT/Duroid 5880
substrate (relative permittivity of 2.2, thickness of 0.508 mm and
loss tangent of 0.0009), as shown in Fig. 7. The size of the filter is
only 59.78 mm2 (9.8 mm × 6.1 mm = 0.116 g × 0.0723 g).
The design process of the filter was discussed in [12], and per-
formance of applied LPF is briefly discussed below.
The measurements were carried out on a HP8757A network ana-
lyzer. As shown in Fig. 8, the measured and simulated results are in
good agreement. The transition band is very sharp, approximately
0.07 GHz from 3.6 to 3.67 GHz with corresponding attenuation lev-
els of −3 and −20 dB, respectively.
Fig. 6. Schematic diagram of the proposed power amplifier.
Fig. 7. The photograph of the fabricated LPF.
Fig. 8. The simulated and measured S-parameters of the proposed LPF.
The applied LPF passes the fundamental frequency (2.6 GHz)
and rejects the unwanted harmonics at the higher frequency. The
frequency response of the proposed filter illustrates three transmis-
sion zeros, at 3.7, 4.3, and 6.2 GHz with attenuation levels of −42.38,
−69.78, and −55.5 dB, respectively. These transmission zeros can
provide the high rejection levels for eliminating the desired har-
monics in the proposed power divider.
4.2. Impedance transformer network
Hence, the applied LPF used as the main block of the impedance
transfer networks in the proposed structure, therefore, the input
LPF as the input impedance transformer network should transfer
50 to the 7.5–12j and the output LPF as the output impedance
transformer network should transfer 50 to the 12–14j . There-
fore, the low impedance stubs are added to the proposed LPF, to
realize the impedance transferring of the LPF from 50 to desired
impedances.
4.2.1. Analysis of two sections impedance transformer
Two sections matching line is one of the most useful
transmission-line that is used for impedance matching in a lot of
applications like the power amplifiers. The two sections line is capa-
ble of achieving ideal impedance matching. Fig. 9 demonstrates the
schematic diagram of the input/output matching network of the
proposed PA.
The input impedance Zin of the two-section line shown in Fig. 10
is obtained as follows:
Zx = Zi2
Rs + jZi2 tan(Âi2)
Zi2 + jRs tan(Âi2)
(1)
where Rs is the optimum source impedances at the 2.6 GHz. There-
fore, the input impedance Zin is
Zin = Zi1
Zx + jZi1 tan(Âi1)
Zi1 + jZx tan(Âi1)
(2)
The input impedance is equal to Z0. Equating Zin to Z0 and solving
(2) for Zx, leads to:
Zx = Zi1
Z0 + jZi1 tan(Âi1)
Zi1 + jZ0 tan(Âi1)
(3)
from (1) and (3):
Zi1
Z0 − jZi1 tan(Âi1)
Zi1 − jZ0 tan(Âi1)
= Zi2
Rs + jZi2 tan(Âi2)
Zi2 + jRs tan(Âi2)
(4)
from (4):
Zi1(Z0 − jZi1 tan(Âi1))(Zi2 + jRs tan(Âi2))
= Zi2(Rs + jZi2 tan(Âi2))(Zi1 − jZ0 tan(Âi1)) (5)

4. M. Hayati, S. Roshani / Int. J. Electron. Commun. (AEÜ) 68 (2014) 1016–1021 1019
Fig. 9. Schematic of the power amplifier, with two sections IMN, OMN.
Fig. 10. Two sections IMN.
The real part of Eq. (5) yields:
Z0Zi1Zi2 + Z2
i1
Rs tan(Âi1) tan(Âi2) = Zi1Zi2Rs
+ Z0Zi1Z2
i2
tan(Âi1) tan(Âi2) (6)
While the imaginary part of (5) yields:
Z0Zi1Rs tan(Âi2) + Z0Zi2Rs tan(Âi1) = Z2
i1
Zi2 tan(Âi1) + Zi1Z2
i2
tan(Âi2)
(7)
On the other hand, the output impedance Zout of the two-section
line shown in Fig. 11 is obtained as follows:
Zy = Zo2
RL + jZo2 tan(Âo2)
Zo2 + jRL tan(Âo2)
(8)
where RL is the optimum load impedances at the 2.6 GHz. Therefore,
the output impedance Zout is
Zout = Zo1
Zy + jZo1 tan(Âi1)
Zo1 + jZy tan(Âi1)
(9)
With the similar process followed for input part, the following
equations are obtained:
Z0Zo1Zo2 + Z2
o1RL tan(Âo1) tan(Âo2) = Zo1Zo2RL + Z0Zo1Z2
o2
× tan(Âo1) tan(Âo2) (10)
Fig. 11. Two sections OMN.
Z0Zo1RL tan(Âo2) + Z0Zo2RL tan(Âo1) = Z2
o1Zo2 tan(Âo1)
+ Zo1Z2
o2 tan(Âo2) (11)
To find the values of the input matching stubs in the conven-
tional PA, assuming normalized value Z0 = 1 , Rs = 0.28 in (6) and
(7), also for further simplification assuming Âi1 = Âi2 = Âi. From (6):
0.72Zi2 + 0.28Zi1(tan Âi)
2
− Zi2(tan Âi)
2
− Zi2(tan Âi)
2
= 0 (12)
and from (7):
(Zi1 + Zi2)(Zi1Zi2 − 0.28) = 0 (13)
from (12) and (13):
f1(zi2) = Z3
i2
(tan Âi)
2
− 0.72Z2
i2
− 0.078(tan Âi)
2
= 0 (14)
The value of Zi2 in (14) is obtained from Fig. 12, which in this
figure, Eq. (14) is plotted for different values of Âi.
From Fig. 12, assuming Âi = 130◦, therefore,
Zi2 = 0.682 × 50 = 34.1 and then Zi1 = 0.41 × 50 = 20.5 .
Similarly to find the values of the output matching stubs in the
conventional PA, assuming normalized value Z0 = 1 , RL = 0.37 in
(10) and (11), also for further simplification assuming Âo1 = Âo2 = Âo.
From (10):
0.63Zo2 + 0.37Zo1(tan Âo)
2
− Zo2(tan Âo)
2
= 0 (15)
and from (11):
(Zo1 + Zo2)(Zo1Zo2 − 0.37) = 0 (16)
From (15) and (16):
f2(Zo2) = Z3
o2(tan Âo)
2
− 0.63Z2
o2 − 0.137(tan Âo)
2
= 0 (17)
The value of Zo2 in (14) is obtained from Fig. 13, which in this
figure, Eq. (17) is plotted for different values of Âo.
Fig. 12. f1(Zi2) for different value of Âi.

5. 1020 M. Hayati, S. Roshani / Int. J. Electron. Commun. (AEÜ) 68 (2014) 1016–1021
Fig. 13. f2(Zo2) for different value of Âo.
From Fig. 13, assuming Âo = 120◦, therefore, Zo2 = 0.6 × 50 = 30 ,
so Zo1 = 0.61 × 50 = 30.5 .
In the input matching network of the proposed PA by replacing
the LPF instead of one stubs (Zi1, Âi1), the values of the another stub
Zi2, Âi2 and Zi1, Âi1 are obtained from (6) and (7) as follows:
Zi1 = 32.4 , Âi1 = 87◦, Zi2 = 8.3 , Âi2 = 23◦, which these are the
dimensions of the proposed input matching network, that is
shown in Fig. 15.
On the other hand, in the output matching network of the pro-
posed PA, replacing the LPF instead of one stub (Zo2, Âo2), the values
of the another stub (Zo2, Âo2 and Zo1, Âo1) are obtained from (10)
and (11) as follow:
Zo1 = 33.84 , Âo1 = 88◦, Zo2 = 8.4 , Âo2 = 22◦, which these are the
dimensions of the proposed output matching network, that is
shown in Fig. 14.
As seen in Fig. 14, at 2.6 GHz the output impedance trans-
former network transfer the 50 to the 12–14j (ZL), which this
impedance is the desired load impedance that is obtained previ-
ously by load pull analysis.
As seen in Fig. 15, at 2.6 GHz the input impedance trans-
former network transfer the 50 to the 7.5–12j (Zin), which
this impedance is the desired source impedance that is obtained
previously by source pull analysis (ZS).
Fig. 14. Simulated results of the proposed OMN.
Fig. 15. Simulated results of the proposed IMN.
Fig. 16. Schematic of the (a) conventional power amplifier, with IMN, OMN and (b)
proposed PA integrated with two LPFs.
4.3. PA implementation and measurements
The structure of the conventional PA and the proposed PA inte-
grated with two LPFs are shown in Fig. 16.
In the conventional PA, IMN and OMN occupy large size and
in the proposed PA applied LPFs lead to ultra size reduction and
efficiency improvement of the proposed PA.
The fabricated power amplifier on a RT/Duroid 5880 substrate
(relative permittivity of 2.2, thickness of 0.508 mm, and loss tangent
of 0.0009) is shown Fig. 17. The measurements were carried out on a
HP8757A network analyzer, and simulations were performed with
the Advanced Design System (ADS) software (Agilent Technologies,
Santa Clara, CA).
Fig. 17. The photograph of the fabricated PA.

6. M. Hayati, S. Roshani / Int. J. Electron. Commun. (AEÜ) 68 (2014) 1016–1021 1021
Fig. 18. Measured and simulated S-parameters of the proposed PA.
Fig. 19. Output power, gain and PAE of the proposed and the conventional PA at
2.6 GHz.
As seen in Fig. 16, the size of the conventional and proposed
PA is about 2070 mm2 (115 mm × 18 mm = 1.136 g × 0.21 g) and
990 mm2 (55 mm × 18 mm = 0.65 g × 0.21 g), respectively, which
presents 52% size reduction in the PA design.
4.4. Measurement results
The proposed power amplifier is characterized by small-signal
and large-signal measurements to evaluate its performance.
4.4.1. Small-signal measurements
The proposed PA is characterized in small signal conditions
(S-parameters) to verify its frequency behavior. A drain bias of
VDD = 28 V is used. The amplifier is biased in class AB (quiescent
drain current of 60 mA and gate voltage of 2.8 V). The simulated
and measured S-parameters are depicted in Fig. 18, which show the
good agreements between simulated and measured results. As seen
in Fig. 18, the small signal gain is 15 dB at the working frequency
(2.6 GHz). The proposed PA suppresses the 2nd–6th harmonics with
high level of attenuation. The measured result shows that, the inser-
tion loss of the 2nd, 3rd, 4th, 5th and 6th harmonics is better than
85, 99, 95, 70 and 96 dB, respectively.
4.4.2. Large-signal measurements
Large-signal continuous wave (CW) measurements are per-
formed to evaluate the power amplifier under steady-state
conditions. The biasing, as same as the small signal measurement is
Table 3
Performance summary of the proposed power divider and previous works.
Ref. PAE Size reduction Harmonic suppression (dB)
2nd 3rd 4th 5th 6th
[1] 46.3% 21% 35 – – – –
[8] 45% – 45 45 – – –
[9] 61.4% – 33 15 20 – –
This work 47% 52% 85 99 95 70 96
used in the large signal measurement. The measured and simulated
results of the fabricated circuits using LPFs and the conventional
matching method are demonstrated in Fig. 19. As shown in Fig. 19,
the maximum PAE at 2.6 GHz is 47%, which shows 6% improvement,
compared to the conventional PA and the gain of the proposed PA
is about 15 dB, which confirms the small signal gain.
The comparison of the proposed PA with previous approaches
matched by filters in output is summarized in Table 3. The results
show excellent dimension reduction with superior harmonics sup-
pressions compared to the reported works. It should be mentioned
that the PAs in [9] is class F thus the PAE is higher.
5. Conclusion
A power amplifier (PA) with a novel technique for size reduction
and nth harmonics suppression has been proposed and imple-
mented. The proposed PA is integrated by two low pass filters
(LPFs). The applied LPFs are used to suppress the harmonics while
work as input and output matching networks. The proposed PA
suppresses 2nd–6th harmonics with high level of attenuation. In
the proposed method, the conventional input and output match-
ing networks are eliminated, which result in 52% size reduction
as compared with the conventional PA. For verification, the pro-
posed power amplifier is fabricated and tested. A good agreement
between the measured and simulated results is achieved.
References
[1] Li YCh, Wu KCh, Xue Q. Power amplifier integrated with bandpass filter for
long term evolution application. IEEE Microw Wireless Compon Lett 2013;23:
424–6.
[2] Ramzi D, Ghannouchi FM. Digital Doherty amplifier with enhanced efficiency
and extended range. IEEE Trans Microw Theory Tech 2011;59:2898–909.
[3] Raab F. Intermodulation distortion in Kahn-technique transmitters. IEEE Trans
Microw Theory Tech 1996;44:2273–8.
[4] Raab F, Asbeck P, Cripps S, Kenington P, Popovic Z, Pothecary N, et al. Power
amplifiers and transmitters for RF and microwave. IEEE Trans Microw Theory
Tech 2002;50:814–26.
[5] Doherty W. A new high efficiency power amplifier for modulated waves. Proc
Inst Radio Engrs 1936;24:1163–82.
[6] Hayati M, Roshani S. A broadband Doherty power amplifier with harmonic
suppression. AEU-Int J Electron Commun 2014;68:406–12.
[7] Nemati H, Fager C, Gustavsson U, Jos R, Zirath H. Design of varactor-based tun-
able matching networks for dynamic load modulation of high power amplifiers.
IEEE Trans Microw Theory Tech 2009;57:1110–8.
[8] Wang Z, Park C. Novel substrate integrated waveguide (SIW)-based power
amplifier using SIW-based filter to suppress up to the fourth harmonic. In:
Asia-Pacific Microw Conf. 2012. p. 830–2.
[9] Wang Z, Gao S, Nasri S, Park C. High power added efficiency power amplifier
with harmonic controlled by UWB filter with notched band at 6.42 GHz. In:
Wireless Microw Tech Conf. 2011. p. 1–4.
[10] Sigg A, Heck S, Brackle A, Berroth M. High efficiency GaN current-mode class-D
amplifier at 2.6 GHz using pure differential transmission line filters. Electron
Lett 2013;49:47–9.
[11] Liao HY, Pan MW, Chiou HK. Fully-integrated CMOS class-E power amplifier
using broadband and low-loss 1:4 transmission-line transformer. Electron Lett
2010;46:1490–1.
[12] Hayati M, Roshani S, Roshani S, Shama F. A novel miniaturized Wilkinson
power divider with nth harmonic suppression. J Electromagn Waves Appl
2013;27:726–35.