1. An Overview of CMOS-MEMS Resonant RF Mixer-Filters and
Bandpass Filters
Tianhao Li1
Abstract— This paper investigates the advantages of
high-Q MEMS resonator for RF bandpass filtering with
tuning ability, and evaluates the possibility of replacing
off-chip bulk passive filters with single-chip CMOS-
MEMS transceiver. This paper also examines the poten-
tial application of CMOS-MEMS resonant mixer-filters
in down-converting RF signals into IF band after filtering
out mixing harmonics. Specific design examples from
research papers are presented and analyzed.
Keywords: resonator, CMOS-MEMS, filter-mixers,
bandpass filters
I. INTRODUCTION
Current radio receivers use ceramic or surface acous-
tic wave filters for image rejection and channel se-
lection. These discrete components limit the miniatur-
ization, and increase the manufacturing cost of radio-
frequency electronics due to the cost associated with
additional assembly and packaging. Even though those
passive filters are capable of achieving quality factors
from 500 to 10,000 needed for RF and IF filtering and
frequency selections, they are all off-chip components
which must interface with transistor-level integrated
circuits at the board-level. This consumes a sizable
space on the electronics board which has become a
valuable resource in modern cellular handsets.
There are four major motivations for CMOS-MEMS
integration aiming at replacing passive filters afore-
mentioned: enabling high volume production, solving
interconnect bottlenecks, improving performance, and
reducing per unit cost. Current radio-frequency systems
require a high-level of integration [1]. For example, the
recent advances in consumer electronics can be related
to the integration of high performance RF electronics
using CMOS based processes, as the data throughput
rate requirement spikes. Single-chip RF transceivers
has been found in many Samsung and Apple hand-
sets[2]. Combined with the constant revolution in
wireless communication standards (3G, 4G-LTE, 5G),
1
T. Li is a graduate student majoring in Electrical Engineering,
at the Georgia Institute of Technology, Atlanta, GA, 30332, USA.
tianhao@gatech.edu
the future need for highly-integrated RF transceiver
with high quality factor keep increases. Researchers
have now produced integrated micro-mechanical cir-
cuits with sufficient Q, thermal stability, and aging sta-
bility [3]. With such advantages, vibrating RF MEMS
devices are perceived more as circuit building blocks
than stand-along devices. Recent micro-mechanical res-
onators can produce Q’s that are greater than 10,000 at
GHz range, while maintain an impressively low thermal
stability of 18ppm over 27-107◦C. Such high speci-
fications promise applications in many fields. In RF
domain, MEMS devices can be used for reconfigurable
channel-selecting filter banks, ultrastable reconfigurable
oscillators, and frequency translators. The inclusion
of MEMS devices as circuit building blocks enables
circuit designer to trade high quality factor for lower
power consumption and better system robustness [3].
This paper first present modern CMOS-MEMS fab-
rication technology with an emphasis on the operation
mechanisms of RF MEMS devices under advanced
CMOS processes. Then the paper further investigates
the design of CMOS-MEMS RF bandpass filter with
tuning ability [4], and the design approaches and tech-
niques for mixer-filter which performs mixing oper-
ation between RF signals and LO signals using dif-
ferential cantilever resonators[1] and clamped-clamped
beam resonators. The advantages and disadvantages
of CMOS-MEMS co-design comparing to CMOS-only
solution are discussed in the end of each section.
II. MEMS TECHNOLOGY
CMOS based MEMS processes can be generally
categorized into pre-CMOS, intra-CMOS, and post-
CMOS micro-machining three types. Micro-machining
must result in a clean and flat silicon surface, thus no
contaminants are allowed in the process [5].
A. Modular Fabrication
One of the popular fabrication approach is modular
fabrication. A modular process means that the fabri-
cation approach can readily adapt to the changes in a

2. Fig. 1. Cross-sections (a) immediately before and (b) after release
of a surface-micro-machining process done directly over CMOS [6]
given module. In CMOS-MEMS context, one example
of these changes is the variation channel length of a
CMOS transistor. Modular fabrication deposits ploy-
silicon layers on to the wafer, and conducts unin-
terrupted CMOS processing [6]. The fabrication then
performs a hydrofluoric acid etching of the sacrificial
oxide to finish final MEMS process module.
Fig. 1 shows cross-sections describing a poly-silicon
surface micro-machining process done directly over
silicon CMOS circuits in a modular fashion, where
process steps for the CMOS transistor and MEMS are
kept separate and in distinct modules.[6] The process
deposits and patterns films above a finished CMOS
circuit with typical CMOS foundry equipment and
facilities. Once the oxide sacrificial layer is removed
from the release etch barrier, the structural poly-silicon
layer is free-standing, and can be used as a beam-
resonator.
B. RF MEMS Process
Certain CMOS-MEMS sensor applications in the
past use above 0.5µm processes. RF MEMS applica-
tions typically require advanced CMOS processes in
order to fabricate high unity-gain bandwidth transistors
due to the high-frequency nature of RF signals. Under
advanced CMOS processes, internal stress gradients,
caused by multi-layer fabrication, are reduced with
better stress matching in 200mm water [1]. Additional
layers of metal create thicker beams which provide
better out-of-plane stiffness. With less stress gradients
and thicker beams, the out-of-plane curl is effectively
reduced. One additional advantage the advanced CMOS
process can provide is the reduction of minimum metal
spacing. By offsetting the embedded metal layers, the
Fig. 2. Cross-sectional schematic of the idealized three-port
clamped-clamped beam micro-mechanical end resonator.
electrode gaps of resonant MEMS devices can be fur-
ther reduced, which lead to even higher quality factor
[8].
C. Clamped-Clamped Beam Resonator
Clamped-clamped beam resonators are base compo-
nents of micro-mechanical filters. A single clamped-
clamped resonator can provide a second-order fre-
quency response. Use of several resonators of this type
in coupled-resonator architecture allows to achieve a
high-order filter, satisfying complex passband specifi-
cations in modern RF communication systems.
The second-order frequency response nature of a
clamped-clamped beam is not only useful in bandpass
filter design, but can be exploited by a mixer-filter as
well. As shown in Fig 2, when signals are applied
to terminals e and b, the difference voltage (Ve - Vb)
contains the sum or difference of the local oscillator
signal VLO and the RF input signal VRF [9]. In addi-
tion, (Ve - Vb) must also contain a dc-bias voltage VP
for filter termination purposes. In a commercial 0.35-
µm CMOS technology, clamped-clamped beams can
be defined and fabricated with the polysilicon module
layers. Since the beam resonator can be intentionally
chose to be built with the bottom polysilicon layer,
once the chip is received from foundry, with the already
mechanized MEMS resonators, the silicon oxide that
surrounds the beam resonators can be removed by
using maskless wet etching techniques, offering a fully
CMOS-compatible fabrication process [10][11]. Under
this approach, resonator-driver gaps down to 40 nm can
be obtained.
III. TUNABLE BANDPASS FILTER DESIGN
Using the on-chip clamped-clamped resonator afore-
mentioned, a tunable bandpass filter can be designed by
leveraging the second-order frequency response of the

3. Fig. 3. System conceptual scheme for the filter and readout circuit
is shown at the top. SEM images of the MEMS resonators, and an
electrical block diagram of the differential amplifier is shown at the
bottom.
beam. In Fig. 3, the MEMS resonators are integrated
monolithically with an on-chip differential amplifier.
The design shown in Fig. 3 utilizes a parallel filtering
structure. The input driver takes in the RF signals
of interest which need to be filtered. To match the
resonance frequency of the clamped-clamped beam
with the passband center frequency of the incoming
RF signals, the resonator needs to be tuned through
accessible measures. This is achieved by means of DC
bias voltages VDC1 and VDC2 at the two sides of the
parallel structure. This mechanism provide a tunable
filter in terms of center frequency and bandwidth. After
the filtering stage, the output capacitive currents are
combined using a readout differential amplifier.
In the design of [4], a resonator-driver gap of 150
nm is defined to ensure the linearity of of resonator
behavior for a wide range of DC bias voltages. CC-
beam resonators are designed to operate at a funda-
mental resonance frequency around 22MHz.
Within the same design, the readout circuit is also
implemented on-chip. In the work presented in [4],
the readout circuit includes two amplification stages.
The resonator capacitive currents are detected with
transimpedance amplifiers (first amplification stage),
one for each resonator. In the second amplification
stage, the two voltage signals obtained from the first
stage are differentially amplified using a common-
source amplifier. In the work presented in [4], a dif-
ferential gain of 87dB is achieved at 20MHz with
Common-Mode-Rejection-Ratio of 40 dB. Considering
the 50-ohm loading of test instrumentation, a source-
follower buffer is integrated on-chip in that design.
Each clamped-clamped can be independently measured
Fig. 4. Resonator characterization with a bias voltage of VDC1 =
5 V in air and vacuum condition. VDC2 is controlled at 0 V .
by switching off the other resonator in the parallel
structure. To shown an example of the high qualify
factor characteristic of an on-chip MEMS clamped-
clamped resonator, Fig. 4. presents the qualify factor
measurements of such a resonator with respect to
resonant frequency.
As shown in Fig. 4, the quality factor value of 220
measured in air condition can be further improved to
4400 in vacuum condition (P = 20µbar) due to the
lowering of air damping. The relationship between the
resonant frequency of the device and the bias voltage
can be represented in equation (1):
fresonance ∝
1
V 2
DC
(1)
Specifically the clamped-clamped beam shown in Fig.
3 has a tuning range of 400 kHz within 20-V variation.
One additional benefit of a clamped-clamped beam
resonant filter is that different combinations of VDC-
value can control the filter bandwidth, which promise
designers another degree of freedom. In Fig. 5, the
measured bandpass-filter frequency response of the
design in [4] is shown. The filter bandwidth can be
widened by increase the difference between the bias
voltages applied to the resonator. It is observed that by
means of parallel filtering, the stop-band (SB) rejection
is increased by more than 10 dB comparing to the case
with only one resonator.
Even though CMOS-MEMS bandpass filter has high
Q and provides tuning ability, the operating frequency
is still in the megahertz range, while similar CMOS RF
circuits can operate in the gigahertz range. Furthermore,
the need for maskless wet etching add additional cost
to the chip, even though the MEMS process is nearly

4. Fig. 5. Frequency response of the CMOS-MEMS filter[4] for
different bias voltages.
fully compatible with CMOS process. However, future
MEMS bandpass filters have the potential to operate
in the gigahertz range by improving the resonator
topology and adapting the circuit design.
IV. MIXER-FILTER DESIGN
Recent advances in MEMS resonant mixer-filters
produced devices that can down-convert and filter sig-
nals with frequencies as high as 3.2 GHz, which covers
a bandwidth up to 2.4 GHz WIFI band. One major
benefit of CMOS-MEMS mixer is the elimination of
mixing feedthrough at low RF frequencies. Mixing
feedthrough has been identified as a major problem
in RF design. The voltage amplitude of LO signal is
usually on the same order with supply voltage VDD, in
order to fully turn on and turn off the LO transistors.
Therefore in a direct downconversion RX topology,
when LO signals propagate through the parasitic path
in the transistor to RX antenna or to IF band, the
noise figure could be significantly deteriorated. Fig. 6
describes the RF-IF, LO-RF, and LO-IF three types of
mixing feedthrough on a block level. LO feedthrough
require CMOS RFIC designers to carefully balance
the mixer structure using double-balanced structure or
other more complex techniques [12]. In CMOS design,
LO-RF feedthrough can be minimized using circuit
techniques such as match the input transistors of a
double-balanced Gilbert cell mixer [12]. Under this
scheme, the parasitic capacitance of two input transis-
tors can be matched, and then the feedthrough signals
can be canceled. However, these techniques increase
the circuit complexity, and require perfect matching.
Fig. 6. Three types of mixing feedthrough during a mixing
operation between RF and LO signals.
Fig. 7. Schematic of the CMOS-MEMS resonant mixer-filter in
[1]. Embedded electrodes and SEM close-up are shown as examples
of multi-layer CMOS metal used in electrode design.
MEMS mixer-filters exploit the nonlinearity of elec-
trostatic force with drive voltage applied on the res-
onators. Mixer-filters down-convert GHz RF input sig-
nals to excite MHz mechanical resonance for inter-
mediate frequency filtering. Such a device further ca-
pacitively transduce the mechanical displacement into
an electrical IF output. Mixing and filtering functions
can be achieved simultaneously as the RF signals
are passing through the MEMS resonator [1]. Fig .7
presents the schematic of a resonant MEMS mixer. The
RF and LO signals are applied across a drive gap of g.
An electrostatic static force with a relationship shown
in equation 2 is exerted on the MEMS structure. The
product term proves that mixing operation can occur in
such a device because the difference between RF and
LO frequencies is generated.
Felectrostatic ∝ (VRF − V LO)2
= V 2
RF + 2VRF VLO + V 2
LO (2)
At resonant frequency, the qualify factor Q of the
resonator amplifies the displacement due to the electro-
static force. In [1], the parasitic capacitance converts the

5. Fig. 8. SEM image of the differential cantilever resonators with
the schematic for mixer-filter.
ac current i to voltage at the output before it is further
amplified by a on-chip pre-amplifier. The complete
equation of Vout is given by:
Vout =
A
Cp
i · dt
=
A
Cp
(ε0teLe)2
2Bg4
VP VLOVRF
1
ωrsin(ωrt)
(3)
where A is the amplifier voltage gain, B is the damping
factor, ωr is the resonant frequency, with other terms
defined in Fig. 7. At the end of the cantilever, the
square frame reduces the feedthrough between VRF and
VLO by distancing the respective electrodes. Evidently,
the split-conductor design is preferred over the single-
conductor as the LO signal appears solely on the
drive electrode and cannot feedthrough into the output
electrode.
The differential resonator structure in Fig. 8 enables
the rejection of common-mode mixing feedthrough.
The LO and RF signals are applied to both resonators
across two electrostatic gaps of 1.3 µm that are routed
under the top-metal mask. The differential amplifier is
implemented on-chip to convert the motional currents
to voltages with an input capacitance.
Fig. 9 shows the output spectrum of the resonant
mixer-filter at around fIF , while the fLO stepped from
10 MHz to 400 MHz. By tuning the DC bias of res-
onators, the spring-softening effect can compensate the
frequency difference at the output of the two resonators
due to manufacturing variations. However, when fLO
increases beyond 1 GHz, mixing feedthrough starts to
increase, as its amplitude rises outside of the resonance
which happens at 435kHz. The mixing feedthrough can
Fig. 9. Mixer output spectrum for fLO =10MHz-400MHz, with
f0=435kHz resonators.
Fig. 10. (a) Mixing output when fLO = 1.8 GHz; (b)fLO = 3.2
GHz.
be eliminated by measuring the feedthrough separately
without dc polarizing voltages, then subtract it out.
One major difference between CMOS Gilbert cell
mixer and this mixer-filter is that while CMOS Gilbert
cell mixer provides a conversion gain at the fundamen-
tal frequency component given as:
C.G. =
Vout
VRF
=
2
π
gmRL (4)
the MEMS mixer design in [1] endures a conversion
loss of 50dB at the mixing frequency. This difference
can be reduced using the preamp which is implemented
on-chip.
While differential cantilever resonators are effective
candidates for mixer-filter design, the clamped-clamped
beam resonator aforementioned in MEMS technology
section and MEMS tunable-bandpass filter section can
also be used for mixing operations at RF frequencies.
Fig. 11 presents a mixer-filter design with clamped-
clamped beam resonators. The electrostatic force on
the beam has a similar relationship with respect to
bias voltage as equation (2). By the taking the partial

6. Fig. 11. (a) Simplified block diagram of a wireless receiver, indicating (with shading) the components replaceable by mixer-filter devices.
(b) Schematic diagram of the described mechanical mixer-filter, depicting the bias and excitation scheme needed for downconversion. (c)
Equivalent block diagram of the mixer-filter scheme.
derivative of the energy E stored on the beam with
respect to the beam displacement z, the force Fz is
then given as:
Fz =
∂E
∂Z
=
∂
∂Z
[
1
2
(C1)(Ve − Vb)2
] (5)
Similarly, equation (5) yields a product term which
enables RF mixing operation.
CMOS-MEMS mixer-filter provides high-Q filtering
at the mixer output for removing higher-order har-
monics. It also features mixing feedthrough elimina-
tion at low LO frequencies. Nevertheless, the mixer-
filter suffers from large conversion loss which requires
the addition of on-chip preamp. It also suffers from
feedthrough at gigahertz frequency due to capacitive
coupling between RF input and mixer currents. Even
though CMOS-MEMS mixer-filter operating at giga-
hertz still need to be further developed in order to meet
commercial communication electronics standards, the
prospects are promising for integrated MEMS radios
in the future.

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