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1. IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013 4419
A Study of Low-Profile, Broadside Radiation,
Efficient, Electrically Small Antennas Based on
Complementary Split Ring Resonators
Ming-Chun Tang, Member, IEEE, and Richard W. Ziolkowski, Fellow, IEEE
Abstract—The designs and performance characteristics of sev-
eral electrically small antennas based on complementary split ring
resonators (CSRRs) are reported. A coaxial-fed monopole is first
integrated with a CSRR that is cut from a grounded finite copper
disc. The presence of the electrically small CSRR element facili-
tates a nearly complete impedance match to the source, a nearly
broadside radiation pattern, and a high radiation efficiency. The
addition of a circular top-hat to the monopole then achieves an
ultra-low profile design and an improved broadside
pattern, while maintaining all other desirable features. Finally, to
enrich their potential usefulness, two additional enhancements of
these designs were accomplished. One is a further miniaturization
that is achieved by introducing a more complex
CSRR element, while maintaining a high, 82%, radiation effi-
ciency. The second is a further enhancement of the directivity and
front-to-back ratio through the introduction of a slot-modified
parasitic disc, while maintaining the original impedance matching,
low-profile and electrically small properties. These designs were
consummated and their performance characteristics evaluated
with the frequency domain ANSYS-ANSOFT High Frequency
Structure Simulator (HFSS) and were confirmed independently
using the time domain CST Microwave Studio (MWS) simulator.
A prototype of the basic system was fabricated and tested; the
agreement between the simulated and measured results validates
the design principles.
Index Terms—Antenna directivity, antennas, complementary
split ring resonator, electrically small antennas, metamaterials.
I. INTRODUCTION
F OR nearly a decade, the split ring resonator (SRR) and
its counterpart, the complementary split ring resonator
(CSRR), have been considered as unit cells in metamaterial
designs. They have attracted much attention and have been
Manuscript received August 06, 2012; revised May 08, 2013; accepted June
03, 2013. Date of publication July 09, 2013; date of current version August 30,
2013. This work was supported in part by the Graduate School of the University
of Electronic Science and Technology of China and in part by NSF contract
number ECCS-1126572.
M.-C. Tang was with the Institute of Applied Physics, University of Elec-
tronic Science and Technology of China, Chengdu 610054, China and also with
the Department of Electrical and Computer Engineering, University of Ari-
zona, Tucson, AZ 85721 USA. He is now with the College of Communication
Engineering, Chongqing University, Chongqing, 400044, China (e-mail: tang-
mingchunuestc@126.com).
R. W. Ziolkowski is with the Department of Electrical and Computer
Engineering, University of Arizona, Tucson, AZ 85721 USA (e-mail: zi-
olkowski@ece.arizona.edu).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2013.2267711
studied and applied extensively because of their many attrac-
tive performance characteristics [1]–[3]. One of the important
applications of SRRs and CSRRs has been in the design of
microwave circuits [4]. Another has been for the design of
small antennas, mainly due to the advantage of their sub-wave-
length resonances [5]–[15]. They have been used for artificial
magnetic conductors (AMCs) to achieve low-profile antennas
[5] and as magnetic loadings to achieve larger bandwidths [6],
[9]. They have been used to obtain electrically small antenna
designs [7] and the miniaturization of known designs [8], [13].
They have been used to realize notched filters in UWB antennas
[12], [15], to increase the number of resonance frequencies in a
single antenna [10], [13], and to achieve impedance matching
[11].
While the SRR and CSRR strategies have provided for an-
tenna miniaturization, one witnesses certain drawbacks in the
resulting designs, which restricts their widespread engineering
application. For instance, the radiation efficiency within the
10 dB impedance bandwidth may be quite low, leading to
small realized gain values [9]. Their fabrication may become
quite cumbersome [6], [8], their inclusion may limit the actual
reduction in size [10], [13], or their materials may decrease the
overall radiation efficiency [14].
In this paper, a CSRR element is introduced in a fi-
nite grounded disc and is then integrated with a traditional
monopole antenna in Section II. The performance character-
istics of the resulting electrically small antenna (ESA, i.e.,
, being the radius of the smallest sphere that com-
pletely encloses the antenna at the operational frequency, ,
and is the free space wave number) are
investigated. It is shown how this combination can produce
a radiation pattern whose maximum is along the axis of the
monopole rather than broadside to it. Next, in Section III, an
ultra-low profile version of this ESA is accomplished
by loading the monopole with a circular top-hat. The perfor-
mance characteristics of this design are parametrically studied
in a comprehensive manner in Section IV. It is demonstrated
that these miniaturized antennas have high radiation efficien-
cies while being impedance matched to the source without any
matching network. Efforts to enhance the usefulness of these
designs for wireless applications are presented in Section V.
In particular, a more complex CSRR element is introduced to
further miniaturize the antenna , while maintaining
its radiation efficiency around 82%. Additionally, a slot-mod-
ified parasitic conducting disc is introduced to enhance the
directivity of the resulting ESA while significantly
0018-926X © 2013 IEEE

2. 4420 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013
increasing its front-to-back ratio (FTBR), increasing its radi-
ation efficiency above 90%, and maintaining its low-profile
nature . As described in Section VI, a prototype
antenna was fabricated and experiments were performed to
validate the basic design principles reported in the previous
sections. It will be demonstrated that the measured results are
in good agreement with their simulated values. Finally, some
conclusions are drawn in Section VII.
We note that in this paper, all of the metallic elements in the
antenna designs are chosen to be copper with its known mate-
rial parameters: , and bulk conduc-
tivity . The numerical simulations
and their optimizations were carried out first using the frequency
domain ANSYS/ANSOFT High Frequency Structure Simulator
(HFSS) [16] and then confirmed with the time domain CST Mi-
crowave Simulator (MWS) tool set [17]. The design frequency
was targeted at 300 MHz, i.e., a 1000 mm free-space wave-
length, simply to facilitate the discussion.
II. DESIGN OF CSRR-AUGMENTED MONOPOLE
Consider first a monopole antenna oriented perpendicular to
a grounded circular copper disc (finite ground plane) and coax-
ially-fed through it. This configuration is shown Fig. 1(a). It is
well known that this type antenna can operate in its fundamental
dipole mode when the monopole length is around one-quarter
wavelength [18]. At this fundamental resonance fre-
quency, the current distributions driven on the monopole and
induced on the top of the disc are shown in Figs. 1(a) and (b),
respectively. In Fig. 1(b), the current distribution on the disc,
which is taken to lie in the xy-plane, is pointing radially out-
ward from the monopole, which is taken to be along the -axis.
In Fig. 1(a), the current on the monopole is along the -axis.
Because it is grounded, the currents on the bottom side of the
disc will be pointing radially inward, leading to a return cur-
rent in the direction along the outer wall of the coax. These
current contributions lead to the well-known dipole (doughnut)
radiation pattern that has its maximum (minimum) orthogonal
(parallel) to the monopole direction [18], [19].
In our original attempt to produce a lower resonance fre-
quency and to miniaturize the antenna, different types of CSRR
structures were cut from the disc. By effectively loading the
grounded disc with these CSRRs, a lower frequency resonance
can be introduced that has the monopole-induced current on the
disc concentrated along the edges of the CSRR element. Sev-
eral of these CSRR-modified discs and the current distributions
induced on them by the monopole are illustrated in Fig. 1(b). It
was found unexpectedly that the currents on the disc dominate
the radiation performance at the lower resonance frequency.
To explain the radiation mechanisms of this class of CSRR-
based antennas, we emphasize the single-CSRR design shown
in Fig. 2. The grounded disc and the CSRR structure have the
same center. A 50 coax feed-line is assumed and was included
in the HFSS model beneath the disc as can be seen in Fig. 2. The
coax design included a Teflon sleeve with a relative permittivity,
, and a loss tangent, tan , i.e., Teflon filled the
space between the inner and outer copper walls of the coax and
extended all the way along the center conductor to end flush with
Fig. 1. Coaxially-fed monopole augmented with a finite, grounded, CSRR-
modified circular copper disc. (a) Three dimensional (3D) view when the disc
is not modified, and (b) two dimensional (2D) views of the current distribu-
tion induced by the monopole on the disc when it incorporates: (I) No CSRR;
(II) a single CSRR; and (III) two, (IV) three, (V) four, and (VI) five symmetric
CSRRs.
the upper surface of the disc. The center conductor of the coax
was itself extended above the disc to form the monopole. The
grounded disc was taken to be 0.5 mm thick. The dimensions of
the 300 MHz design are given in Fig. 2. They were adjusted
to achieve nearly complete impedance matching to the 50
source, i.e., a zero input reactance and an input resistance equal
to 50 at 300 MHz.
The HFSS-simulated performance characteristics of this
CSRR-based antenna are given in Figs. 3 and 4. For
comparison, the corresponding results for the same sized
disc-loaded monopole antenna shown in Fig. 1(a) are also

3. TANG AND ZIOLKOWSKI: A STUDY OF LOW-PROFILE, BROADSIDE RADIATION, EFFICIENT, ELECTRICALLY SMALL ANTENNAS 4421
Fig. 2. 3D view of the coax-fed monopole antenna augmented with the single
CSRR-modified grounded circular disc designed to operate near 300 MHz.
Its design dimensions have the values in mm: , ,
, , , and .
Fig. 3. The simulated values for the antennas whose grounded discs have
or lack a single CSRR element. The subplots show the current distributions on
the upper surface of the grounded disc for the CSRR-based antenna at its two
lowest resonance frequencies.
given. The appearance of the desired low frequency res-
onance is clearly seen in Fig. 3 at 300 MHz where the
minimum value . This resonance fre-
quency is more than a factor of two lower than the reference
monopole fundamental resonance frequency, 671.4 MHz,
where . The 10 dB impedance band-
width at 300 MHz is 2.334 MHz, which means the fractional
bandwidth is 0.78%. At 300 MHz, the value of this antenna
is 0.854 and its radiation efficiency is 98.11%. Despite its
electrically small size, it radiates very efficiently and the
values demonstrate that the CSRR-based antenna is nearly
completely matched to the 50 source. We note that there is a
second resonance, similar in nature to the reference case one.
While it is located at a slightly higher frequency, it shows a
better match to the source.
The simulated current distributions at the frequen-
cies are also shown in Fig. 3. At the lower 300 MHz resonance
frequency, the majority of the current on the grounded disc is
concentrated in the CSRR gap and along its edges near the gap,
as was already shown in Fig. 1(b). The resulting radiation pat-
tern is given in Fig. 4(a). Its behavior is similar to a dipole
asymmetrically positioned with respect to the origin and ori-
ented along the x-axis, i.e., similar to “an asymmetrical dipole”
Fig. 4. The E- (zx-plane) and H-plane (yz-plane) directivity patterns for the
CSRR-based antenna at 300 MHz exhibit an asymmetrical dipole behavior (a).
The corresponding zx-plane and yz-plane directivity patterns at 628.3 MHz
for the CSRR-based monopole and at 671.4 MHz for the reference monopole
having no CSRR in its grounded disc (b).
with horizontal polarization along the x-axis [19]. We note that
the edge-length from the middle point of the inside edge of the
CSRR element to the middle point of the outside edge at 300
MHz is , where
(1)
i.e., it is nearly in length. Note that there are two distinct
features of the radiation pattern at the 300 MHz resonance fre-
quency given in Fig. 4(a). The peak radiation direction is
away from z-axis in the zx-plane (E-plane), instead of being
strictly in the direction. Concurrently, the two nulls in the
E-plane occur at 80.6 and , instead of strictly being
at and ( direction), respectively. Although the
pattern is tilted from broadside, the broadside directivity (in
direction) is still high at 3.60 dB.
At the higher resonance frequency, 671.4 MHz, the CSRR-
based antenna operates, like the reference case at its resonance
frequency, 628.3 MHz, in a fundamental monopole mode. The
vertically polarized monopole-like radiation patterns associated
with these resonances are given in Fig. 4(b). The radial current
distribution on the grounded disc shown in Fig. 3 further con-
firms this monopole behavior. Comparing the current distribu-
tions associated with these two resonances, the CSRR structure
clearly created a new current pathway on the grounded disc,
leading to the pattern being broadside to it.
The effect of the monopole length on the impedance match
in the lower resonance frequency range is illustrated in Fig. 5. It
shows that the input impedance is impacted by the length of the
monopole. Note that the resonance frequency decreases as the
monopole length and, hence, its inductance increases. This be-
havior indicates, as with many other metamaterial-inspired an-
tennas [20], that tuning the shape of the driven element, here the
height , is a simple and effective way to refine the impedance
match.
Returning now to Fig. 1(b), one finds that the CSRR element
with a single gap produces the lowest resonance frequency. One
way to understand this behavior is to remember that the res-
onance wavelength is closely related to the total edge length
of the slot. Smaller slots lead to higher resonance frequencies.
A circuit element view would recall that each gap acts like a
capacitor. More cuts (gaps) lead to more capacitors in series

4. 4422 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013
Fig. 5. Impact of the antenna height on the magnitude of the reflection coeffi-
cient .
and, hence, to an overall decrease in the capacitance. Since the
resonance frequency of such a metamaterial resonator is given
by , where and
are, respectively, the effective inductance and capacitance of
the system, a decrease in leads to a higher resonance fre-
quency.
An important aspect of the current behavior is the location of
the resultant dipole, i.e., the highly localized, large in-phase cur-
rent density in the gap. As noted this causes the tilt in the pattern.
One might expect that multiple cuts could help with un-tilting
the pattern. However, the symmetrical placement of the gaps
with respect to the x-axis about the monopole causes their net
dipole moment to remain along the x-axis, and it is always dis-
placed from the origin simply by the location of the slots. Also
note that one can move the effective horizontal dipole closer to
the origin to un-tilt the pattern by asymmetrically locating the
slot relative to the origin, i.e., by maintaining the slot shape but
shifting the gap closer to the origin. We found, however, that this
caused the vertical currents on the monopole and the opposite
pointing currents in the gap to become too close to one another,
which in turn had a significant negative impact on the radiation
efficiency. Therefore, in the remainder of this paper, all designs
will emphasize CSRR structures with a single gap and whose
centers coincide with the coax center-line.
III. ULTRA-LOW PROFILE DESIGN WITH NEARLY BROADSIDE
RADIATION
As shown in Figs. 2 and 5, the single CSRR-based antenna
had a reasonably long monopole length, i.e., . It
was desired to modify the design to make it and, hence, the en-
tire system low profile. However, because of the behavior shown
in Fig. 5, the height parameter leads only to fine tuning. While
making the monopole thinner increases the inductance which in
turn would allow for a smaller height (increases the capacitance)
to maintain the resonance frequency (net reactance being zero),
the changes are small.
In order to achieve a lower profile design, while maintaining
the resonance frequency and nearly complete impedance
matching, it was found that standard top-hat monopole tech-
niques proved useful. A copper disk was added on the top of the
Fig. 6. Configuration of the top-hat loaded, CSRR-based monopole antenna.
(a) 3D view, (b) top view, and (c) side view. The design parameters have the
values in mm: , , . All other
parameters remain unchanged from those given in Fig. 2.
driven monopole antenna. This configuration is shown in Fig. 6
along with its dimensions. We were able to optimize the radius
of the top-hat and the height of the monopole to maintain the
impedance match. While various combinations are possible,
we report here the very low-profile, 300 MHz CSRR-based
antenna design, whose total height .
The HFSS and CST MWS simulation results for the top-
loaded, CSRR-based antenna are shown in Fig. 7. The HFSS
(CST) predicted values given in Fig. 7(a) indicate that
the resonance frequency is 300.0 (299.3) MHz with
( 28.61) dB and its 10 dB impedance bandwidth is
2.40 (2.77) MHz. Fig. 7(b) shows the E- and H-plane directivity
patterns. At the resonance frequency, , the direc-
tivity in the direction is 3.91 dB (3.63 dB), and the radiation
efficiency is 98.03% (94.47%). Moreover, the peak radiation
direction is 2.5 (2.3 ) away from the -axis in the E-plane,
while two nulls appear at and 90.1 (90.1 )
in the E-plane.
The HFSS-predicted currents on all of the horizontal sur-
faces are shown in Fig. 8. The majority of the driven current
on the top-hat’s top surface is concentrated on the edge above
the CSRR gap. In contrast, it is concentrated near the end of
the monopole, near the center of the disc, on its bottom sur-
face. Stronger radial currents are present on the CSRR-modified
grounded disc near the coax. Nonetheless, the largest currents
appear in the gap of the CSRR element and along its edges near
the gap. Particularly important is the observed fact that the cur-
rents on the tops (and bottoms) of both discs are in-phase. The
presence of the currents on the top of the top-hat disc aids in
un-tilting the radiation patterns. On the other hand, because the
impedance matching is best when the radius of the top-hat disc
and the inner radius of the CSRR element are nearly the same,
the currents on the bottom of the top-hat disc and those in the
gap on the top of the CSRR-modified grounded disc are oppo-
site to each other. Consequently, it is found that decreasing the

5. TANG AND ZIOLKOWSKI: A STUDY OF LOW-PROFILE, BROADSIDE RADIATION, EFFICIENT, ELECTRICALLY SMALL ANTENNAS 4423
Fig. 7. HFSS and CST simulation results of the top-hat loaded, CSRR-based
antenna shown in Fig. 6. (a) values as a function of the frequency, and
(b) E- and H-plane directivity patterns at the resonance frequency: 300 MHz
(HFSS) and at 299.3 MHz (CST).
Fig. 8. Current distributions on the various horizontal surfaces of the top-hat
loaded, CSRR-based antenna. On the (a) top and (b) bottom of the top-hat disc.
On the (c) top and (d) bottom of the CSRR-modified grounded disc.
height further will lead to a significant decrease in the radia-
tion efficiency. We note that the total edge-length of the CSRR
element along the dashed line path shown in Fig. 8(c) and cal-
culated with (1) at 300 MHz is .
Fig. 9. Impact of the height on several design parameters and the resulting
performance characteristics. (a) Radius values necessary to maintain the 300
MHz resonance, and (b) the corresponding values at 300 MHz and
10 dB impedance bandwidth value.
These results demonstrate that the top-hat approach has an
additional benefit, i.e., the broadside nature of the radiation
pattern is improved considerably. This improvement can be
ascribed to two reasons: (1) The length of the monopole has
been reduced significantly, which proportionately decreases
the residual vertical polarization effects. (2) As noted above,
additional in-phase horizontal currents appear on the top-hat
disc. Getting the maximum of the radiation patterns precisely
along the z-axis with this design requires a similar slight
trade-off with one of the other performance characteristics. The
following parameter studies helped optimize this final design.
IV. DESIGN PARAMETER STUDIES
Designs of the top-hat loaded CSRR-based antenna were con-
sidered in which the height was varied from 115 mm to 1.5
mm. As shown in Figs. 9(a) and 9(b) it was necessary to enlarge
the radius of top-hat disc from 1.0 mm to 84.0 mm to maintain
the impedance match at 300 MHz as was varied. Once
was set, the R_CSRR and R_hat values were varied to obtain
a resonance at 300 MHz. Fig. 9(b) also shows that the 10 dB
impedance bandwidth remained about the same value for most
of the choices of . Furthermore, it shows that all the
values were lower than 20 dB, indicating that one can main-
tain nearly complete impedance matching for different form fac-
tors of this antenna.
Additional performance characteristic variations with
were studied. It was found that once is above 5 mm, it
has little impact on the values of the directivity, gain, and
realized gain in direction and the radiation efficiency. It
does, however, have a significant impact, as shown by Fig. 10,
on the directions of the maximum directivity and its nulls. The
directivity in the direction varies between 3.60 dB and 3.91
dB, the maximum occurring for the smallest values. The
directivity maximum does cross the z-axis, but for a height
that is about 4 times larger and at a cost of some maximum
directivity and bandwidth. When is less than 20 mm, the tilt
of the maximum directivity is within 3 , and the tilt of the two
dips is within 4 . In addition, the half-power beamwidths in
the E- and H-planes are as wide as 87 and 112 , respectively.
On the other hand, the radiation efficiency is nearly constant,
around 98%, for most values. Note that when is further
decreased from 5.0 to 1.5 mm (i.e., to only 0.5 mm air gap
between the top-hat and the grounded discs), the top-hat radius
quickly changes from 68.5 mm to 85.5 mm and, hence, nearly
completely covers the CSRR element. As noted above, the

6. 4424 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013
Fig. 10. Impact of on the tilt of the radiation pattern. (a) Angle of the max-
imum directivity from the direction, and (b) angles of the minimum direc-
tivities near the x-axis.
currents on the bottom of the top-hat effectively cancel those
in the CSRR gap and the radiation efficiency drops sharply to
only 21.7%. In this limit the ESA is a poor radiator, operating
just like a parallel-plate capacitor or high-Q cavity.
It was also found that the impedance matching is not affected
much when the various radii values are changed. On the other
hand, since they impact the perimeter length of the CSRR, the
width and gap sizes of the CSRR were expected to have a much
larger impact. However, as those parameters were varied, it was
found that the radii values had to also be varied to maintain
the resonance frequency with a value below 20 dB.
Impedance matching was relatively straightforward for varia-
tions from 13 to 26 mm in the CSRR width and gap values along
with variations of 110 to 160 mm in the grounded disc radius.
Better than 25 dB values were obtained in all cases.
Finally, the effects the top-hat radius, the CSRR radius, the
grounded disc radius, the CSRR width and the CSRR gap size
were studied. As shown in Fig. 11(a), only small changes in the
top-hat radius, the CSRR radius, and the CSRR width were nec-
essary to tune the system. The points indicate values at which
the 300 MHz resonance frequency was achieved with a
value below 20 dB. Fig. 11(b) shows that both the 10 dB
impedance bandwidth and the radiation efficiency increase
only slightly when the CSRR width increases. Essentially the
same behaviors were obtained when the CSRR gap size was
varied. Similarly, as Fig. 11(c) shows, changing the radius of
the grounded disc requires small changes in the top-hat and
CSRR radii. Fig. 11(d) shows that when the grounded disc size
is increased, both the 10 dB impedance bandwidth and the
radiation efficiency also increase. While the radiation efficiency
variations are small, the bandwidth is increased substantially.
We note that over this disc size variation, the value changes
from 0.66 up to 0.97.
V. ENRICHING THE ULTRA-LOW PROFILE DESIGN
The design of choice given in Section II took into account
the trade-offs identified in these parameter studies. While the
design accomplished the desired goals, further enhancements
were considered. In particular, designs having and
having a , were obtained.
A. Design With
To achieve a smaller value, the resonator had to be re-
designed to be smaller, yet produce the same resonance fre-
quency. The resulting design and its dimensions are shown in
Fig. 11. Impacts of the top-hat radius, the CSRR radius, the grounded disc
radius, the CSRR width and the CSRR gap size on the antenna performance
characteristics. Geometry parameter variations (a), and corresponding changes
in the bandwidth and radiation efficiency at 300 MHz (b), when the CSRR width
varies. Geometry parameter variations (c), and corresponding changes in the
bandwidth and radiation efficiency at 300 MHz (d), when the ground disc radius
varies.
Fig. 12. Configuration of the miniaturized top-hat loaded, CSRR-based an-
tenna. (a) 3D view, (b) top view, and (c) side view. The design parameters in
mm: , , , ,
, , , ,
and .
Fig. 12. Basically it allows for a doubling of the edge length
of the slot. The HFSS and CST simulated results are given
in Figs. 13 and 14. The HFSS (CST) resonance frequency in
Fig. 13(a) is 300.185 MHz (299.1 MHz), where
( 20.28 dB). The 10 dB bandwidth is 0.452 MHz
(0.500 MHz). The radiation efficiency is 81.88% (78.19%) for
this (0.495) design. Substituting perfect electric
conductors for the copper elements, it was found that the 20%
reduction in the radiation efficiency from the de-
sign is due to the copper losses.
The E- and H-plane directivity patterns at the resonance fre-
quency are given in Fig. 13(b). The HFSS (CST) predicted peak
radiation direction is now only away from z-axis

7. TANG AND ZIOLKOWSKI: A STUDY OF LOW-PROFILE, BROADSIDE RADIATION, EFFICIENT, ELECTRICALLY SMALL ANTENNAS 4425
Fig. 13. HFSS and CST simulation results for the miniaturized top-loaded,
CSRR-based antenna. (a) values versus frequency, and (b) E- and H-plane
directivity patterns at the resonance frequency: 300.185 MHz (HFSS) and
299.10 MHz (CST).
Fig. 14. Current distributions on all of the horizontal surfaces of the top-hat
loaded, CSRR-based antenna. On the (a) top and (b) bottom of the top-hat. On
the (c) top and (d) bottom of the CSRR-modified grounded disc.
in the E-plane, and the two E-plane nulls appear at
and 89.4 (89.2 ). The directivity in the direction
is 3.69 dB (3.40 dB). While the radiation efficiency has been
reduced, this design exhibits an excellent broadside radiation
result while maintaining its ultra-low profile .
Fig. 15. Top-hat loaded, CSRR-based antenna augmented with a slot-modified
parasitic copper disc with the end of its slot near the edge of the CSRR element.
The dimensions in mm: , , , ,
, , , , ,
, and . The copper parasitic disc is taken to be 2.0 mm thick.
The HFSS-predicted current distributions on the horizontal
surfaces of this miniaturized CSRR-based antenna are shown in
Fig. 14. In general, these current behaviors are very similar to
those shown in Fig. 8 for the larger design. The most important
difference, of course, is that there are now two large net x-di-
rected current densities, which are in-phase with each other. The
length of the dashed line in Fig. 14(c) is
, which explains this in-phase outcome.
B. Design With Higher Directivity
Even though the top-loaded, CSRR-based antenna exhibits a
relatively high directivity despite its electrically small size, it
generates nearly equal lobes in both the and directions.
This is undesirable for many wireless applications in general.
A CSRR-based design, which has a high broadside directivity
and a large FTBR, while maintaining its electrically small size
and low profile, was desired. We note that a conventional patch
antenna has a reasonable FTBR because its ground plane size is
electrically large. On the other hand, it would be possible to aug-
ment the ESA with a structured ground plane to remove the back
lobe [21]. Unfortunately, this is a complex solution. We decided
to introduce, as was done for another type of metamaterial-in-
spired antenna in [22], a slot-modified, parasitic conducting disc
to achieve the same effects.
Following the design philosophy introduced in [22], a para-
sitic copper disc, centered on—but not connected to—the coax
feedline was introduced below the original antenna. Initial sim-
ulations indicated that some care in the design would have to
be exercised to prevent the currents on the parasitic disc from
shorting-out those on the original antenna. Radiation efficiency
was impacted by the size of the circular hole in the center of the
disc. With it being large, one could maintain a high radiation
efficiency, but then without much impact on the FTBR. Then,
abiding by the original design principles to increase the FTBR
value [22], a pair of meander-line slots were cut into this par-
asitic disc starting from the outside edges, proceeding inwards
towards the inner edge, and centered along the y-axis (i.e., in
the H-plane). To maximize its impact on the current produced
on the top of the parasitic disc, the head of the meander-line
slot was placed approximately halfway between the outer and
inside edges of the parasitic disc, just under the outer edge of
the CSRR element. This configuration is shown in Fig. 15.
To enhance the directivity and FTBR value, the length of
one edge of the slot is set on the order of one-quarter wave-
length (around ). Subsequently, the ground size had to be

8. 4426 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013
increased slightly to accommodate two slots of this size. The
distance between the top face of the parasitic disc and the top
face of the CSRR element is .
The optimized antenna system is presented in Fig. 15. The coax
feedline extends from the base of the CSRR element above
to below the parasitic disc. As for the top-hat loaded, CSRR-
based antenna itself, only two of its dimensions had to be ad-
justed from those given in Fig. 6, i.e., and
, to compensate for the presence of the
parasitic disc in the impedance matching.
The HFSS and CST simulation results for this antenna
are shown in Figs. 16 and 17. The HFSS (CST) results
in Fig. 16(a) demonstrate that the antenna is nearly com-
pletely matched to the source at 299.98 MHz (299.2 MHz),
with ( 23.24 dB) and the 10 dB
impedance bandwidth is 0.675 MHz (0.708 MHz). The pres-
ence of the parasitic disc further decreases an already small
bandwidth. On the other hand, the E- and H-plane directivity
patterns given at the resonance frequencies in Fig. 19(b) show
that the directivity in the direction is 7.25 dB (7.47 dB) and
the (27.33 dB). The radiation efficiency
is 92.70% (89.05%) for this design. The E-plane
directivity pattern at the resonance frequency in Fig. 16(b)
is asymmetric because of the asymmetric CSRR structure
itself. The current distribution on the top of the parasitic disc
is shown in Fig. 17. It exhibits the same behaviors as those
observed in the original study [22]. In particular, the large
in-phase x-directed currents along the inside edges of the slots
are responsible for cancelling the back lobe and reinforcing the
front lobe. The net result of these x-directed edge currents and
the image current of the CSRR structure is that the parasitic
disc acts like a dipole in-phase with the CSRR structure, i.e.,
the overall system is an effective two element end-fire array
with closely spaced elements which produces the observed
high directivity, as asserted in [23].
At the HFSS-predicted resonance frequency, one has
, which is not electrically small because of the requisite
radius of the parasitic disc. Nonetheless, in comparison with
a standard half-wavelength patch antenna whose ground plane
size is twice its patch size, as reported in [14], the patch an-
tenna’s directivity is slightly larger, 7.34 dB, but its FTBR, 17
dB, is lower. We also note that, while effective, this method
increased the overall diameter of the entire system to ,
which is near to the size of the radiating element of the patch an-
tenna. Thus a reduction in size of this configuration was sought.
We found that by reversing the direction of the slot and its
origin, the size of the parasitic disc could be reduced. This con-
figuration is shown in Fig. 18. In particular, it radius was de-
creased 34%. The R_hat and R_CSRR values again had to be
tuned to 56.0 mm and 83.65 mm, respectively, to achieve a
nearly complete impedance match at 300 MHz.
The HFSS and CST simulation results for this antenna
are shown in Figs. 19 and 20. The HFSS (CST) results
in Fig. 19(a) demonstrate that the antenna is nearly com-
pletely matched to the source at 300.02 MHz (300.6 MHz),
with ( 21.22 dB) and the 10 dB
impedance bandwidth is 1.07 MHz (1.01 MHz). The E- and
H-plane directivity patterns given at the resonance frequencies
Fig. 16. HFSS and CST simulation results of the top-hat loaded, CSRR-based
antenna shown in Fig. 15. (a) values as a function of the frequency, and
(b) E- and H-plane directivity patterns at the resonance frequency: 299.98 MHz
(HFSS) and at 299.2 MHz (CST).
Fig. 17. HFSS-predicted current distribution on the upper surface of the slot-
modified parasitic copper disc of the antenna shown in Fig. 15 at the resonance
frequency, 300.2 MHz.
in Fig. 19(b) show that the directivity in the direction is
7.11 dB (7.37 dB) and the (9.84 dB). The
radiation efficiency is 94.68% (91.04%) for this
design.
Fig. 20 shows the HFSS-predicted current distribution on the
top surface of the parasitic disc. Strong in-phase x-directed cur-
rents again are concentrated on the heads of the meander-line

9. TANG AND ZIOLKOWSKI: A STUDY OF LOW-PROFILE, BROADSIDE RADIATION, EFFICIENT, ELECTRICALLY SMALL ANTENNAS 4427
Fig. 18. Top-hat loaded, CSRR-based antenna augmented with a slot-modified
parasitic copper disc with the end of its slot near the edge of the parasitic disc.
The dimensions are in mm: , , , ,
, , , , ,
, and .
Fig. 19. HFSS and CST simulation results of the top-hat loaded, CSRR-based
antenna shown in Fig. 18. (a) values as a function of the frequency, and
(b) E- and H-plane directivity patterns at the resonance frequency: 300.02 MHz
(HFSS) and at 300.6 MHz (CST).
slots. Here, they are located on the outside edges of the par-
asitic disc. The total length of one edge of the slot, as mea-
sured along the black dashed line on the right side of center-cut
disc, is , slightly larger than one-quarter
wavelength. Because the slots are now connected to the interior
cut-out disc, their lengths could be made this requisite distance
without substantially increasing the size of the parasitic disc. It
is noted that the current induced near the x-axis is very weak on
the surface of the parasitic disc. This area is labeled by the red
Fig. 20. HFSS-predicted current distribution on the upper surface of the slot-
modified parasitic copper disc of the antenna shown in Fig. 18 at the resonance
frequency, 300.02 MHz.
Fig. 21. Impact of the design parameter H_disc on the FTBR values of the
antenna given in Fig. 18.
dotted line; it has little to do with the radiation processes. This
was confirmed by simulating the configuration in which two 5.0
mm wide slots, centered along the vertical dotted red line, were
cut. Consequently, the large currents near the ends of the mean-
derline slots, which occur near the y-axis edges of the parasitic
disc, are responsible for increasing the FTBR.
Thus, this electrically small design with the reversed slot in
the parasitic disc increased the bandwidth, decreased the asym-
metry in the pattern, increased the radiation efficiency, but de-
creased the FTBR value, all while remaining low profile and
maintaining about the same maximum directivity. It was found
that the FTBR ratio could be increased at the cost of the low
profile characteristic. By increasing H_disc, the height of the
original antenna above the parasitic disc, the FTBR near 300
MHz was found to increase, as illustrated in Fig. 21. This curve
was obtained by optimizing the total length of meander-line slot
to operate with its maximum FTBR value near 300 MHz. Note
that when H_disc increases to 50 mm , the FTBR value
is improved to 10.7 dB; when it increases to 115 mm ,
the FTBR value reaches 20.5 dB. Changing this height param-
eter changes the magnitudes and phases of the currents at the
ends of the slots, which leads to a more effective cancellation of
the back lobe and, hence, to the improved FTBR values.
VI. MEASUREMENTS
An experiment was carried out to validate the basic design
principles in support of the reported simulation results. We se-

10. 4428 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013
Fig. 22. Configuration of the coaxially-fed, top-hat loaded, CSRR-based an-
tenna with a sleeve balun introduced to mitigate any cable currents. (a) 3D
view, (b) top view, (c) side view, and (d) the fabricated antenna. The design
parameters in mm: , , ,
, , , ,
, , , , , ,
, , and .
lected the ESA design illustrated in Fig. 6 as the proof-of-con-
cept example; it represents the basic antenna concepts in our
studies. However, because the lowest possible frequency for an
antenna under test (AUT) in the UESTC anechoic chamber is
800 MHz, the basic design was rescaled for operation near 1.0
GHz.
The antenna design and dimensions are shown in Fig. 22.
After the first test, which was based on the original design, it was
determined that the top loading required mechanical stability
to maintain its height above the CSRR element. Three posts,
which were each combinations of Teflon-based screws, blind-
nuts and pads, were installed symmetrically around the -axis
and, hence, the coax-feed, as well as with respect to the tab in the
CSRR slot. According to our simulation studies, these posts had
little effect on the antenna performance because, as positioned,
they had essentially no impact on the current pathways on the
central portion of the CSRR element.
Because of the presence of the slot, the fields radiated by the
AUT couple significantly to the coax feedline and, hence, to
the vector network analyzer (VNA) cable. Consequently, the
cable effects in the initial measurements were very large and
had a negative impact on them. In order to reduce these leakage
currents on the cable, a sleeve balun was designed and placed
directly beneath the AUT to act as a choke and to isolate the
AUT and the VNA cable. The parameters of the sleeve balun,
which are also given in Fig. 22, were generated from known
designs and HFSS simulations [24]–[26].
Fig. 23. Comparison of the electric field distributions in the z0x-cut plane sim-
ulated for different phase angles at the resonance frequencies of the CSRR-based
antennas that are fed with a 100 mm long coaxial cable. (a) AUT without the
sleeve balun, ; and (b) AUT with sleeve balun,
.
Fig. 24. Input impedance reflection coefficient for various configura-
tions of the CSRR-based antenna illustrated in Fig. 22.
The indispensability of the sleeve balun to the success of the
experiments is emphasized in Figs. 23 and 24. It is readily ob-
served from the simulation results in Fig. 23 that the presence
of a properly designed sleeve balun prevents the presence of
the leakage currents on the outer wall/shield of the feedline and
cable. It is also apparent in comparing Figs. 23(a) and 23(b),
as expected [27], that these stray currents have a significant
negative impact on the impedance matching and the near- and
far-field radiation pattern characteristics of the AUT.
As demonstrated in Fig. 24, when a coax cable having a small
length, 5.0 mm, is employed in the design of the fabricated pro-
totype CSRR-based antenna, which is similar to the idealized
designs given in Figs. 2, 6, 15, and 18, it can be well matched
to the 50 source at 1.002 GHz with
and a 10 dB impedance bandwidth equal to 9.25 MHz. Also,
its radiation pattern is similar to the corresponding one given
in Fig. 7(b). However, when the length of the coaxial cable is
increased, it is found that the resonance frequency shifts to a
higher value, 1.115 GHz; the impedance match becomes poor,
; and the radiation pattern is deteriorated.
As shown in Fig. 23(a), these deteriorated performance charac-
teristics are a result of the leakage current on the feedline.

11. TANG AND ZIOLKOWSKI: A STUDY OF LOW-PROFILE, BROADSIDE RADIATION, EFFICIENT, ELECTRICALLY SMALL ANTENNAS 4429
Fig. 25. Simulated and measured radiation patterns of the ESA with balun; (a)
E-plane (z0x-plane), (b) H-plane (z0y-plane).
With the presence of the sleeve balun, which still perturbs the
fields and currents of the AUT, the resonance frequency shift is
much smaller and the impedance match remains quite good. Ac-
cording to our HFSS simulation results, also shown in Fig. 24,
an acceptable matching level is obtained at
with and a 10 dB impedance band-
width equal to 6.15 MHz when the sleeve balun is optimized.
The corresponding measured results, also shown in Fig. 24,
are in very reasonable agreement, i.e., with
and a 10 dB impedance bandwidth
equal to 5.25 MHz. The difference between the simulated and
measured resonance frequency is thus only 0.98%. Given this
outcome, the 3.2% difference between the base design without
the balun and the measured results with the balun, which it-
self is quite reasonable, could be alleviated, if desired, with a
straightforward retuning of the design parameters. The reason
for the reduction of the impedance bandwidth from the smaller
coax length case is that the sleeve balun is also a narrow-band
resonator [24], [25]. In addition, the simulation studies also re-
vealed that the input impedance and the radiation patterns of
the AUT with the sleeve balun remained basically unchanged
regardless of the length of the feed line.
Fig. 25 gives the simulated and measured E- and H-plane di-
rectivity patterns. The measured values show very good agree-
ment with the simulated results. They are both similar to those
given in Fig. 7(b) for the idealized case. We also note that the
cross-polarizations (especially in the H-plane) are a little higher
than those associated with the idealized design. One finds that
the peak gain around the -axis is 3.48 dB, which is a little
lower than the idealized case value, 3.82 dB. The measured
value was 2.76 dB. Notice that in Fig. 25 the measured peak
gain value in the E-plane is lower and tilted away from the
z-axis around 6.5 in comparison with the simulated results.
Moreover, the cross-polarization levels in both planes become
slightly higher. The main reasons for the differences in the sim-
ulated and measured behaviors are dimension errors associated
with the fabrication tolerances, particularly since both the AUT
and the balun are narrow-bandwidth resonators.
VII. CONCLUSION
Several CSRR-based antennas were introduced and their
performance characteristics were investigated numerically.
Their properties: electrically small, nearly complete impedance
match, nearly broadside radiation, and high radiation efficiency,
TABLE I
VALUES FOR THE REPORTED ANTENNA DESIGNS
were confirmed with two computational electromagnetics
solvers, one in the frequency domain (HFSS) and one in the
time domain (CST). The inclusion of the CSRRs in the designs
led to several interesting properties, particularly their directivity
patterns with their maxima nearly orthogonal to the CSRR ele-
ment, rather than orthogonal to their driven monopole element.
The addition of a copper hat to the monopole facilitated an
ultra-low profile design. This design was modified to enhance
its properties further for potential applications. One was an
antenna that was miniaturized further; the other was an antenna
with higher directivity and a high FTBR value.
We note that the bandwidths of all of the presented CSRR-
based ESAs are quite narrow. Their corresponding values
are compared with the electric-based lower bound [28]–[30]:
, where , and
, in Table I. Nonetheless, given their low-pro-
file, electrically small and high radiation efficiency properties,
these types of ESAs are simple and may be attractive for a
variety of narrow bandwidth wireless applications, including
precise transponders (e.g., friend-foe identification); sensor de-
vices; and GPS, RFID and wireless power transfer (WPT) sys-
tems. Moreover, given the NFRP nature of the CSRR antenna
design, we anticipate that we will be able to extend it with both
frequency agile and non-Foster augmentations to enhance its
operational frequency range and instantaneous bandwidth, re-
spectively, for other applications. These possibilities are cur-
rently under investigation.
A prototype system based on the basic design was fabricated
and tested near 1.0 GHz to coincide with the measurement setup
capabilities. Reasonable agreement between the simulated and
measured performance characteristics was obtained, confirming
the design principles reported.
REFERENCES
[1] J. B. Pendry, A. J. Holden, D. J. Robbins, and W. J. Stewart, “Mag-
netism from conductors and enhanced nonlinear phenomena,” IEEE
Trans. Microw. Theory Tech., vol. 47, no. 11, pp. 2075–2084, Nov.
1999.
[2] F. Falcone, T. Lopetegi, J. D. Baena, R. Marques, F. Martin, and M.
Sorolla, “Effective negative-epsilon stopband microstrip lines based
on complementary split ring resonators,” IEEE Microw. Wireless
Compon. Lett., vol. 14, no. 14, pp. 280–282, Jun. 2004.
[3] J. D. Baena, J. Bonache, F. Martin, R. M. Sillero, F. Falcone, T.
Lopetegi, M. A. G. Laso, J. Garcia-Garcia, I. Gil, M. F. Portillo, and
M. Sorolla, “Equivalent-circuit models for split-ring resonators and
complementary split-ring resonators coupled to planar transmission
lines,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 4, pt. 2, pp.
1451–1461, Apr. 2005.
[4] F. Martin and R. Marqués, “Application of split ring resonators to mi-
crowave circuit design,” in Applications of Metamaterials, F. Capolino,
Ed. Boca Raton, FL, USA: CRC Press, 2009, ch. 9.

12. 4430 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013
[5] A. Erentok, P. Luljak, and R. W. Ziolkowski, “Antenna performance
near a volumetric metamaterial realization of an artificial magnetic con-
ductor,” IEEE Trans. Antennas and Propagat., vol. 53, pp. 160–172,
Jan. 2005.
[6] P. Ikonen, S. Maslovski, and S. Tretyakov, “PIFA loaded with arti-
ficial magnetic material: Practical example for two utilization strate-
gies,” Microw. Opt. Tech. Lett., vol. 46, no. 3, pp. 205–210, Aug. 2005.
[7] K. B. Alici and E. Ozbay, “Electrically small split ring resonator an-
tennas,” J. Appl. Phys., vol. 101, p. 083104, Apr. 2007.
[8] F. Bilotti, A. Alú, and L. Vegni, “Design of miniaturized metamate-
rial patch antennas with -negative loading,” IEEE Trans. Antennas
Propag., vol. 56, no. 6, pp. 1640–1647, Jun. 2008.
[9] M. Palandoken, A. Grede, and H. Henke, “Broadband microstrip an-
tenna with left-handed metamaterials,” IEEE Trans. Antennas Propag.,
vol. 57, no. 2, pp. 331–338, Feb. 2009.
[10] H. Zhang, Y. Q. Li, X. Chen, Y. Q. Fu, and N. C. Yuan, “Design
of circular/dual-frequency linear polarization antennas based on the
anisotropic complementary split ring resonator,” IEEE Trans. Antennas
Propag., vol. 57, no. 10, pp. 3352–3355, Oct. 2009.
[11] M. Selvanayagam and G. V. Eleftheriades, “A compact printed
antenna with an embedded double-tuned metamaterial matching net-
work,” IEEE Trans. Antennas Propag., vol. 58, no. 7, pp. 2354–2361,
Jul. 2010.
[12] M.-C. Tang, S. Xiao, T. Deng, D. Wang, J. Guan, B. Wang, and G.
Ge, “Compact UWB antenna with multiple band-notches for WiMAX
and WLAN,” IEEE Trans. Antennas Propag., vol. 59, no. 4, pp.
1372–1376, Apr. 2011.
[13] Y. Dong, H. Toyao, and T. Itoh, “Design and characterization of
miniaturized patch antennas loaded with complementary split-ring
resonators,” IEEE Trans. Antennas Propag., vol. 60, no. 2, pp.
772–785, Feb. 2012.
[14] R. O. Ouedraogo, E. J. Rothwell, A. R. Diaz, K. Fuchi, and A. Temme,
“Miniaturization of patch antennas using a metamaterial-inspired tech-
nique,” IEEE Trans. Antennas Propag., vol. 60, no. 5, pp. 2175–2181,
May 2012.
[15] C.-C. Lin, P. Jin, and R. W. Ziolkowski, “Single, dual and
tri-band-notched ultra-wideband (UWB) antennas using capaci-
tively loaded loop (CLL) resonators,” IEEE Trans. Antennas Propag.,
vol. 60, pp. 102–109, Jan. 2012.
[16] ANSYS/ANSOFT High Frequency Structure Simulation (HFSS) ver.
13.0, ANSYS Corp. [Online]. Available: www.ansoft.com
[17] CST Microwave Studio CST of America, Inc. [Online]. Available:
http://www.cst.com
[18] C. A. Balanis, Antenna Theory: Analysis and Design, 3rd ed. New
York, NY, USA: Wiley Interscience, 2005.
[19] C. A. Balanis and D. Decarlo, “Monopole antenna patterns on finite size
composite ground plane,” IEEE Trans. Antennas Propag., vol. AP-30,
no. 4, Jul. 1982.
[20] R. W. Ziolkowski, P. Jin, and C.-C. Lin, “Metamaterial-inspired engi-
neering of antennas,” Proc. IEEE, vol. 99, pp. 1720–1731, Oct. 2011.
[21] P. Jin and R. W. Ziolkowski, “High directivity, electrically small, low-
profile, near-field resonant parasitic antennas,” IEEE Antennas Wire-
less Propag. Lett., vol. 11, pp. 305–309, 2012.
[22] M.-C. Tang and R. W. Ziolkowski, “Efficient, high directivity, large
front-to-back-ratio, electrically small, near-field-resonant-parasitic an-
tenna,” IEEE Access, vol. 1, pp. 16–28, May 2013.
[23] A. D. Yaghjian, T. H. O’Donnell, E. E. Altshuler, and S. R. Best,
“Electrically small supergain end-fire arrays,” Radio Sci., vol. 43, no.
RS3002, pp. 1–13, 2008.
[24] C. Icheln, M. Popov, P. Vainikainen, and S. He, “Optimal reduction
of the influence of RF feed cables in small antenna measurements,”
Microw. Opt. Tech. Lett., vol. 25, no. 3, pp. 194–196, May 2000.
[25] S. A. Saario, J. W. Lu, and D. V. Thiel, “Full-wave analysis of choking
characteristics of sleeve balun on coaxial cables,” Electron. Lett., vol.
38, no. 7, pp. 304–305, Mar. 2002.
[26] C. Icheln, J. Krogerus, and P. Vainikainen, “Use of balun chokes
in small-antenna radiation measurements,” IEEE Trans. Antennas
Propag., vol. 53, no. 2, pp. 498–506, Apr. 2004.
[27] J. Oh and K. Sarabandi, “Low profile, miniaturized, inductively cou-
pled capacitively loaded monopole antenna,” IEEE Trans. Antennas
Propag., vol. 60, no. 3, pp. 1206–1213, Mar. 2012.
[28] L. J. Chu, “Physical limitations of omni-directional antennas,” J. Appl.
Phys., vol. 19, pp. 1163–1175, Dec. 1948.
[29] H. L. Thal, “New radiation limits for spherical wire antennas,” IEEE
Trans. Antennas Propag., vol. 54, no. 10, pp. 2757–2763, Oct. 2006.
[30] S. R. Best, “Low Q electrically small linear and elliptical polarized
spherical dipole antennas,” IEEE Trans. Antennas Propag., vol. 53, pp.
1047–1053, Mar. 2005.
Ming-Chun Tang (S’12–M’13) received the B.S.
degree in physics from the Neijiang Normal Univer-
sity, Neijiang, China, in 2005 and the Ph.D. degree
in radio physics from the University of Electronic
Science and Technology of China (UESTC), in 2013.
From August 2011 to August 2012, he was also
with the Department of Electrical and Computer En-
gineering, The University of Arizona, Tucson, AZ,
USA, as a Visiting Scholar. He is currently an As-
sistant Professor in the College of Communication
Engineering, Chongqing University, China. His re-
search interests include electrically small antennas, RF circuits, metamaterial
designs and their applications.
Prof. Tang was a recipient of the Best Student Paper Award in the 2010 In-
ternational Symposium on Signals, Systems and Electronics (ISSSE2010) held
in Nanjing, China. He is serving as a reviewer for IEEE journals including the
IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS and IEEE ANTENNAS
AND PROPAGATION MAGAZINE.
Richard W. Ziolkowski (M’97–SM’91–F’94)
received the Sc.B. degree in physics (magna cum
laude with honors) from Brown University, Prov-
idence, RI, USA, in 1974 and the M.S. and Ph.D.
degrees in physics from the University of Illinois at
Urbana-Champaign, IL, USA, in 1975 and 1980, re-
spectively. He was awarded an Honorary Doctorate,
Doctor Technish Honoris Causa, from the Technical
University of Denmark (DTU) in 2012.
He was a member of the Engineering Research
Division, Lawrence Livermore National Laboratory,
from 1981 to 1990 and served as the Leader of the Computational Electronics
and Electromagnetics Thrust Area for the Engineering Directorate. He currently
is serving as the Litton Industries John M. Leonis Distinguished Professor
in the Department of Electrical and Computer Engineering, University of
Arizona, Tuscon, AZ, USA. He holds a joint appointment with the College
of Optical Sciences. His research interests include the application of new
physics and engineering ideas to linear and nonlinear problems dealing with
the interaction of electromagnetic waves with complex media, metamaterials,
and realistic structures.
Prof. Ziolkowski is an IEEE Fellow and an OSA Fellow. He was the President
of the IEEE Antennas and Propagation Society in 2005. He continues to be very
active in the IEEE, OSA, and APS professional societies.