A Second-Order Low-Power ΔΣ Modulator For Pressure Sensor Applications Tero Nieminen and Kari Halonen Aalto University School of Science and Technology Department of Micro- and Nanosciences P.0 Box 11000, FIN-00076 Aalto, Finland Email: firstname.lastname@example.org.ﬁ Telephone: +358 9 470 22275 Abstract—In this paper, an 1-bit second-order low-power ΔΣ clkinmodulator for pressure sensor applications is presented. Themodulator utilizes correlated double-sampling (CDS) in order toreduce the ﬂicker (1/f) noise. Due to the 1-bit output, the feedbackDAC is inherently linear. The modulator is designed with 0.35- Clockµm CMOS process. Measured signal-to-noise and distortion ratio divider(SNDR) is 86dB (14bits), while the current consumption is 14µA. I. I NTRODUCTION Clock Clock In modern sensor applications, accurate analog-to-digital generator generatorconverters (ADCs) with a low sampling frequency are needed.Thus, a ΔΣ technique is a good alternative to be used.Concerning stability, robustness and power consumption ofthe modulator, a second-order implementation is a tempting In+ outchoice compared to higher-order ones, despite of less ag- S/H ΔΣgressive noise-shaping. Utilizing one-bit output, achievable In-signal-to-noise ratio (SNR) is decreased compared to multi-bit implementations, but on the other hand, feedback DAC Fig. 1. The system block diagramnonlinearity is not an issue. In this work, a second-order one-bit ΔΣ modulator isimplemented in order to meet the speciﬁcations of the data DACconversion needed in the pressure sensor application. The -1 -1paper is organized as follows. In Section II the modulator 1 1/7 1 7/2 1/6 1architecture is presented. Section III presents the circuit de- in z-1 z-1 outscription. Measured performance is shown in Section IV andconclusions in V. Fig. 2. Block diagram of the modulator II. S YSTEM DESCRIPTION clock, and clock generators create non-overlapping clock sig- The top level block diagram of the system is shown in Fig. 1. nals for the SH and integrators. The clock generation circuitryThe system consists of a sample/hold (SH) front-end stage, the is more detailed discussed in section III.modulator, the clock divider and the clock generators. The SHstage is utilized in order to sample the resistive sensor element III. C IRCUIT D ESCRIPTION(Wheatstone bridge) to large on-chip capacitors to reducethe system current consumption. The modulator architecture A. The SH stageis selected to be cascade-of-integrators-feedback (CIFB)  In this work, low power consumption is one of the keyas depicted in Fig 2. The ﬁrst integrator utilizes correlated speciﬁcations of the system. However, resistance of the sensordouble-sampling to reduce the ﬂicker noise. elements is in order of some kΩs, which would lead to system All clock signals are generated internally from a single current consumption of ≈ 1mA. This is clearly too high, andsquare 32.768kHz clock input. SH circuit needs separate thus the design utilizes a SH circuit that samples the sensorclock signals with different duty cycle than 50/50. Hence, to large on-chip capacitors. To reduce the average powera frequency-by-two clock divider is employed for the input consumption, the sensor element current from positive to978-1-4244-8971-8/10$26.00 c 2010 IEEE
1.4pF 0.3pF Vref 2 Vref 2 1 1 1 fb+ fb- 1 1 fb+ fb- Vcm Vcm x1 9.8pF 1.8pF 2 1 1 2 1 2 1 Vin+ − − + out+ 2 + + + 1 1.4pF 2 1.4pF - 1 1.05pF 2 - - out- Vin- + + - 1 2 1 1 x2 2 1 1 2 1 9.8pF 1.8pF Vcm Vcm fb- fb- fb+ 2 2 1 fb- fb+ 2 1 A fb+ Vref 1 Vref 1 1.4pF 0.3pF xA Fig. 4. Implemented 2nd order ΔΣ modulator B xB Vdd VBIASP Fig. 3. The clock timing M9 M10 VCASP M7 M8negative supply is cut (phase B, see Fig 3), and the current is VCASN Vo- Vo+ﬂowing only a short period when the sensor element (resistive Vin+ Vin-Wheatstone bridge) voltage is sampled to the capacitors (phase M5 M1 M2 M6A). The SH clock operates either at 1/4 or 1/8 of the modulatorfrequency. Fig 3 presents the clock timing (in the ﬁgure, VBIASNSH sampling frequency is 1/4). Phases 1 and 2 are the ﬁrst M3 M0 M4 VCMFBintegrator sampling and integrating phases, respectively. A isthe input sampling phase of the SH. The sampling capacitorscan be selected to be 25pF or 50pF (control bit sel_cap). Fig. 5. Folded-Cascode OTAThe capacitors have to be much larger than the ﬁrst integratorsampling capacitor to minimize the voltage drop at everyintegrator sampling phase. The differential input is sampled(phase A), doubled and fed into the ﬁrst integrator. clock period), or alternatively impractically large SH sampling capacitors should be used.B. The modulator All switches are implemented as transmission gates, and The modulator is implemented as presented in Fig 4. PMOS transistors are dimensioned three times as large asThe ﬁrst integrator is implemented as CDS  (correlated NMOS devices (ratio of the transconductance parameter) todouble sampling) integrator to reduce 1/f noise. This results have symmetrical on-resistance with respect to the signalin one additional capacitor and three additional switches for common-mode level. It was simulated that the linearity ofboth branches (+ and -) in the ﬁrst integrator. Otherwise, 16-bits can be achieved with the switches utilized. Thus, itboth integrators are implemented as conventional, noniverting is not necessary to use any bootstrapped switches (almost(with delays) SC integrators. Feedbacks are implemented with constant on-resistance of the switch, which reduces the switchone voltage reference (Vref in the ﬁgure) by inverting the nonlinearity and harmonic distortion).other branch (in both integrators), as presented in Figure 4. In the modulator, folded cascode operational transconduc-With two references, the input sampling and the feedback tance ampliﬁer (OTA, Figure 5) is selected, due to its largecapacitors in the ﬁrst integrator could be combined as one gain-bandwidth (GBW) and open-loop DC-gain. Simulatedcapacitor, because the input signal and the feedback constants open-loop DC-gain and GBW of the OTA are 88dB andare equal. Also thermal noise would be minimized with a 3.0MHz (with 1pF load), respectively. Common-mode feed-common capacitor. The need for only one reference is chosen back (CMFB) is realized by switched capacitors (SC) .rather than savings of capacitors (and few switches). Anotheradvantage of using separate capacitors is that the signal- The quantizer is implemented as a regenerative comparatordependent loading for the voltage reference is avoided. Size (Fig 6). During the reset phase (phase 2), the outputs areof the sampling capacitors are determined by speciﬁed input shorted by a CMOS switch. At the end of the phase 2, voltagesignal amplitude range, oversampling ratio (OSR) and targeted is regenerated across the CMOS reset switch. At the beginningsignal-to-noise ratio (SNR). However, the input capacitors of the latch phase (phase 1), NMOS latch is turned on and(1.4pF in this case) cannot be very large, since the modulator is logic levels on the outputs are generated. The result is storeddriven by a sample/hold (SH) stage. Too large modulator input to the D ﬂipﬂop at the end of the latch phase. Flowing ofcapacitors lead to increased voltage drop on the SH output static current is prevented by the switch transistors Ms1 −Ms2 .(the sensor is sampled either every fourth or eighth modulator Connected between the input transistor drains and the outputs,
Vdd Vdd M5 M6 Vout- Ms2 latch Ms1 Vout+ reset Vin+ Vin- M1 M2 reset Vbias M3 M4 MB2 latch Mlatch Fig. 8. The modulator top-level layout Fig. 6. Regenerative comparator TABLE I T HE MODULATOR SUMMARY clkin Process 0.35µm CMOS Supply voltage 2.0-3.3V clkin clkin Input range 0.7Vppdiﬀ clkin clk= Bandwidth 30Hz clkin/2 OSR 256 clk/4 or clk/8 Area (core+IOs) 2.29 mm2 fsel Current 14µA clkin Mode 5014 5018 2514 2518 Fig. 7. The clock divider Peak SNR[dBc] 86.5 85.5 85 83these switches also reduce the kickback noise. core occupies 0.3mm2 , half of which is occupied by the SHC. Clock generation capacitors (100pF totally) and the other half by the modulator. Separate clock signals are needed for the SH and the Measured output spectrum with 50pF sampling capacitorsmodulator. To obtain the timing as presented in Fig 3, a clock and 1/4 frequency (smallest voltage drops on the SH output) isdivider (Fig 7) is employed. The input clock frequency is presented in Fig 9. The spectrum is calculated from 524288-divided by 2 (for the modulator) and 8 or 16 (for the SH), point Fast-Fourier Transform (FFT) with Kaiser windowingdepending on the control bit ’fsel’. With possibility to reduce (β=13). From the ﬁgure it can be seen that the measuredthe SH frequency, sensor element current consumption can be HD3 is -97.5dBc. SNR can be calculated to be 86dB, whichreduced. Nonoverlapping clocks for the SH and the modulator corresponds to 14-bit accuracy. Current consumption of theare created by a clock generator constructed of inverters and modulator is 14μA. The output spectrum is measured for allNOR gates. Since the clock frequency is low (16.384kHz operating modes (four different combinations of the samplingfor the modulator), relatively long nonoverlap time (10ns) capacitors/SH sampling frequencies). To demonstate the effectfor the clocks are used. Instead of long inverter chain, 0.5pF of the sampling capacitor size and the SH sampling frequency,capacitors are placed to internal nodes of the clock generator. an output spectrum with 25pF capacitors and 1/8 frequency (largest voltage drops on the SH output) is presented in Fig 10. With this cambination, SNR is decreased about 3dB (0.5bits). IV. M EASUREMENTS The largest interference component in the spectra is at 50Hz, The chip was fabricated in double-poly 4-metal 0.35μm originating from the network disturbances. The modulator isCMOS process, and packaged to 20-CDIP ceramic package. summarized in Table I. The current consumption includesMeasurements were done from a 4-layer printed circuit board only the modulator (to which the SH sampling frequency does(PCB). The modulator top-level-layout is shown in Fig 8. not affect). In the table, mode ’5014’ means 50pF input SHTotal area including the core and the IO-pads is 2.29 mm2 . The sampling capacitors and 1/4 SH frequency, and so on.
0 R EFERENCES  R. Schreier and G. C. Temes, Understanding Delta-Sigma Data Convert- −20 ers. USA: John Wiley & Sons, Inc., 2005.  R. Naiknaware and T. Fiez, “142 dB Delta Sigma ADC with a 100nV −40 LSB in a 3V CMOS Process,” in Proc. Custom Integrated Circuits Conf., 21-24 May 2000, pp. 5–8.  M. Dessouky and A. Kaiser, “Input switch conﬁguration suitable for rail-Relative Power [dBc] −60 to-rail operation of switched opamp circuits,” Electronics Letters, vol. 35, Jan. 1999.  S. R. Norsworthy, R. Schreier, and G. C. Temes, Delta-Sigma Data −80 Converters:Theory, Design, and Simulation. USA: IEEE Press, 1997. −100 −120 −140 −1 0 1 2 3 10 10 10 10 10 Frequency [Hz] Fig. 9. Measured output spectrum, mode ’5014’ 0 −20 −40Relative Power [dBc] −60 −80 −100 −120 −140 −1 0 1 2 3 10 10 10 10 10 Frequency [Hz] Fig. 10. Measured output spectrum, mode ’2518’ V. C ONCLUSION A second-order ΔΣ modulator for pressure sensor applica-tions (with selection modes of the SH sampling capacitorsand sampling frequency) in 0.35μm CMOS process wasintroduced. Architecture, implementation and most essentialdesign issues were discussed. Measurements show that withfull-scale input, SNR of 86.5dB can be achieved with 50pFsampling capacitors and 1/4 sampling frequency. ACKNOWLEDGMENT The authors would like to thank Mikko Snygg from Mi-croAnalogSystems. Mr. Vesa Turunen is acknowledged forhis help in clock generation design and top-level layout. Mr.Matthew Turnquist gave valuable help in sub-block layoutdrawing.