82                                  ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)UDC 621.373.5:621.382.3(...
ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)                                         83from the VCO tank...
84                               ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)     Fig. 4. An ac equivale...
ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)                                                           8...
86                                   ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)        Substituting Ct...
ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)                                            87       Substit...
88                              ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)      For both Clapp and Col...
ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)                                          89                ...
90                          ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)                                ...
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Colpitts startup

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Colpitts startup

  1. 1. 82 ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19)UDC 621.373.5:621.382.3(045) 1 V. V. Ulansky, D.Sc. 2 M.S. Fituri, Ph.D. 3 I. A. Machalin, Ph.D.MATHEMATICAL MODELING OF VOLTAGE-CONTROLLED OSCILLATORS WITH THE COLPITTS AND CLAPP TOPOLOGY 1,2 Al Fateh University, Tripoli, Libya, e-mail: 1 vulanskyi@yahoo.com, 2mfituri@hotmail.com, 3 National Aviation University, e-mail: migor@svitonline.com An analytical modeling is presented for the Colpitts and Clapp voltage-controlled oscillators, from which more precise equations are derived for calculation of oscillation frequency and small-signal negative resistance. Modeling is performed with taking into account a field-effect transistor parasitic capacitances. Finally, a comparison of start-up conditions is made for the Colpitts and Clapp oscillators verified by SPICE simulations. The derived equations assure a more accurate modeling, designing and implementing the Colpitts and Clapp voltage-controlled oscillators. Introduction. LC oscillators are important building blocks in modern communicationsystems. They are widely used as voltage controlled oscillators (VCOs) in phase locked loops to up-and down-convert signals. There are several VCO topologies possible, but the most popular are theColpitts and Clapp topologies [1 – 8]. The Colpitts and Clapp VCO design requires precise circuitmodeling due to an accurate prediction of the oscillating frequency and quantitative estimation ofthe start-up conditions. For these reasons, the modeling of the Colpitts and Clapp VCOs is paidgreat attention [1 – 3], [8; 9]. At present, the small-signal models of the mentioned VCO topologiesdo not include transistor parasitic capacitances. These parasitic elements change the frequencyresponse of the oscillator circuits. Consequently, the oscillation frequency, start-up condition, andother performance characteristics are affected and the real circuit operation differs from thepredicted operation with ideal components. The goal of this article is to present a more accuratemodel for the Colpitts and Clapp VCOs. VCO topologies. The VCOs having the Colpitts and Clapp topologies are shown in fig. 1. a b Fig. 1. VCO with the Colpitts (a) and Clapp (b) topology The VCO`s circuit elements are here represented by the varactor diodes VD1 and VD2 whosecapacitances are controlled by the dc control voltage Vctrl , inductor L, coupling capacitor CC ,source impedance Z S , tank capacitors C1 and C 2 , resistor RC isolating the dc control voltage line
  2. 2. ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) 83from the VCO tank, and resistor RG providing zero bias voltage for the gate of transistor Q in Fig.1(a). For a single oscillating frequency the varactor diodes can be replaced by the capacitors withcapacitances CV 1 , CV 2 and C0  CV 1CV 2 CV 1  CV 2  as shown in fig. 2 (a) and (b). Since thevaractors are usually matched, then CV 1  CV 2  CV and C0  0,5CV . a b Fig. 2. Single-frequency Colpitts (a) and Clapp (b) oscillators The junction field-effect transistor (JFET) Q in fig. 1 and fig. 2 can be replaced with anyactive device, such as a bipolar junction transistor (BJT), a depletion metal-oxide-semiconductorFET (MOSFET) or a metal-semiconductor FET (MESFET). As seen from fig. 2(b), the Colpittsoscillator is a special case of the Clapp oscillator in which an additional capacitor C 0 is used forcontrolling the frequency of oscillation. Indeed, if C0   the circuit of fig. 2(b) becomesequivalent to the circuit of fig. 2(a). Theoretical analysis. The Colpitts and Clapp oscillators were usually analyzed using afeedback system approach [2]. An alternative approach providing more insight into the oscillationphenomenon can be based on the concept of “negative resistance” [3], [8]. This concept isillustrated in fig. 3. Fig. 3. RLC tank with negative resistance created by the oscillator active network As seen from fig. 3, for overcoming the energy loss from a finite RP, active circuit must forma small-signal negative resistance Rneg  0  , which is required to replenish the energy loss everycycle of oscillation. Thus the negative resistance can be interpreted as a source of energy. Here, RPdenotes the equivalent parallel resistance of the tank and, for oscillation start-up, it is necessary that RP  Rneg  0 . As the oscillation amplitude increases, the amplifier will start to saturate decreasingthe loop gain until it reaches unity, thereby satisfying Barckhausen criterion [2]. In the steady statethe two resistances must be of equal amplitude, that is Rneg  RP . For determining a negative resistance, the input impedance of the circuit shown in fig. 2 (b)will be derived. An ac equivalent circuit of fig. 2 (b), ignoring the inductor L, for calculating theinput impedance is shown in fig. 4, where C gd is the capacitance between gate and drain, C gs is thecapacitance between gate and source and C ds is the capacitance between drain and source.
  3. 3. 84 ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) Fig. 4. An ac equivalent circuit of the Clapp oscillator ignoring the inductor L and capacitor C0 and indicating the transistor terminal parasitic capacitances Assuming Z S  1 j  C2  Cds  , rds  1 j  C2  Cds  , rgs  1 j  C1  C gs  ,RG  1 jC gd , and replacing the transistor Q by its simplified small signal model we obtain theequivalent circuit shown in fig. 5, where rds and rgs are, respectively, small-signal drain-source andgate-source resistances, g m is the small-signal transconductance of the transistor. Fig. 5. Small-signal equivalent circuit of the Clapp oscillator From fig. 5 the input impedance can be written as Z in  vin iin . (1) The loop equations are given by vin  i2 X C 1,Cgs  X C 2,Cds   g m v gs X C 2,Cds ; (2) v gs  i2 X C 1,Cgs , (3)where X gd i2  iin . (4) X gd  X C 1,Cgs  X C 2,Cds Substituting i2 from (4) into (2) and (3) gives X gd X C 1,Cgs  X C 2,Cds  vin  iin  g m v gs X C 2,Cds ; (5) X gd  X C 1,Cgs  X C 2,Cds X gd X C 1,Cgs v gs  iin . (6) X gd  X C 1,Cgs  X C 2,Cds
  4. 4. ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) 85 Now by substituting v gs from (6) into (5) we obtain X gd X C 1,Cgs  X C 2,Cds  X gd X C 1,Cgs X C 2,Cds vin  iin  iin g m . (7) X gd  X C 1,Cgs  X C 2,Cds X gd  X C 1,Cgs  X C 2,Cds Taking the ratio of vin to iin in (7) then yields X gd Z in  X C1,Cgs  X C 2,Cds  g m X C1,Cgs X C 2,Cds  . (8) X gd  X C 1,Cgs  X C 2,CdsSince X gd  1 jC gd , X C1,Cgs  1 j  C1  C gs  and X C 2,Cds  1 j  C2  Cds  , then we have from (8): 1  1 gm  Z in    2 , (9) C gd C gd  jCS   C1  C gs   C2  Cds   1    C1  C gs C2  Cdswhere C S  C1  C gs C 2  C ds  C1  C gs  C 2  C ds  . From (9) follows that the input impedance of the circuit shown in fig. 5 is represented by aseries connection of a negative resistance   C gd Cgd   rneg   g m  2  C1  C gs   C2  Cds  1   , (10)  C1  C gs C2  Cds       with a capacitance  C gd C gd  C SP  C S 1   C C  . (11)  1 gs C 2  C ds   It should be noted that with C gs  C gd  C ds  0 , eq. (10) is reduced to a well-knownexpression [3], [9] rneg   g m 2C1C2 . Connecting a coil L with the capacitor C 0 and a series total loss resistance rt to the inputimpedance Z in we obtain a series equivalent circuit of the Clapp oscillator shown in fig. 6 (a). a b Fig. 6. Series equivalent circuit of the Clapp (a) and Colpitts (b) oscillator As seen from fig. 6(a), the total tank circuit capacitance is given by  Cgd C gd  0,5CV CS  1    C1  C gs C2  Cds  Ct    ; (12)  C gd C gd  0,5CV  CS  1    C1  C gs C2  Cds   
  5. 5. 86 ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) Substituting Ct from (12) into the frequency setting equation 2  1 LCt , (13)gives 1   C gd C gd    0,5LCV CS  1    C1  C gs C2  Cds   2      . (14)   0,5C  C 1  C gd Cgd   S      V  C1  C gs C2  Cds    After substituting 2 from (14) to (10), the final expression for rneg can be represented as  gm L rneg  . (15)   C gd C gd  C1  C gs  C2  Cds 1  2 CS 1   C C     CV  1 gs C 2  C ds   From (15) follows that rneg depends on the transistor parasitic capacitances and ratio C S CV .As seen, the greater is ratio C S CV the less will be rneg . Figure 7 illustrates the dependence ofrneg on the ratio C S CV calculated with g m  3mS, L  100nH, C1  C2  20pF, C gs  C gd  2 pF,and Cds  0, 25pF . Fig. 7. Dependence of rneg on the ratio C S CV As it could be observed in fig. 7, big ratios of CS to CV result in small magnitude of rneg . Thisimpedes the start-up for the Clapp VCO. Figure 6 (b) shows the series equivalent circuit of the Colpitts oscillator. For this circuit thenegative resistance and total tank circuit capacitance are determined as follows:   C gd C gd  rneg   g m  2  CV  C gs   CV  Cds  1    ; (16)  CV  C gs CV  Cds      Ct  CV C gs CV  C ds   1  C gd  C gd  . (17) 2CV  C gs  C ds  CV  C gs CV  C ds    By substituting (17) into (13), the oscillation frequency is found to be 1   CV  C gs   CV  Cds   C gd Cgd  2   L 1    (18)  2CV  C gs  Cds  CV  C gs CV  Cds    
  6. 6. ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) 87 Substituting (18) into (16) then yields gm L rneg   . (19) 2CV  C gs  C ds According to (19), the negative resistance is inversely proportional to CV . Moreover, for theColpitts VCO resistance rneg does not depend on parasitic capacitance C gd , but it depends onC gs and Cds , which reduce the magnitude of rneg . The dependence of rneg on the value of varactor capacitance CV for the Clapp and ColpittsVCOs is illustrated in fig. 8 (a) and (b) respectively for the same data as in fig. 7. a b Fig. 8. Dependence of rneg on the varactor capacitance value CV for Clapp (a) and Colpitts (b) VCO Comparing the plots shown in fig.8, one can conclude that the Clapp VCO has an inherentlymore difficult start-up condition than the Colpitts VCO because rneg Colpitts  rneg Clapp . Therefore, ahigher small-signal transconductance and greater power consumption are required for the Clapposcillator to achieve reliable start-up. Referring to the equivalent circuits of fig. 6 we find that to enable oscillation, the negativesmall-signal resistance rneg added by the transistor should overcome the losses represented by rt .Taking into account that any VCO operates in a fixed frequency range where the varactorcapacitance changes from CV min to CV max , the guaranteed start-up condition for both VCOs in termsof the series equivalent circuits of fig. 6 is given by rneg  rt , (20) minwhere rneg is the minimum magnitude of the negative resistance in the selected frequency range. minAs seen from Fig. 8, rneg corresponds to CV min for the Clapp VCO and to CV max for the Colpitts minVCO. In the steady state a nonlinear mechanism reduces gm to a large-signal transconductance GmGm  g m  so that at any oscillation frequency rneg  rt .
  7. 7. 88 ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) For both Clapp and Colpitts VCOs the total series loss resistance rt is found as rt  rL  2 rVwhere rL and rV are, respectively, the coil and the varactor diode series resistance. The start-up condition for both VCOs can also be represented in terms of the parallelequivalent circuit shown in Fig. 3. Indeed, performing a series to parallel transformation thenegative resistance loading the parallel tank circuit is determined as   C gd C gd  Rneg   2CV  C gs  C ds   g m CV  C gs CV  C ds 1  2   , (21)  C C CV  C ds     V gs for the Colpitts VCO. Analogically, the parallel negative resistance Rneg can be derived for the ClappVCO. The equivalent parallel resistance of the tank for the Colpitts VCO topology can easily befound as L RP  . rt CV  C gs CV  Cds  1  C gd  C gd    2CV  C gs  C ds  CV  C gs CV  C ds    It is evident from the above given theoretical analysis that the VCO performancecharacteristics are dependent on the transistor parasitic capacitances C gs , C gd and C ds . For example,assuming g m  3mS, CV  10pF, C gs  C gd  2 pF, and Cds  0, 25pF one can calculate from (21)that Rneg  985 , while for the ideal VCO model we have Rneg   4 g m  1333 . Thusneglecting the transistor parasitic capacitances a 35 % error is introduced in estimation of Rneg .Therefore, at high frequencies the parasitic capacitances should be taken into account. Simulation results. To confirm the validity of theoretical modeling, we compare the resultsof calculations and SPICE simulations, performed with MULTISIM, for the Colpitts and Clapposcillator circuits shown in fig. 2. A JFET BF245B was selected for both oscillators with pinch-offvoltage VP  2,31V and I DSS  6, 2mA , where I DSS is the transistor drain-source current insaturation with zero gate-source voltage. For both oscillators the coil inductance was chosen to be100nH and VDD  5V . The source impedance circuit for both oscillators is shown in fig. 9. Fig. 9. The source impedance circuit The capacitances were selected as follows: CV 1  CV 2  CV  10pF – for the Colpittsoscillator and C1  C2  40pF, C0  9pF – for the Clapp oscillator. The dc value of the drain-sourcecurrent was calculated from the Shockley equation resulting in I DQ  1, 27mA . Using well-knownequations and SPICE file of JFET BF245B the other parameters were calculated giving thefollowing results: g m  2, 43mS, C gs  1,33pF, C gd  0,90pF and C ds  0 . The simulatedmaximum values of rt providing sustained oscillations at f  200, 2MHz and calculated values ofrneg for both oscillators are given in table.
  8. 8. ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) 89 Simulated and theoretical results Simulated value Theoretical value rt max (Ω) rneg (Ω) Colpitts Clapp Colpitts Clapp oscillator oscillator oscillator oscillator 9,3Ω 0,5 -11,4 -0,54 As seen from table 1, the Colpitts oscillator can provide sustained oscillations with maximumtotal loss resistance of 9,3Ω, while the Clapp oscillator can do the same with only rt max =0,5Ω. Thusthe Colpitts oscillator has an inherently more powerful starting-up than the Clapp oscillator. Itshould be noted that for both oscillators rneg  rt max , which proves the validity of the start-upcondition (20). The time required to reach a steady-state oscillation level  ss  is an important performancecharacteristic for any oscillator. Figure 10 illustrates the Colpitts (a) and Clapp (b) oscillatorstarting waveform for case when rt  0, 25 . As seen, the output voltage amplitude is within 3 dBof the steady-state output level in approximately 350 nS and 5,8 μS correspondingly for the Colpittsand Clapp oscillator. The fact that  ss Colpitts    ss Clapp once again demonstrates that the Clapposcillator has less powerful start-up than the Colpitts oscillator. Therefore, the Clapp oscillator canonly be designed with a high-Q inductors and varactors rising in price of fabricated VCOs. Whilethe Colpitts oscillator can be designed with low-Q and low-cost components. This is very importantadvantage of the Colpitts VCO arising from the results of this paper. a b Fig. 10. Starting waveform: a – Colpitts oscillator; b – Clapp oscillator Conclusions. Unlike the traditional approach of modeling the Colpitts and Clapp oscillatorswith neglected parasitic elements, the proposed mathematical model takes into consideration thetransistor terminal capacitances as well. For both series and parallel equivalent circuits of theoscillators, the analytical expressions have been derived for precise prediction of the oscillationfrequency and small-signal negative resistance. These expressions have been verified by SPICEsimulations. For the first time it was shown that the Colpitts oscillator has inherently more powerfulstart-up condition than the Clapp oscillator. From this fact follows that the Colpitts voltage-controlled oscillators can be designed with low-cost inductors and varactors, while the Clapposcillators require more expensive components.
  9. 9. 90 ISSN 1990-5548 Електроніка та системи управління. 2009. №1(19) References 1. Rhea R. W. Oscillator design and computer simulation. – 2nd ed. – Atlanta: Noble, – 2008. – 303 p. 2. Leenarts D., Van der Tang J., Vaucher C. Circuit design for RF transceivers. – Boston: Kluwer. Academic Publishers, – 2001. – 311 p. 3. Razavi B. Design of Analog CMOS Integrated Circuits. – Boston: McGraw–Hill, 2001. – 684 p. 4. Van der Tang J., Kasperkovitz D., Van Roermund A. High-frequency oscillator design for integrated transceivers. – London: Springer: 2003. – 343 p. 5. Aparicio R., et al. A Noise-Shifting Differential Colpitts VCO // IEEE J. Solid-State Circuits. – 2002. – Vol. 37. – P. 1728 – 1736. 6. Hegazi E., Rael J., Abidi A. The designer`s guide to high purity oscillators. – London: Springer, – 2004. – 204 p. 7. Odyniec M. RF and microwave oscillator design. – Artech House Publishers, – 2002. – 416 p. 8. Trimless IF VCO: Part 1: Design considerations // Application note 2032. – http: // www.maxim-ic.com/an 2032. 9. O`Connor C. Develop a trimless voltage-controlled oscillators // Microwaves and RF. – January 2000. – P. 94 – 105.В. В. Уланский, М. Ф. Абусаид, И. А. МачалинМатематическое моделирование генераторов, управляемых напряжением с топологиейКолпица и КлапаПредставлено аналитическое моделирование генераторов управляемых напряжением cтопологией Колпица и Клапа, из которого выведены более точные уравнения для расчетачастоты генерируемых колебаний и отрицательного сопротивления в режиме малогосигнала. Моделирование выполнено с учетом паразитных емкостей полевого транзистора.Дается сравнение условий самовозбуждения для генераторов Колпица и Клапа,подтвержденное SPICE имитацией. Приведенные уравнения гарантируют более точноемоделирование, проектирование и изготовление генераторов управляемых напряжением стопологией Колпица и Клапа.В. В. Уланський, М. Ф. Абусаід, І. О. МачалінМатематичне моделювання генераторів, керованих напругою з топологією Колпіця таКлапаПодано аналітичне моделювання генераторів керованих напругою з топологією Колпиця таКлапа, з якого виведено більш точні рівняння для розрахунку частоти генерованих коливаньта негативного опору в режимі малого сигналу. Моделювання виконано з урахуваннямпаразитних ємностей польового транзистора. Подано порівняння умов самозбудження длягенераторів Колпіця та Клапа, підтверджуване SPICE імітацією. Наведені рівняннягарантують більш точне моделювання, проєктування та виготовлення генераторів керованихнапругою з топологією Колпіця та Клапа.

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