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- 1. IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011 897 A High-Efﬁciency PV Module-Integrated DC/DC Converter for PV Energy Harvest in FREEDM Systems Zhigang Liang, Student Member, IEEE, Rong Guo, Member, IEEE, Jun Li, Student Member, IEEE, and Alex Q. Huang, Fellow, IEEE Abstract—The future renewable electric energy delivery andmanagement (FREEDM) system provides a dc interface for alter-native energy sources. As a result, photovoltaic (PV) energy can beeasily delivered through a dc/dc converter to the FREEDM system’sdc bus. The module-integrated converter (MIC) topology is a goodcandidate for a PV converter designed to work with the FREEDMsystem. This paper compares the parallel connected dc MIC struc-ture with its counterpart, the series connected MIC architecture.From the presented analysis, the parallel connected architecturewas shown to have more advantages. In this paper, a high-efﬁciencydual mode resonant converter topology is proposed for parallel con-nected dc MICs. This new resonant converter topology can changeresonant modes adaptively depending on the panel operation con-ditions. The converter achieves zero-voltage switching for primary-side switches and zero-current switching for secondary-side diodesfor both resonant modes. The circulation energy is minimized par-ticularly for 5–50% of the rated power level. Thus, the convertercan maintain a high efﬁciency for a wide input range at differentoutput power levels. This study explains the operation principle ofthe proposed converter and presents a dc gain analysis based onthe fundamental harmonic analysis method. A 240-W prototype Fig. 1. Part of the FREEDM system diagram.with an embedded maximum power point tracking controller wasbuilt to evaluate the performance of the proposed converter. The predicted to become the biggest contributors to electricity gen-prototype’s maximum efﬁciency reaches 96.5% and an efﬁciency eration among all renewable energy generation candidates byincrease of more than 10% under light load conditions is shown 2040 [2], [3]. In 2009, almost 7.5 GW of new PV capacity waswhen compared with a conventional LLC resonant converter. added worldwide and it is expected that the global installed PV Index Terms—DC-DC power converters, photovoltaic systems, capacity could reach 10 GW in 2010 [4].smart grid, solar power generation. The large-scale utilization of renewable energy depends on an advanced smart grid infrastructure where the users have the ability to manage their energy consumption as well as use plug- I. INTRODUCTION and-generate and plug-and-store energy devices at home and HE global demand for electric energy has continuouslyT increased over the last few decades. Energy and the en-vironment have become serious concerns in today’s world [1]. in industrial applications [5], [6]. The future renewable electric energy delivery and management (FREEDM) system is an in- telligent electric power grid integrating highly distributed andAlternative sources of energy generation have drawn more and scalable alternative generating sources and storage with exist-more attention in recent years. Photovoltaic (PV) sources are ing power systems to facilitate a renewable energy-based soci- ety [5]. The 400-V dc bus in the FREEDM system provides an alternative interface for PV converters. Fig. 1 shows part of the Manuscript received July 1, 2010; revised January 9, 2011; accepted January FREEDM system including an Intelligent Energy Management10, 2011. Date of current version May 13, 2011. Recommended for publication (IEM) module. As a result, PV converters in a FREEDM sys-by Associate Editor J. M. Guerrero. tem only need to have a dc/dc stage to interface with the dc bus. Z. Liang and A. Q. Huang are with the Future Renewable Electric EnergyDelivery and Management (FREEDM) Systems Center, Department of Electri- Generally, this structure has several advantages.cal and Computer Engineering, North Carolina State University, Raleigh, NC 1) Since the solid state transformer (SST) is the component27695 USA (e-mail: zliang2@ncsu.edu; aqhuang@ncsu.edu). interfacing with electric grid, the PV converters’ controller R. Guo is with the International Rectiﬁer Rhode Island Design Center,Warwick, RI 02818 USA (e-mail: rguo1@irf.com). does not require a phase locked loop, current regulator, or J. Li is with the ABB U.S. Corporate Research Center, Raleigh, NC 27606 anti-islanding controller. Thus, the control task becomesUSA (e-mail: jun.li@us.abb.com). much simpler. Color versions of one or more of the ﬁgures in this paper are available onlineat http://ieeexplore.ieee.org. 2) The PV converter can be comprised of a single power Digital Object Identiﬁer 10.1109/TPEL.2011.2107581 stage. 0885-8993/$26.00 © 2011 IEEE
- 2. 898 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011 tipartial cloud capability, and the fact that any single failure of an MIC will not impact any other part of the system. As a result, MICs in a parallel conﬁguration have higher fault tolerance and reliability that make them more promising for PV application in a FREEDM system. However, the high gain requirement usually compromises its efﬁciency. The topologies suitable for this application can be categorized into two groups: nonisolated topologies and isolated topologies. For nonisolated topologies, boost, buck–boost, zeta, cuk, or theirFig. 2. Two types of dc MIC structure: (a) parallel connection and (b) series derivatives [23]–[32] are commonly used. Isolated topologiesconnection. mainly include ﬂyback [33]–[39], current-fed push–pull [40], [41], and resonant converters [42], [43]. The typical maximum Therefore, it is very possible to reduce the system cost for end efﬁciency of these converters is around 80–97% [10]–[12], [19].users. At present, signiﬁcant research effort has been made to Among these topologies, the half-bridge LLC resonant con-improve the performance of PV converters [7]–[9]; PV module- verter is a good candidate due to its several unique advan-integrated converters (MICs) are gaining increasing amounts of tages [44]–[46]. However, it is difﬁcult for an LLC resonantattention due to their distinctive features [10]–[20]. converter to maintain high efﬁciency for a wide input range un- 1) The MIC is an integrated part of the PV panel. MICs re- der different load conditions. In this paper, a new resonant dc/dc move losses due to the mismatch between panels and sup- converter with dual operation modes is proposed. By chang- port panel level maximum power point tracking (MPPT). ing operation modes adaptively according to VPV and PPV , the For a string inverter or a centralized inverter, a string or converter’s efﬁciency is improved. multistring of PV panels shares a single MPPT controller, but the mismatch loss is serious in partial shading condi- tions [21]. Considering the mismatch loss together with III. OPERATION PRINCIPLE OF THE NEW the dc/ac conversion loss contributing to the whole PV RESONANT CONVERTER system loss, string/centralized inverters may have lower system efﬁciency than MICs due to higher mismatch loss Fig. 3 shows a circuit diagram of the proposed resonant con- although they usually have higher dc/ac conversion efﬁ- verter. S1 and S2 are two power MOSFETs; DS 1 , CS 1 and ciency than MICs. DS 2 , CS 2 are the body diodes and parasitic capacitances of S1 2) Panel level hot-spot risk is removed [11] and panel life- and S2 , respectively. Cr is the resonant capacitor; Lr and Lm time can be improved. Hot spot takes place when a shaded are the magnetizing inductance of transformers Tx2 and Tx1 , cell within a partially shaded panel becomes reverse bi- respectively. Llkg is the sum of the leakage inductance of Tx1 ased and dissipates power in the form of heat [22]. For and Tx2 . D1 , D2 and Co1 , Co2 form a voltage doubler at the series connected PV panels used with a string/centralized secondary side of Tx1 . A half-wave rectiﬁer (HWR) formed by inverter, a by-pass diode is added to each panel in practice. D3 , S3 , D4 , and CO 3 is added to the secondary side of trans- For the MIC solution, the by-pass diode is not necessary former Tx2 . Diode D3 blocks the conductive path of the body because each panel has its own MIC, leading to no direct diode of S3 . Thus, D3 and S3 form a unidirectional switch to en- connection between PV panels. able or disable the HWR. When the HWR is enabled, the HWR 3) Its “plug and play” feature simpliﬁes system installation. and voltage doubler will support the 400-V dc bus with their In summary, the MIC solution allows for more ﬂexible PV summed outputs. Table II summarizes the operation modes forproject planning and multifacet PV panel installation. the proposed converter and Vth is a predeﬁned threshold volt- age that is usually equal to the nominal voltage Vnom . For the ﬁrst three operation conditions listed in Table II, the HWR is II. COMPARISON OF MICS IN SERIES AND disabled by turning off switch S3 . As a result, the converter PARALLEL CONNECTIONS behaves like a traditional LLC resonant converter with a voltage Both dc MICs and ac MICs are available in the market. Only doubler [46]: an equivalent resonant inductor Lr , comprised ofdc MICs will be discussed in this paper, as they are suitable for Lr and Llkg , participates in the resonant circuit formed by Lmthe FREEDM system. As shown in Fig. 2, dc MICs have two and Cr . Diode D4 is conducting to provide a path for the loadkinds of connection structures. Fig. 2(a) shows a type I dc MIC current. Once VPV is smaller than Vth and PPV is lower thanconﬁguration, consisting of multiple parallel connected MICs 50% of the rated power (Prated ), the PV panel is working underdirectly interfaced with a dc bus. Type II dc MICs, shown in condition #4 and the converter will operate in Mode II.Fig. 2(b), need to form a series connection to obtain a voltage For one switching period, the operation of the converter inhigh enough for interfacing with the dc bus. Generally, the power Mode II can be divided into nine stages. The equivalent circuitrating of both types of dc MICs is around 200 W–300 W. for each stage is shown in Fig. 4 and its key waveforms are The two system structures have different features. Table I depicted in Fig. 5. For the description of circuit operation (andsummarizes the comparison results of the two MIC structures: for the subsequent dc gain derivation in the next section), thethe parallel connection is more ﬂexible due to its stronger an- following assumptions are made.
- 3. LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 899 TABLE I COMPARISON OF TWO TYPES OF DC MIC STRUCTUREFig. 3. Circuit diagram of the proposed resonant converter. TABLE II 3) The turn ratio NT X 2 (Npri : Nsec ) of transformer TX 2 isSUMMARY OF OPERATION MODES FOR THE PROPOSED RESONANT CONVERTER the half of NT X 1 . Deﬁne NT X 2 = 1/2 NT X 1 = N . The operation processes of Mode II are speciﬁed as follows. Stage 1 (t0 –t1 ): When S2 is turned off at t = t0 , stage 1 be- gins. Since Ipri is negative, capacitor Cs2 (Cs1 ) will be charged (discharged) and the switching node voltage Vsw will increase accordingly. Inductors Lm , Lr , and Llkg are all in resonance with Cr . Vcr continues to decrease and no current ﬂows through the secondary side of either transformer. The output capacitors 1) All the components are ideal. The body diodes and par- Co1 , Co2 together with Co3 supply the load current and VC o1 – asitic capacitance of S1 and S2 have been taken into ac- VC o3 all decrease in this period. count. The output capacitors have equal values (Co1 = Stage 2 (t1 –t2 ): At time t = t1 , Vsw reaches Vpv . Ds1 is Co2 = Co3 ). forward biased and starts to conduct a current Ipri . Ipri starts 2) Inductor Llkg includes the leakage inductance of TX 1 and to decrease. Once Ipri becomes smaller than the magnetizing TX 2 ; it also includes the wire parasitic inductance. currents IL r and IL m , the resonance of [Lm , Lr , Llkg ] and Cr
- 4. 900 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011Fig. 4. Equivalent circuits for each operation stage (Mode II operation).is stopped. Lr and Lm will be out of the resonance following changes its direction at t = t3 . The leakage inductor Llkg stillthis. The difference between Ipri and IL m will ﬂow in the sec- resonates with Cr , and Ipri keeps increasing. The magnetizingondary side of Tx1 . Similarly, the secondary side of Tx2 will currents IL r and IL m continue to increase with the same slopeconduct the current difference between Ipri and IL r . Thus, the as in Mode 2. The rectiﬁer diodes D1 and D3 conduct currentvoltage across the primary side of Tx1 and Tx2 is clamped by and power is delivered to the load. This stage ends when Ipri isVout . IL r and IL m start to decrease linearly. equal to IL m . Stage 3 (t2 –t4 ): This stage begins when S1 is turned on at t = Stage 4 (t4 –t5 ): At t = t4 , Ipri and IL m are equal. The outputt2 . At this moment, the primary-side current Ipri is negative and current of the transformer Tx1 reaches zero. Transformer Tx1 ’sﬂows through the body diode of S1 . Thus, ZVS turn on of S1 secondary voltage is lower than the output voltage. The outputcan be achieved at t2 . The current Ipri continues to decrease and is separated from transformer Tx1 . Meanwhile, since Ipri is still
- 5. LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 901 balance of the transformers Tx1 and Tx2 has still been preserved. Further, if a full-wave rectiﬁer (FWR) is added instead of the HWR, Ipri will become symmetrical and the other character- istics of the converter will remain. The theoretical analysis of the aforementioned Mode II operation has been veriﬁed by the simulation with Simetrix. Fig. 6 shows the simulation results of the proposed converter with following operation conditions: Vpv = 22 V, Vout = 400 V, Pout = 120 W (50% of Prated ), fs = 83 kHz. IV. DC GAIN ANALYSIS FOR THE PROPOSED CONVERTER OPERATION IN MODE II Understanding of the dc gain characteristic for a resonant con- verter has equal importance as knowing its operation principle. Since the dc gain characteristic for Mode I operation is the same as LLC resonant converter, only Mode II operation requires a new analysis to be developed. The fundamental harmonic analy- sis (FHA) method is widely used for dc gain analysis of resonant converters [47]–[50] and it is also valid for the analysis devel-Fig. 5. Key waveforms of the proposed converter (Mode II operation). oped in this paper. This approach is based on the assumption that the power transfer from the source to the load through the resonant tank is almost completely dependent on the fundamen-larger than IL r , the output current of Tx2 is not zero and power tal harmonic of the Fourier expansion of the currents and theis delivered to the load through Tx2 . During this stage, Lm voltage involved. The voltage at the input of the two rectiﬁersparticipates into the resonance again and the resonance between Vosq (t) can be expressed as[Llkg , Lm ] and Cr begins. Stage 5 (t5 –t6 ): Switch S1 is turned off at t = t5 . The current Vosq (t) = Vab (t) + Vcd (t) (1)Ipri is positive and switching node voltage will decrease due to where Vab (t) and Vcd (t) are the secondary-side terminal volt-charging (discharging) of Cs1 (Cs2 ). ages of transformers TX 2 and TX 1 (see Fig. 3). Like the con- Stage 6 (t6 –t7 ): At time t = t6 , Vsw drops to zero that causes ventional LLC resonant converter, the current in the secondarythe conduction of the body diode Ds2 . With the drop of Vsw , the side is quasi-sinusoidal and the voltage Vosq (t) reverses whenvoltage applied to Lm (VL m ) decreases to zero and continues to the current becomes zero. Therefore, Vosq (t) is an alternativebecome more negative. Once VL m is higher than a certain level, square wave in phase with the rectiﬁer current. The Fourierdiode D2 on the secondary side of Tx1 will be forward biased. expression of Vosq (t) isThus, the voltage applied to Lm is clamped and IL m will droplinearly. Lm is out of resonance with Cr . Instead, only Llkg 4 1 Vosq (t) = Vout sin(n2πfsw t). (2)resonates with Cr and Ipri decreases steeply. This stage ends π n =1,3,5,... nwhen IL r is equal to Ipri . For convenience, the phase angle of Vosq (t) is assumed to be Stage 7 (t7 –t8 ): At time t = t7 , IL r is equal to Ipri ; no zero in (2). Its fundamental component Vo FHA (t) ismore current will ﬂow in the secondary side of Tx2 . The outputis separated from Tx2 . D3 is turned off with ZCS. The voltage 4 Vo FHA (t) =Vout sin(2πfsw t). (3)applied to Lr is not clamped and Lr participates in the resonance πagain with Cr and Llkg . The current Ipri is positive and continues The rms amplitude of Vo FHA (t) isto ﬂow through Ds2 , which creates the ZVS condition for S2 if √ 2 2S2 is turned on at this moment. Vo FHA = Vout . (4) Stage 8 (t8 –t10 ): At t = t8 , S2 is turned on with ZVS. The πcurrent Ipri continues to decrease due to the resonance between Deﬁne the fundamental part of the rectiﬁer current to be √[Lr , Llkg ] and Cr . The transformer Tx1 delivers power to the irect (t) = 2Irect sin(2πfsw t). (5)output. This stage ends when current Ipri = IL m . Stage 9 (t10 –t11 ): At t = t9 , Ipri = IL m . No more current will The phase angle of Irect is also zero since it is in phase withﬂow in the secondary side of Tx1 . The voltage applied to Lm Vo FHA (t). Thus, the average value of Iout can be calculated asis not clamped anymore and Lm participates in the resonance TSW √ 2 2 2 2Irectagain with Lr , Llkg , and Cr . At t = t11 , S2 is turned off and a Iout = irect (t)dt = . (6) TSW 0 πnew switching cycle begins. From the aforementioned analysis, the energy transferred by Iout can be expressed asTx1 and Tx2 is different. The positive and negative parts of the Voutcurrent Ipri are not symmetrical. However, the voltage-second Iout = . (7) Rout
- 6. 902 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011Fig. 6. Simulation results of the proposed converter operating in Mode II.Fig. 7. Equivalent FHA resonant circuit model for the proposed converter operation in Mode II. Equation (8) can be derived by combining (6) and (7) as substituted by an equivalent transformer Txe with turn ratio N .follows: The resulting expression for the dc gain of the converter can be √ derived through a circuit analysis based on the model in Fig. 7. 2πVout Irect = . (8) Deﬁne the dc gain 4Rout Insert (8) into (5) N Vo FHA M= . (11) √ Vi FHA √ 2πVout πVoutirect (t) = 2 sin(2πfsw t) = sin(2πfsw t). 4Rout 2Rout Consider (9) Vdc 2 1 The equivalent ac output impedance Ro ac can be derived by VSW (t) = + Vdc sin(n2πfsw t). (12)combining (4) and (8) as follows: 2 π n =1,3,5,... n Vo FHA 8Rout Ro ac = = . (10) vi FHA (t) is the fundamental part of VSW (t) Irect π2 The expression for Ro ac is the same as the one for a conven- 2 vi FHA (t) = Vdc sin(2πfsw t). (13)tional LLC resonant converter. With the known Ro ac , the equiv- πalent FHA resonant circuit model can be obtained, as shown in Vi FHA can be derived as follows:Fig. 7. In this model, Vi FHA is the rms value of the fundamental √ 2component of the voltage at the switching node SW (VSW ). The Vi FHA (t) = Vdc . (14)voltage VSW is generated by the controlled switches S1 and S2 . πThe output current Iout is produced from Irect after the rectiﬁer Combining with (4), (11), and (14), the input-to-output volt-network and ﬁlter capacitors. From a turn ratio perspective, the age conversion ratio isconversion gain of a transformer with turn ratio 2N followedby a voltage doubler is equal to a transformer with turn ratio N . Vout 1Therefore, transformer Tx1 together with voltage doubler can be = |M | . (15) Vdc 2N
- 7. LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 903 From the FHA model, Zout is the impedance seen from theprimary side of the two transformers N 2 Ro ac · Lm r · S Zout = (16) N 2 Ro ac + Lm r · Swhere Lm r = Lm + Lr . The dc gain M can be derived asfollows: Zout M (S) = . (17) (1/S · Cr ) + S · Llkg + Zout By substituting S = j2πfSW , the amplitude of M (S) is, asshown (18), at the bottom of this page. For convenience, (18) can be rewritten as 1 M (fn ) = . (19) (1 + λ − (λ/fn 2 ))2 + Q2 (f − (1/f ))2 n n Fig. 8. Series of example of dc gain curves of a new resonant converter with The parameters in (19) are deﬁned as follows: different Q value (Mode II). 1 fr = (20) 2π Llkg · Cr Z0 Q= (21) N2 · Ro ac Llkg λ= (22) Lm + Lr Llkg Z0 = (23) Cr f SW fn = . (24) fr Equations (19)–(24) reveal the dc gain characteristics forMode II operation. It is interesting that Mode II operation has Fig. 9. Series of example of dc gain curves for a new resonant converter withsimilar dc gain expression to Mode I but with different parame- different Q value (Mode I).ters for the resonant tank. A series of example of dc gain curvesof Mode II operation under different load conditions (with differ- V. DC GAIN VERIFICATION AND COMPARISONent Q values) are plotted in Fig. 8. For very light load conditions(small Q), the gain has a large peak. On the contrary, the gain To verify the dc gain expression derived in section IV, abecomes ﬂat under heavy load conditions (large Q). Similar to series of simulations have been performed for different Vpv foran LLC converter, the dc characteristic of Mode II operation a given load condition. The converter’s switching frequency fscan be divided into ZVS and ZCS regions, and the converter is recorded. Equation (19) is used to calculate the dc gain resultshould be prevented from entering the ZCS region. With proper at a given fs for the same operation condition. Through thechoice of the resonant tank, Mode II operation can stay in the comparison between the dc gain from simulation (Msimulation )ZVS region for Vpv and Ppv variations. The ZVS region can be and the theoretical analysis result (Mcalculation ), the accuracyfurther divided into regions I and II due to slightly operation of (19) can be evaluated. Table III shows the comparison resultsdifferences. In practical designs, the converter has unity gain at for a 50% load condition where Msimulation is deﬁned byVpv = Vnom and the converter enters Mode II operation only Vout · N Msimulation = . (25)when Vpv ≤ Vnom . Therefore, it is impossible for the proposed Vpv /2resonant converter to work in region I after entering Mode IIoperation. Mode II operation can only be active in region II. From Table III, Mcalculation matches with Msimulation veryFurthermore, the discussion about Mode II operation in the last well. Therefore, (19) is accurate enough for engineering designsection is dedicated for region II. On the contrary, Mode I op- of the proposed converter. Furthermore, a comparison of the dceration can only be active in region I (see Fig. 9) because the gain between Mode I and II operations is conducted in order torequired dc gain should be lower than 1 in Mode I (Vpv > Vnom ). reveal the general dc gain features of the proposed converter. 32π 2 ·Cr ·Lm r ·Rout ·fSW ·N 2 2M= . (32π 2 ·Cr ·Lm r ·Rout ·fSW ·N 2 − 8·Rout ·N 2 + 32·π 2 ·Cr ·Llkg ·Rout ·fSW ·N 2 )2 + (−2·π 3 ·Lm r ·fSW + 8·π 5 ·Cr ·Lm r ·Llkg ·fSW )2 2 2 3 (18)
- 8. 904 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011 TABLE III TABLE IV DC GAIN COMPARISON BETWEEN SIMULATION AND CALCULATION LIST OF PARAMETERS OF THE PROPOSED CONVERTER FOR GAIN ANALYSISThe normalized frequency fn has a different base for Mode Iand II operations since they have different fr : fSW fn M o deI = , fr M o deI 1 where fr M o deI = (26) 2π (Llkg + Lr ) · Cr fSW 1 fn M o deII = , where fr M o deII = . fr M o deII 2π Llkg · Cr (27) For further analysis, fn needs to be uniﬁed using the samebase, for fn M o deI : fSW fr M o deII fn M o deI = · fr M o deII fr M o deI fr M o deII = fn M o deII · = α · fn M o deII . (28) fr M o deI Both the dc gain expressions for Modes I and II can be writtenas functions of fn M o deII , as shown (29) and (30), at the bottomof this page. Table IV gives the resonant tank parameters for example de-sign. For comparison, the equations for calculating several keyparameters are also listed in Table IV. The gain curves for thetwo operation modes can be plotted in the same ﬁgure, as shownin Fig. 10. Fig. 10. DC gain comparison between Modes I and II at 50% rated power. From Fig. 10, the two curves reach their peaks at the samefrequency fn M deﬁned by fM 1 2) The frequency difference becomes larger with higher inputfn M = = . voltage. Fig. 10 takes Vpv = 22 V and Vpv = 32 V as ex- fn M o deII 2π (Lr + Llkg + Lm ) · Cr · fn M o deII (31) amples. It shows the switching frequency almost doubles Similar to the LLC resonant converter, operation in the region if the converter operates in Mode II with 32-V input.where fn < fn M is forbidden. In the region fn M < fn < 3) The gain curve of Mode II becomes much ﬂatter at highf0 , MM o deI is always higher than MM o deII . On the contrary, frequency. The gain is almost constant and stops decreas-MM o deI becomes lower than MM o deII in region fn > f0 . For ing. Considering that higher Vm pp requires smaller dca desired dc gain in the latter region, the following conclusion gain, this implies that the PV panel voltage may be out ofcan be drawn. regulation in Mode II when Vm pp is too high. Therefore, 1) Mode II operation needs a higher switching frequency it is reasonable to keep the converter operating in Mode I than Mode I operation. when Vm pp is higher than a certain value. 1 MM o deI (fn M o deII ) = (29) (1 + λM o deI − (λM o deI /(α · fn M odeII )2 ))2 + Q2 o deI (α · fn M M o deII − (1/α · fn M odeII ))2 1 MM o deII (fn M o deII ) = . (30) (1 + λM o deII − 2 (λM o deII /fn M odeII ))2 + Q2 o deII (fn M M odeII − (1/fn M odeII ))2
- 9. LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 905 TABLE V TABLE VI CIRCUIT PARAMETERS FOR EXPERIMENT LOSS BREAKDOWN OF THE PROPOSED CONVERTER IN MODE II WITH 10% OF P ra te d (V pv ≤ 32 V) TABLE VII LOSS BREAKDOWN OF THE LLC CONVERTER WITH 10% OF P ra te d (V pv ≤ 32 V)Fig. 11. Efﬁciency improvement of the proposed converter in Mode IIoperation. VI. DESIGN EXAMPLE AND EFFICIENCY ANALYSIS The MIC will be operated with PV panels that normally haveVm pp of around 22–40 V. Vnom for this design is 32 V andPrated is equal to 240 W. The transformer primary side is thelow-voltage side and it has high resonant current circulating.In order to minimize the conduction loss, a 75-V MOSFET Fig. 12. System diagram for the experiment with a work ﬂow chart for thewith low Rdson is preferred and multistrand Litz wire should be dc/dc controller.used to reduce the ac resistance of the primary winding of thetransformer. There is no strict limitation on volume and size forMICs. Thus, a lower switching frequency fs (<200 kHz) canbe adopted to beneﬁt the converter efﬁciency. Table V gives component parameters for the MIC prototype.The threshold voltage Vth for operation mode decision is chosento be equal to Vnom . One can design Cr , Lr , Lm , and Tx1with a conventional design procedure for an LLC converter.Then, a secondary winding is added to Lr such that it formsthe transformer Tx2 . The devices D3 , D4 , and S3 in HWRhave the same current rating as D1 and D2 in voltage doubler.Considering that a practical transformer has a certain leakageinductance, the value of Llkg can be chosen to be 5–15% of(Lr + Lm ). A comprehensive loss analysis has been conducted to eval-uate the efﬁciency of the designed converter. For comparison,the efﬁciency of a traditional LLC resonant converter with thesame circuit parameters is also analyzed. Their efﬁciency dif- Fig. 13. Picture of a 240-W MIC prototype.ference is plotted in Fig. 11 for 5–50% of Prated . The efﬁciency
- 10. 906 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011Fig. 14. Waveforms of an MIC prototype: (a) Mode I (ch1: 10 V/div; ch4: 10 A/div; t = 4 μs) and (b) Mode II (ch1: 50 V/div; ch2: 200 V/div; ch3: 1 A/div;ch4: 10 A/div).Fig. 15. Waveforms to verify the ZVS operation in Mode II (ch1: 10 V/div; ch2: 20 V/div; ch4: 10 A/div). (a) V in = 22 V, 20% of P ra te d (verify upper sideswitch ZVS) and (b) V in = 22 V, 20% of P ra te d (verify lower side switch ZVS).improvement drops when Ppv increases. When Ppv approaches operation reduces the transformer core loss by causing smaller50% of Prated , the efﬁciency improvement is reduced to almost variation of the magnetic ﬁeld strength in a switching period.zero. Therefore, there is no beneﬁt to keep converter running As a result, the total loss is dramatically reduced by Mode IIin Mode II when Ppv > 50% of Prated and mode change is operation.required. To get a better understanding of the efﬁciency improvementin Mode II operation, a loss breakdown is conducted for both VII. EXPERIMENTAL RESULTSMode II operation and normal LLC operation with Vpv < 32 V An experimental prototype has been built to verify the per-and Ppv = 10% of Prated . Tables VI and VII give the analysis formance of the proposed converter. Fig. 12 depicts the systemresults. As discussed in the previous section, Mode II operation diagram for experiment and Fig. 13 shows a picture of the pro-will increase the switching frequency. Thus, the switching loss totype. An MPPT controller implemented in a microcontrollerof MOSFET may increase due to the increase in the number of will provide a reference voltage Vpv ref that will be used by theswitching events. However, the data in Table VI show a signiﬁ- dc/dc controller to determine the converter’s operation modecant decrease in the total switching loss. This is because higher based on the criteria described in Table II. The dc/dc controllerfrequency operation leads to a much lower resonant current will check Vpv and Ppv every few minutes and its operationthrough the MOSFET during its turn-off event. Due to the same follows the work ﬂow chart in Fig. 12.reason, the MOSFET conduction loss and transformer copper Fig. 14 shows the operation waveforms of MIC prototype inloss are also greatly reduced. Moreover, the higher frequency Modes I and II. In Mode II, only the positive part of current
- 11. LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 907 converter’s performance have been validated by the experiment results from a 240-W prototype. Future work includes the com- pletion of an advanced energy controller design for the MIC that can receive commands from the IEM and allows for a ﬂexible control of the power generation proﬁle. ACKNOWLEDGMENT The authors would like to thank Edward Van Brunt’s help during the manuscript revision. This work made use of ERC shared facilities supported by the National Science FoundationFig. 16. Measured efﬁciency improvements with HWR (Mode II) for 5–50% under Award Number EEC-0812121.of P ra te d (V pv ≤ 32 V). REFERENCES [1] B. K. Bose, “Global warming: Energy, environmental pollution, and the impact of power electronics,” IEEE Ind. Electron. Mag., vol. 4, no. 1, pp. 6–17, Mar. 2010. [2] European Renewable Energy Council (2004, May). Renewable Energy Scenario to 2040 [Online]. 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Zacharias, “New inverter technology and harmonic dis- tortion problems in modular PV systems,” in Proc. Eur. Photovoltaic Solar Energy Conf. Exhib., 1997, pp. 2207–2210. Zhigang Liang (S’10) was born in Sichuan, China,[32] J. M. A. Myrzik, “Novel inverter topologies for single-phase standalone or in 1981. He received the B.S. and M.S. degrees grid-connected photovoltaic systems,” in Proc. Power Electronics Drive in electrical engineering from Zhejiang University, Systems (PEDS), 2001, pp. 103–108. Hangzhou, China, in 2003 and 2006, respectively.[33] D. C. Martins and R. Demonti, “Photovoltaic energy processing for utility He is currently working toward the Ph.D. degree connected system,” in Proc. IEEE Ind.l Electron. Soc. (IECON), 2001, in the Future Renewable Electric Energy Delivery pp. 1292–1296. and Management (FREEDM) Systems Center, North[34] D. C. Martins and R. Demonti, “Grid connected PV system using two Carolina State University, Raleigh. energy processing stages,” in Proc. IEEE Photovoltaic Spec. Conf., 2002, From 2006 to 2007, he was a System Engineer with pp. 1649–1652. Monolithic Power Systems (MPS), Inc., Hangzhou,[35] T. Shimizu, K. Wada, and N. Nakamura, “A ﬂyback-type single phase China. His research interests include high-efﬁciency utility interactive inverter with low-frequency ripple current reduction on power conversion, micro inverters and MICs for Photovoltaic applications, and the DC input for an AC photovoltaic module system,” in Proc. IEEE Power energy management in dc microgrid. Electron. Spec. Conf. (PESC), 2002, pp. 1483–1488.[36] T. Shimizu, K. Wada, and N. Nakamura, “Flyback-type single-phase utility interactive inverter with power pulsation decoupling on the dc input for an ac photovoltaic module system,” IEEE Trans. Power Electron., vol. 21, no. 5, pp. 1264–1272, Sep. 2006.[37] N. P. Papanikolaou, E. C. Tatakis, A. Critsis, and D. Klimis, “Simpliﬁed high frequency converters in decentralized grid-connected PV systems: Rong Guo (M’10) was born in Hunan, China, in A novel low-cost solution,” in Proc. Eur. Conf. Power Electron. Appl., 1982. She received the B.S. degree in electrical engi- [CD-ROM], 2003. neering and automation from Xi’an Jiaotong Univer-[38] N. Kasa, T. Iida, and A. K. S. Bhat, “Zero-voltage transition ﬂyback sity, Xi’an, China, in 2003, the M.S. degree in power inverter for small scale photovoltaic power system,” in Proc. IEEE Power electronics from Zhejiang University, Hangzhou, Electron. Spec. Conf. (PESC), 2005, pp. 2098–2103. China, in 2006, and the Ph.D. degree in electrical[39] A. Fernandez, J. Sebastian, M. M. Hernando, M. Arias, and G. Perez, “Sin- engineering from North Carolina State University, gle stage inverter for a direct ac connection of a photovoltaic cell module,” Raleigh, in 2010. in Proc. IEEE Power Electron. Spec. Conf. (PESC), 2006, pp. 93–98. 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- 13. LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 909 Jun Li (S’07) was born in Liaoning, China, in 1981. Alex Q. Huang (S’91–M’94–SM’96–F’05) received He received the B.S. degree in automation from the B.Sc. degree in electrical engineering from Tianjin University, Tianjin, China, in 2004, the M.S. Zhejiang University, Hangzhou, China, in 1983, the degree in power electronics from Zhejiang Univer- M.Sc. degree in electrical engineering from the sity, Hangzhou, China, in 2006, and the Ph.D. degree Chengdu Institute of Radio Engineering, Chengdu, in power electronics from North Carolina State Uni- China, in 1986, and the Ph.D. degree from versity, Raleigh, in 2010. Cambridge University, Cambridge, U.K., in 1992. He is currently a Senior R&D Engineer in ABB From 1994 to 2004, he was a Professor with the U.S. Corporate Research Center, Raleigh, NC. His Center for Power Electronics Systems, Virginia Poly- research interests include topology and control of technic Institute and State University, Blacksburg. high-power multilevel converters for MV drives and Since 2004, he has been a Professor of Electricalrenewable energy generation. Engineering with North Carolina State University (NCSU), Raleigh, and the Director of NCSU’s Semiconductor Power Electronics Center. He is also the Progress Energy Distinguished Professor and the Director of the new National Science Foundation’s Engineering Research Center for Future Renewable Elec- tric Energy Delivery and Management Systems, Department of Electrical and Computer Engineering, North Carolina State University, Raleigh. His research areas are power management, emerging applications of power electronics, and power semiconductor devices. He has published more than 200 papers in jour- nals and conference proceedings, and holds 14 U.S. patents. Prof. Huang is the recipient of the NSF CAREER Award and the prestigious R&D 100 Award.

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