Welcome to the training module on Analog Devices High Speed Amplifiers - Part 1 . This training module introduces the basic knowledge of high speed operational amplifiers and some characteristics.
It is generally accepted that amplifiers with bandwidths greater than 50MHz are considered high speed (HS) amplifiers. High speed amplifiers tend to be more concerned with AC specifications, such as, wide bandwidth, fast slew rates and quick settling time. Precision amplifiers on the other hand have bandwidths less than 50MHz and are more concerned with DC specs, such as offset voltage and drift. High speed amplifiers are used in applications such as communications, video, medical imaging, commercial imaging, consumer electronics, industrial and data acquisitions.
Here’s a few typical High-speed amplifiers specifications. The Bandwidth range is from 50MHz to 2.1GHz; The Slew rate is from 30V/ µ sec to 5500V/ µ sec; and finally The Settling Time is from 1 µ sec to 4nsec.
Before examining the input and output circuit structures of op amps, we must first understand the two popular op amp topologies and their differences. The two popular op amp topologies are voltage feedback-VFB and current feedback-CFB. This is a simple model of a VFB op amp. This topology has two high impedance inputs and an internal source of large voltage gain or A(s), which is a function of frequency. With feedback applied as shown, the small differential input error voltage is equal to v. Large values of A(s) produces small values of v. The classic equation which relates the output voltage to the input voltage is shown and can be derived using simple algebra. The second equation shows that the term 1/[1+R2/R1] is defined as the feedback factor, β. For large values of A(s), the gain is simply 1+R2/R1.
This is the classic gain versus frequency plot on a log-by-log scale for a VFB op amp. If the op amp has a single-pole response, which by the way is desirable for good transient response and stability, then the open-loop transfer function A(s) rolls off at 6dB/octave above the corner frequency. The noise gain, 1+R2/R1, intersects the open-loop gain at the closed-loop bandwidth, f CL . Over most of the usable range of closed-loop gain, the product of the noise gain and the closed-loop bandwidth is constant and equal to the product of gain and bandwidth of the op amp. The quantity; 1+R2/R1 is referred to as noise gain because it is the gain a small voltage noise source in series with either input terminal sees referenced to the output. It is also equal to the non-inverting mode gain.
The second op amp topology is called current feedback or CFB. The topology is referred to as CFB because the error signal is a current and not voltage. In this diagram, the non-inverting input voltage is buffered with a unity-gain buffer and applied to the inverting input. Ideally, the output impedance of this buffer, R O , is zero. Feedback produces a small error current at the inverting input, i. This error current is mirrored across the open-loop trans-impedance gain term, T(s), and the resulting voltage is buffered to the output. In a CFB op amp, the T(s) term is analogous to the A(s) term in a VFB op amp. Note that the CFB op amp has a high non-inverting input impedance, and a low inverting input impedance. Simple algebra as shown, can be used to derive the equations for the CFB topology. The denominator of the equation determines the op amp’s closed-loop frequency response. Note that the equation for V OUT /V IN for small R O shows that the closed loop bandwidth is proportional to R2 and independent of R1. Therefore, if R2 is held constant, the closed-loop gain 1+R2/R1 can be modified by changing R1, and the closed-loop bandwidth will not be affected.
The closed-loop frequency response for a CFB op amp is quite different than for a VFB op amp. Note that, the gain-bandwidth is not a constant. The feedback resistor, R2, is usually fixed for optimum performance. The closed-loop gain is determined by the selected value of R1, with little effect on the closed-loop bandwidth. This makes current feedback op amps useful for wide bandwidth programmable gain amplifiers or PGAs, where the gain can be changed without affecting the bandwidth. However, higher R2 values (above the optimum value) proportionally reduce the closed-loop bandwidth. Lower R2 values (below the optimum value) may cause instability because of the proportional increase in closed-loop bandwidth.
This is a simplified schematic of a current feedback op amp; implemented on a complementary bipolar process with matching PNP and NPN transistors. Q1 and Q2 act as the buffer between the non-inverting and inverting inputs. The input error current is mirrored via Q3 and Q4 to the high impedance node represented by R T and C P . The voltage developed at the high impedance node is then buffered to the output. For simplification, the bias circuits are not shown, but they are extremely important to the proper operation of a CFB op amp.
Shown on this slide, is the model for a CFB op amp, along with the corresponding Bodie plot. The Bodie plot is plotted on a log—to-log scale. The open-loop gain is expressed as trans-impedance T(s) in ohms. The finite output impedance of the input buffer is modeled by R O . The input error current is i . By applying the principles of negative feedback, we can derive the expression for the op amp transfer function as shown. Examination of this equation, quickly reveals that, the closed-loop bandwidth of a CFB op amp is determined by the internal dominant-pole capacitor, C P , the external feedback-resistor R2, and is independent of the gain-setting resistor-R1. This ability to maintain constant bandwidth, independent of the gain makes CFB op amps ideally suited for wide-band programmable gain amplifiers. A CFB op amp is usually optimized for a specific R2 , b ecause, the closed-loop bandwidth is inversely proportional to the external feedback resistor, R2. Increasing R2 from its optimum value lowers the bandwidth, and decreasing it may lead to oscillation and instability because of high frequency parasitic poles.
Here is a simplified schematic for a typical high speed VFB op amp. This circuit, takes advantage of the fast PNPs available on a CB process. The differential signal currents in the collector sides of Q1 and Q2 are fed to the emitters of a PNP cas-code transistor pair; hence the term folded cascode . The collectors of Q3 and Q4 are loaded with the current mirror, D1 and Q5 whiles Q4 provides voltage gain. Note that the slew-rate and full-power bandwidth is determined by how fast the bias current I T can charge and discharge the stray capacitance, C STRAY . Higher slew rate requires a larger input tail current, and hence more power. The small-signal unity-gain bandwidth product is proportional to trans-conductance or gm of the input stage which is also proportional to the input stage tail current. A typical example will show that, the small-signal unity-gain bandwidth product can be 20 to 50 times the full-power bandwidth. Therefore, most VFB designs of this type, actually require emitter de-generation resistors to reduce the input stage trans-conductance and hence reduce the small-signal bandwidth, adding stability and making the unity gain-bandwidth product more in line with the full-power bandwidth. The emitter de-generation resistors also serve to linearize the input stage gm transfer function and thus lower distortion.
Shown here is a simple model for the single-stage amplifier and the corresponding Bodie plot. This is the model for a simple VFB op amp in terms of frequency response. The Bodie plot is constructed on a log-to-log scale for convenience. Note that the high frequency response is determined solely by g m and C P . The equations demonstrates the fundamental property of VFB op amps: The closed-loop bandwidth multiplied by the closed-loop gain is a constant , i.e., the VFB op amp exhibits a constant gain-bandwidth product over most of the usable frequency range. As noted previously, some VFB op amps (called de-compensated ) are not stable at unity gain, but designed to be operated at some minimum amount of closed-loop gain. However, even for these op amps, the gain-bandwidth product is still relatively constant over the region of stability.
In the past, amplifiers that had both low noise and high slew rates were considered mutually exclusive and we had to settle for: a Differential pair that had low noise, but also low slew rate Or if we used an H bridge, we could obtain high slew rates, but not very low noise, It wasn’t until the introduction of the AD8099 that featured a new input structure called a “common mode linearized input stage” that we could obtain both low noise and high slew rate simultaneously
The key point here is that the 2 transistor differential pair architecture offers low noise which is good but also features low slew rates. There are four main noise contributors in the differential pair noise calculation Q1&Q2, Ru, Rb, and Rd. The root of the sum-of-their squares (RSS) equals the total noise of the amplifier.
Until recently, op amp designers had to make the above tradeoffs between the input g m stage quiescent current, the slew-rate and distortion performance. ADI has patented a circuit core which supplies current-on-demand , to charge and discharge the dominant-pole capacitor, C P , while allowing the quiescent current to be small. The added current is proportional to the fast slewing input signal and adds to the quiescent current. The quad-core (g m stage) consists of transistors Q1, Q2, Q3, and Q4 with their emitters connected together as shown. Consider a positive step voltage on the inverting input. This voltage produces a proportional current in Q1 that is mirrored into C P1 by Q5. The current through Q1 also flows through Q4 and C P2 . At the dynamic range limit, Q2 and Q3 are correspondingly turned off. Notice that the charging and discharging current for C P1 and C P2 is not limited by the quad core bias current. However, in practice, small current-limiting resistors are required forming an &quot;H&quot; resistor network as shown. Q7 and Q8 form the second gain stage (driven differentially from the collectors of Q5 and Q6), and the output is buffered by a unity-gain complementary emitter follower (X1).
Compared with the folded cas-code differential pair, this circuit is more complicated. However it provides higher slew rate, but with more noise. It still has four noise components, Qi & Qo, 4 Rhs, Rms and one Rc. There are a lot more of them in the H bridge. More components means more noise, but this amplifier does provide much more slew rate as previously mentioned.
Here is the Common Mode linearized input architecture. There are only 2 noise components. The current source I t is split between the 2 pairs of NPN transistors, and therefore, only contribute half the noise. The higher slew rate circuitry is common mode, so there isn’t any noise contribution from those stages. This architecture enables amplifiers, to offer high slew rate, low noise, excellent distortion and wide bandwidth all in one amplifier. In 2003 ADI released the first high speed amplifier with a Common Mode linearized input. The part number is AD8099. Other devices with common mode linearized inputs are the ADA4899 and ADA4898.
Noise gain, is the amount by which a small amplitude noise voltage source in series with the input terminal of an op amp is amplified when measured at the output. The input voltage noise of an amplifier is modeled in this way. Noise gain must be distinguished from signal gain. This slide shows both inverting and noninverting op amp configurations. Notice that the noninverting noise gain is equal to the signal gain; in this case 1+R2/R1. However, in the inverting configuration the noise gain does not change, but the signal gain changes to –R2/R1. The closed loop bandwidth is the Unity gain bandwidth divided by the noise gain.
The follower circuit –A- and inverter circuit B-, show a general method for increasing the noise gain without increasing the signal gain of the circuit. The signal gain of the follower is +1 and that of the inverter is –1. In the example, the noise gain is increased to approximately 11. A drawback to this trick is that both the DC offset and input noise of the amplifier are raised by the value of the noise gain, when R D is DC-connected. However, when C D is used in series with R D , the offset voltage of the amplifier is not raised, and the gained-up AC noise components are confined to a frequency region above 1/(2 •R D •C D ). A further caution is that, this technique can be somewhat tricky when separating these operating DC and AC regions, and should be applied carefully with regard to settling time. It should be noted that, some amplifiers such as the AD817 and AD847-series are designed to be stable under any capacitive load. The penalty of this adaptive compensation is increased ringing on low-level outputs, and increased distortion because of the resulting increase in non-linearity.
It is quite common to use a capacitor in the feedback loop of a voltage feedback amplifier to shape the frequency response as in a simple single-pole low-pass filter. The resulting noise gain is plotted on the left side of this page and on a Bodie plot to analyze stability and phase margin. Stability is determined by the intersect of the net slope of the noise gain and the open loop gain. For unconditional stability, the noise gain plot must intersect the open loop response with a net negative slope of less than 12dB/octave. In this case the net slope where they intersect is 6dB/octave indicating a stable condition. Notice that figure A shows the second pole in the frequency response occuring at a considerably higher frequency than F-sub-U. In the case of the CFB amplifier, the same analysis is used except that the open loop trans-impedance gain T(s) is used to construct the Bodie plot. In the CFB current noise gain plot, the first pole is determined by R2 and C2. As the frequency continues to increase, C2 becomes a short circuit, and all the inverting input-current flows through R0.
The definition of noise gain -for the purposes of stability analysis- for a CFB op amp must be redefined in terms of a current noise source attached to an inverting input. This current is reflected to the output by an impedance that is defined to be the &quot;current noise gain&quot; of a CFB op amp. The equation of the CURRENT NOISE GAIN is shown. The CFB op amp is normally optimized, for best performance, by a fixed feedback resistor R2. Additional poles in the trans-impedance gain -T(s)- occur at frequencies above the closed loop bandwidth fcl (set by R2). Note that the intersection of the CFB current noise gain with the open loop T(s) occurs where the slope of the T(s) function is 12db/octave. This indicates instability and possible oscillation. It is for this reason that CFB op amps are not suitable in configurations that require capacitance in the feedback loop, such as integrators or low pass filters. They can be used in some filter configurations such as a Sallen-Key filter.
This diagram shows the noise sources found in a typical op amp circuit. Each component contributes to the overall noise of the amplifier. Each element’s noise contribution is calculated and then root sum squared. It is easy to see the contribution that resistors make and hence why low values should be used. The noise current multiplied by the resistors create a noise voltage. The total noise calculation is referred to the output or RTO. If you wanted to figure out the input referred noise, you would need to divide by the gain of the amplifier.
This table indicates how the individual noise contributors are referred to the output. After calculating the individual noise spectral densities in this table, they can be squared, added, and then the square root of the sum of the squares, yields the RSS value of the output noise spectral density since all the sources are uncorrelated. This value is multiplied by the square root of the noise bandwidth i.e noise bandwidth = closed-loop bandwidth multiplied by a correction factor of 1.57 to obtain the final value for the output rms noise.
This figure shows an example of a calculation of total output noise for the AD8011 CFB op amp. All six possible sources are included in the calculation. Noise for RF gain blocks used in communication systems is normally specified as the Noise Figure, NF. However, many wideband op amps find their way into communications systems, so it is useful to understand how NF is calculated for an op amp. NF of an op amp is highly circuit dependent. It is determined by the input voltage & current noise, the closed-loop gain, the actual values of the feed-forward and feed-back resistors, and the source resistance. Noise figure is simply defined as the dB ratio of the total output noise to the output noise due to just the source resistor. Noise figure is useful in communication receiver design, since it can be used to measure the decrease in signal-to-noise ratio.
The first step is therefore to calculate the total output noise. Once the total output noise is calculated, the noise figure can be calculated. In this circuit, the NF depends on whether the input driving source is un-terminated or terminated. For the un-terminated case, the output noise due to the source resistance is simply the resistor noise multiplied by the gain of 2. For the terminated case, additional resistor noise is created by the termination resistor, and the resulting NF is 6dB worse. Note that the input noise-current flows through 50 in the un-terminated case and 25 in the terminated case. The effects on the output noise due to this difference are negligible.
There are a few consideration for selecting high speed OP AMP as it relates to noise. For VFB, read specs are ok. However, for current feedback op-amps, note that even though the voltage noise is low the current noise is NOT. Therefore one needs to be careful when selecting amplifiers. Don’t just look at the voltage noise. You need to check out the current noise as well. Remember that the current noise gets multiplied by the feedback resistor, the noise gain of the amplifier and finally it also gets multiplied Rp.
High speed op amps are optimized for bandwidth and settling time, and not for precision DC characteristics as found in lower frequency op amps. High speed op amps do have reasonably good DC performance. The model shows how to reflect the input offset voltage and the offset currents to the output. Again remember that R2/R1+1 is the noise gain of the amplifier. If large values of resistors are used, then it will incur large offset voltages at the output. The output offset voltage due to the input bias currents can be nulled by making the effective source resistance, R3, equal to the parallel combination of R1 and R2.
Here is the summary of the input offset voltage and the offset currents in high speed op amps.
The addition of external load capacitance to an op amp output often creates instability because of the extra parasitic pole which is introduced. External capacitance can be stray capacitance of a long run on a PC board, or the input of the next stage in the system. This is shown in the left-hand diagram, where C L causes an unwanted breakpoint in the open-loop response. The noise gain of the circuit is 1, and it intersects the open-loop gain at a slope which is equal to – 12dB/octave, an unstable situation. Oscillation or excessive ringing may result. One method to restore stability is to increase the noise gain as shown in the right-hand diagram. Here, the intersection of the noise gain with the open-loop gain occurs where the slope is – 6dB/octave, a region of stability. Note that an obvious penalty of the increase in noise gain is a proportional reduction in closed-loop bandwidth as well as increased sensitivity to input offset voltage and noise voltage.
Another solution to load capacitance problems is to use a snubbing resistor Rx or Rs. Shown here is the AD811 CFB op amp, but this technique can be used on any amplifier. The snubbing resistor isolates the load capacitance from the output resistance and pushes the parasitic pole out to where it doesn’t cause a problem. The Drawback to this approach is loss of bandwidth as Rx works against CL and the loss of output voltage if Rx gets too large. Typically Rx is empirically determined, but consult the datasheet to see if there is a recommended value.
Fast op amps are useful as current-to-voltage converters in such applications as high speed photodiode preamplifiers and current-output DAC buffers. A typical application using a VFB op amp as an I/V converter is shown. The total input capacitance C1 is the sum of the photodiode amplifier capacitance and any parasitic capacitance from the PCB. The shunt resistance of the photodiode is neglected since it is much larger than the feedback resistor. The net input capacitance forms a pole at fp. Note that we are neglecting the effect of C2 as we are assuming it is small compared to C1. If left uncompensated the phase shift at the frequency of intersection fx will cause instability and oscillate. Introducing a zero at fx by adding the feedback cap C2 stabilizes the circuit and yields a phase margin of about 45 degrees. In practice C2 is typically determined empirically.
High speed photodiode pre-amps such as shown are real challenges, especially if the circuit is to be optimized to operate with as much bandwidth as possible. A FET input op amp with low noise and offset is mandatory. The value of R2 is generally selected to produce the required output voltage for a given amount of photodiode current. Typically, this resistance is at least 100k and may be as high as 1G in low frequency precision applications. The total input capacitance C1 is the sum of the diode capacitance and the input capacitance of the op amp. This causes a pole in the frequency response as shown. The diode is operated under reverse bias to minimize its capacitance which can be as low as a few pF. Left uncompensated, instability will occur. Therefore, C2 must be added to cause another breakpoint in the noise gain plot at f 2 . The equations will yield an approximate value for C2 which will produce the maximum bandwidth possible for the given set of conditions while maintaining a phase margin of 45 ° . In practice, C2 is generally optimized in the actual circuit for best performance. In the equations, f 2 represents the signal bandwidth 1/(2 C2R2). Note that the final equation gives f 2 in terms of the unity gain-bandwidth product of the op amp fu, the feedback resistor R2, and the total input capacitance C1. The ratio of f u to the input capacitance of the op amp can be used as an effective selection criteria for this type of op amp. It should be noted that this same second-order system analysis is suitable to other applications, such as an I-by-V converter for a high output impedance DAC.
Thank you for taking the time to view this presentation on “ High Speed Amplifiers Part 1 ” . If you would like to learn more, or go on to purchase some of these devices, you may either click on the part list link, or simply call our sales hotline. For more technical information, you may either visit the Analog Devices site, or if you would prefer to speak to someone live, please call our hotline number, or even use our ‘live chat’ online facility.
High Speed Amplifiers Part 1
High Speed Amplifiers Part 1 <ul><li>Source: Analog Devices </li></ul>
Introduction <ul><li>Purpose </li></ul><ul><ul><li>This training module introduces the basic knowledge of high speed operational amplifiers and some characteristics. </li></ul></ul><ul><li>Outline </li></ul><ul><ul><li>High Speed Amplifiers Defined </li></ul></ul><ul><ul><li>Architectures </li></ul></ul><ul><ul><li>Noise Gain vs. Signal Gain </li></ul></ul><ul><ul><li>Noise Analysis </li></ul></ul><ul><ul><li>Voltage Offset </li></ul></ul><ul><ul><li>Driving Capacitive Loads </li></ul></ul><ul><ul><li>Compensating for Input Capacitance </li></ul></ul><ul><li>Content </li></ul><ul><ul><li>33 pages </li></ul></ul>
How to Classify High Speed Amplifiers? <ul><li>High Speed Amplifiers </li></ul><ul><ul><li>Have Bandwidths > 50MHz </li></ul></ul><ul><ul><li>AC specification driven </li></ul></ul><ul><ul><ul><li>Wide bandwidths </li></ul></ul></ul><ul><ul><ul><li>High slew rates </li></ul></ul></ul><ul><ul><ul><li>Fast settling time </li></ul></ul></ul><ul><li>Precision Amplifiers </li></ul><ul><ul><li>BW < 50MHz </li></ul></ul><ul><ul><li>Offset voltage < 1mV </li></ul></ul><ul><ul><li>Low drift </li></ul></ul><ul><li>HS AMPs Applications: Communications, video, medical imaging, commercial imaging, consumer electronics, industrial and data acquisition </li></ul>
Typical High Speed Amplifier Specifications <ul><li>Bandwidth: 50MHz to 2.1GHz </li></ul><ul><li>Slew Rate: 30V/ µ sec to 5500V/ µ sec </li></ul><ul><li>Settling Time: 1 µ sec to 4nsec </li></ul>
Voltage Feedback (VFB) OP-AMP Model ~ + – v – A(s) v A(s) = OPEN LOOP GAIN R2 R1 V OUT V IN V OUT V IN 1 + R2 R1 1 + 1 A(s) 1 + R2 R1 = 1 + R2 R1 1 + 1 A(s) β =
Gain-Bandwidth Product for Voltage Feedback OP AMPS GAIN dB OPEN LOOP GAIN, A(s) IF GAIN BANDWIDTH PRODUCT = X THEN Y · f CL = X f CL = X Y WHERE f CL = CLOSED-LOOP BANDWIDTH LOG f f CL NOISE GAIN = Y Y = 1 + R2 R1
Current Feedback (CFB) OP AMP Model + – i – T(s) i T(s) = TRANSIMPEDANCE OPEN LOOP GAIN T(s) R O i ×1 R2 R1 V IN V OUT V OUT V IN 1 + R2 R1 1 + R2 T(s) 1 + R O = 1 + R2 R1 1 + R2 T(s) R1 + R O R2 ASSUME R O << R1, AND R1 R2, THEN V OUT V IN ×1
Frequency Response for Current Feedback OP AMPS <ul><li>Feedback resistor fixed for optimum performance. Larger values reduce bandwidth, smaller values may cause instability. </li></ul><ul><li>For fixed feedback resistor, changing gain has little effect on bandwidth. </li></ul><ul><li>Current feedback op amps do not have a fixed gain-bandwidth product. </li></ul>GAIN dB G1 G2 G1 · f 1 G2 · f 2 f 1 f 2 <ul><ul><li>LOG f </li></ul></ul>
AD8011 Output Noise Analysis / H z 4 n V / H z 5 p A / H z 4 n V / H z AD 80 11 5 p A / H z 0 . 9 n V / H z 2 n V / H z R1 1 k R2 1 k (G • R S ) (G ) ( 1 ) (R 2) (- R 2/ R 1) G = 1 + R2 R1 (G ) + - R S 5 0 f CL = 180MHz 1 . 8 n V / H z 0 . 5 n V / H z 4 n V / H z 4 n V / H z 5 n V / H z 4 n V OUT PUT NO ISE SP ECTRAL DENS ITY = 8.7nV/ Hz TOTAL NOISE = 8.7 1.57 X 180 X 10 6 = 146 V rms
AD8011 Noise Figure for Unterminated and Terminated Input Conditions Unterminated 50 1k 1k + – R S V no(total) = 8.7nV / Hz, from previous slide V no(Rs) = G = 2 4kTR G 4kTR 0.9nV/ Hz = 1.8nV/ Hz NF = 20 log 8.7 1.8 = 13.7 dB 50 1k 1k + – R S V no(total) 8.7nV / Hz (See Note) V no(Rs) = G = 2 4kTR G kTR 0.9nV/ Hz = 0.9nV/ Hz NF = 20 log 8.7 0.9 = 19.7 dB 50 Note: Input noise current (I n+ ) flows through 50 (unterminated case) or 25 (terminated case), but the overall effect of this is negligible. I n+ I n+ Terminated AD8011 AD8011
High Speed OP AMP Noise Summary <ul><li>Voltage Feedback Op Amps: </li></ul><ul><ul><li>Voltage Noise: 2 to 20nV/ Hz </li></ul></ul><ul><ul><li>Current Noise: 0.5 to 5pA/ Hz </li></ul></ul><ul><li>Current Feedback Op Amps: </li></ul><ul><ul><li>Voltage Noise: 1 to 5nV/ Hz </li></ul></ul><ul><ul><li>Current Noise: 5 to 40pA/ Hz </li></ul></ul><ul><li>Noise Contribution from Source Negligible if < 100 </li></ul><ul><li>Voltage Noise Usually Dominates at High Gains </li></ul><ul><li>Reflect Noise Sources to Output and Combine Root Sum Squared (RSS) </li></ul><ul><li>Errors Will Result if there is Significant High Frequency Peaking </li></ul>
Model for Calculating Total Op Amp Output Voltage Offset V O = ±V OS 1 + + I b+ R3 1 + - I b- R2 IF I b+ = I b- AND R3 = R1||R2 V O = ±V OS 1 + I b - V OS + - R3 R2 R1 R2 R1 I b+ R2 R1 R2 R1 V O
Output Offset Voltage Summary <ul><li>High Speed Bipolar Op Amp Input Offset Voltage: </li></ul><ul><ul><li>Ranges from 0.1mV to 3mV for VFB and CFB </li></ul></ul><ul><ul><li>Offset TC Ranges from 5 to 15µV/°C </li></ul></ul><ul><li>High Speed Bipolar Op Amp Input Bias Current: </li></ul><ul><ul><li>For VFB Ranges from 0.1 to 5µA </li></ul></ul><ul><ul><li>For CFB Ranges from 5 to 15µA </li></ul></ul><ul><li>Bias Current Cancellation Doesn't Work for: </li></ul><ul><ul><li>Bias Current Compensated Op Amps </li></ul></ul><ul><ul><li>Current Feedback Op Amps </li></ul></ul>
External Load Capacitance to An OP AMP Output Capacitive loading on op amp generally reduces phase margin and may cause instability, but increasing the noise gain of the circuit improves stability.
Open-loop Series Resistance Isolates Capacitive Load for AD811 Current Feedback Op Amp 6 7 4 3 2 - + A D 8 1 1 R I N 30. 9k R F 750 R X 1 2 R L 500 V OUT C L 1n F 0.1 F +1 2V -1 2V 100 F/ 2 5V 100 F/ 2 5V 0.1 F R B 1 k V IN
Compensating For Input Capacitance in a Current-to-voltage Converter Using VFB OP AMP
Generalized Model for High Speed Photodiode Preamp C2 R2 C1 I + _ 1 f 2 f 1 f u f NOISE GAIN OPEN LOOP GAIN UNCOMPENSATED COMPENSATED f 1 = f 2 = f 2 = f 1 • f u C2 = 1 2 R2 C1 1 2 R2 C2 C1 2 R2 f u FOR 45° PHASE MARGIN GAIN f u = OP AMP UNITY GAIN BW PRODUCT f 2 = SIGNAL BW f u Total Input Capacitance f 2 = f u 2 R2 C1 – V B
Additional Resource <ul><li>For ordering high speed operational amplifiers, please click the part list or </li></ul><ul><li>Call our sales hotline </li></ul><ul><li>For additional inquires contact our technical service hotline </li></ul><ul><li>For more product information go to </li></ul><ul><ul><li>http://www.analog.com/en/amplifiers-and-comparators/operational-amplifiers-op-amps/products/index.html </li></ul></ul>Newark Farnell