Successfully reported this slideshow.
We use your LinkedIn profile and activity data to personalize ads and to show you more relevant ads. You can change your ad preferences anytime.

EVALUATION OF RADIATED EMISSIONS FROM PCB AND CABLES AT POST-LAYOUT LEVEL

104 views

Published on

EMC ROME 1996: EVALUATION OF RADIATED EMISSIONS FROM PCB AND CABLES AT POST-LAYOUT LEVEL

Published in: Technology
  • Be the first to comment

  • Be the first to like this

EVALUATION OF RADIATED EMISSIONS FROM PCB AND CABLES AT POST-LAYOUT LEVEL

  1. 1. EVALUATION OF RADIATED EMISSIONS FROM PRINTED CIRCUIT BOARDS AND CABLES AT POST-LAYOUT LEVEL E. Leroux High Design Technology Corso Trapani, 16 10139 Torino Italy S. Caniggia ITALTEL Castelletto di Settimo Milanese (MI) 20019 Italy F. Canavero Politecnico di Torino Corso Duca degli Abruzzi, 24-10100 Torino Italy B. Demoulin University of Lille 59655 Villeneuve d’ascq France Abstract - Because it costs to solve ElectroMagnetic Compatibility (EMC) problems late in the development process, new methods have to predict radiated electromagnetic emissions at the design stage. In the case of complex Printed Circuit Boards (PCBs) with attached cables, a tradeoff between accuracy and simulation time must be found for this evaluation. In this paper the Radiated Emissions (RE) of single boards with and without an attached cable are investigated. A feasibility study is proposed for the modelling of these emissions. The results are compared with some measurements for validation. Some ideas are given to improve the models and to introduce them into an EMC post- layout environment. 1. INTRODUCTION Since January 1996, electronic products have also to pass the requirements of the european norms in ElectroMagnetic Compaltibility (EMC) to obtain the EEC mark and be sold in the european market. In many cases electronic equipment is mostly composed of Printed Circuit Boards (PCBs), so PCB's manufacturers will have properly to limit the Radiated Emissions (RE) of the boards they produce. Currently, the most common method of handling ElectroMagnetic (EM) emissions is through compliance testing of the first prototype, already implemented. In the case of a PCB it would be necessary to have the first board made by the manufacturer, and if it does not pass the tests, repeat again all the manufacturing processes. And this may delay the product's completion date and increase the unit cost of the product because the designer has not as many options available for correcting an EMC problem late in the development process. A solution to limit this possible iteration is to predict the EM Field at the design stage, thanks to a computer-oriented analysis of EM radiated Interferences (EMI). This proposal seems to be quite hard to solve because of the number of factors that influence the radiated EM field. As the microstrip structures that compose the nets play a major role especially in the frequency bands of the norms, a method [1] has been developped to predict their differential mode emissions. This algorithm uses the dyadic Green's functions of the actual medium (the PCB) and is able to take into account of the dielectric layers in the field calculation. In particular dielectric covers modify the radiation spectrum of PCBs and have to be considered in a predictive method. But the presence of unintended common mode currents on the external cables of electronic equipment is often the primary source of radiated ElectroMagnetic Interferences (EMI) at frequencies above 30 MHz. In this paper a feasibility study on the modelling of RE of simple two-layered PCBs with and without an attached cable is presented. Simulations and measurements are conducted on two simple test boards made of a microstrip and a stripline. Some ideas are given to improve the models and to introduce them into an EMC post-layout environment. 2. DESCRIPTION OF THE COUPLING AND THE TEST BOARDS A metal plane is commonly thought of as a low impedance path for returning currents and one which is of constant potential across its area. In fact, these currents give rise to voltage gradients in the plane which act as sources of common-mode current. The majority of the RF current flowing along a signal trace on a multi-layer PCB returns on the ground plane directly beneath the signal trace. However, a small portion of the ground-plane current also can return via indirect paths causing the PCB and attached cables to produce common-mode radiation similar to a dipole antenna. To investigate the effect of these radiation sources in actual PCBs, two structures are considered (see Figure 1): one wire or strip over a ground plane separated by dielectric (microstrip structure), one wire or strip between two ground planes with dielectric (stripline structure). Both the structures were tested with and without an attached cable. Models were built to simulate the radiation of these two PCBs with attached cables powered by the PCB ground noise. In order to verify the models, some experiments were conducted. For every structure a 8-MHz oscillator, with output resistance of 50 W, was mounted on a board enclosed in a shielded box of
  2. 2. size: 5.5´5´2.5 cm (see Figure 2). The output connector was a SMA type. This provided a 8 MHz trapezoidal pulse train having roughly a 50-percent duty cycle, a rise and fall time of approximately 2 ns and amplitude of 5 V without load. A battery- powered supply consisting of a 9 V battery and a 7805 regulator provided the 5 V DC for the oscillator. It is important to note that we never used connection to the commercial power system. M icro strip S trip line h 2 h P lanes: W id th= 1 9 cm L ength= 3 0 cm er= 4 .4 w ire: rad ius= 0 .5 m mh= 1 .6 m m Z 0 = 1 5 3 W Z 0 = 1 2 5 W Figure 1: Examined structures for multilayer circuit boards: microstrip and stripline. Shielded box PCB local plane tfall=2 nsec Period of 125 nsec trise=2 nsec 5 Volts Source: Digital gate AC 244 50W PCB track 100 W 8Mhz Figure 2: The 8 MHz digital circuit with the output waveform used in shielded box The models use the envelope of the generator output spectrum. In Figure 3, a comparison between the measured spectrum of the source on the 50W input of a spectrum analyser and the envelope of the calculated spectrum is given. 1 10 7 1 10 8 1 10 950 60 70 80 90 100 110 120 Figure 3: Generator output spectrum on a 50 W load. For the calculations reported in this paper, we have always used the spectral bounds minus 3 dB to take into account the rms values measured by the spectrum analyser. In this paper the radiated emissions were measured at 3 meters of distance with the antennas at 1.1 m height from the metallic ground floor, in horizontal polarisation. In order to maximise the emission the PCB was orientated with the plane in vertical position and the center of the PCB was 1.1 m above the ground plane of the chamber. The PCB track was terminated with a 100 W resistance. 3. THE MICROSTRIP STRUCTURE 3.1 Differential mode emissions of the PCB alone Firstly the differential mode emissions of the PCB are investigated by means of measurements and simulations. For the microstrip structure, as no dielectric cover is present, the Hertzian Radiating Dipoles Method [2] has been used to model the differential mode emissions: · the current distribution on the track is obtained using the Transmission Line Theory (TLT) from the envelope of the generator calculated spectrum · the track is divided into segments for which the lenght L << l, l is the considered wavelenght · the classical elementary Hertzian dipole formula is used to calculate the emissions of each segment · the PCB local plane and the floor of the semi- anechoic chamber are taken into account using image theory The results are reported in Figure 4. Note that the calculated differential mode emissions fit quite well with the higher measured results. Frequency in MHz |V| in dBmV envelope of the calculated spectrum measured spectrum
  3. 3. 10 100 1 10 320 10 0 10 20 30 40 50 60 70 80 EN55022 norm simulation measurement frequency in MHz |E| in dBmV/m Figure 4: Microstrip configuration: RE of only PCB 3.2 Modelling of common mode currents on the PCB Secondly, a more accurate model to predict radiated emission from a microstrip structure is presented (Figure 5). In this figure 5, one can observe that the total radiated field is the sum of two contributions: a differential mode radiation EDM due to the differential signal current Isign; a common mode radiation ECM due to the ground noise Vn always present in a PCB with a finite dimensions ground plane. IDM» + Etot=EDM+ECM ECM EDM ICM Vn Ltrans= Lplane - Mplane/track Isign»IDM Ground noise Vn Vn=j w Ltrans Isign Etot,EDM,ECM are vectorial quantities Figure 5: Radiated emission from a PCB as sum of common mode (ground noise) and differential mode (signal current). This ground noise depends on the coupling between the track, the PCB local plane and the metal floor of the semi-anechoic chamber. The finite ground plane of a PCB has an associated partial inductance Lgnd [3]. But it is not only the self-inductance of the plane that matters but also its coupling to the (much larger) inductance of the track [4]. Together these parameters determine the voltage drop over the PCB local plane, if a current flows trough the track and returns through the plane. This voltage drop is proportional to the signal differential current Isign and is responsible for common mode emission through the distributed stray capacitance of the ground plane or by a cable attached to the PCB (see [4] [5]). In absence of an I/O cable, the contribution of the common mode radiation ECM could also be neglected in the case of a PCB having a large ground plane and placed in vertical position. This is not the case when a cable is attached to the PCB. 3.3 Emissions of the PCB with an attached cable 3.3.1 The used model Finally the emissions of the PCB with an attached cable are investigated. A cable of 1 m is attached to the structure of Figure 1, in horizontal position and at 1.1m from the ground of the chamber floor, it is not connected at the other extremity (high impedance). The model of Figure 6 has been used. It has already been explained in [4] and especially in [6] in which the feasibility study started. In the present article some improvement respect to [6] are presented. cable Vn=j wLtrans Isign C10 metal floor of semi-anechoic room Ltrans= Lplane -Mplane/track H Figure 6: The used model for the cable exitation The modelling follows three steps: · The determination of the ground noise. C10 is the capacitance between the PCB local plane and the metal floor of the semi-anechoic chamber. Ltrans is a type of a “transfer inductance”, it depends on the mutual inductance between the PCB local plane and the track. Respect to [6] these elements have not been calculated using 2D approximations or analytical formula but using the Partial Element Equivalent Circuit (PEEC) method (see 3.3.2). A purpose of this article is to see if the prediction of the radiated emissions can be by this way improved · The propagation of the ground noise on the cable.
  4. 4. The Transmission line theory (TLT) is used to calculate the current distribution at any abscissa on the cable. The distance H between the PCB local plane and the floor of the semi-anechoic chamber defines the maximum frequency for which TLT can approximatively apply: fmax . = 3108 l with l = H =1.3 meters, fmax = 230 Mhz This frequency correspond to the maximum frequency for which the emissions of the cable is much bigger than the ones of the PCB (see 3.3.3). In fact the method is used until this maximum frequency. · The calculation of the cable emissions. The Hertzian Radiating Dipoles Method has been used dividing the cable in short segments to calculate the RE of the cable. The floor of the semi- anechoic chamber is taken into account using image theory 3.3.2 The use of PEEC method to determine the concentrated desired elements The PEEC method allows the evaluation of parasitic parameters in threedimensional structures of conductors and was introduced in 1974 by Ruehli [7] [8]. The key idea is to subdivide all conductors in volume cells where conduction or polarization current flows and in surface cells where free or boundary charge is located. Inductance and coefficient of potential matrices are then calculated, whose elements are called partial, as they are referred to the single cell and not to the whole conductor. Calculated values can be directly used as circuit parameters (inductances and capacitances), or can be grouped in order to obtain macromodels, as it has been done in this application. This method allows to model very general structures, because the only restriction in choosing cells is due to their shape: their size must be parallel to cartesian axes. Using PEEC approach it is possible to study PCBs placed either parallel or perpendicular to the floor of the semianechoic chamber, only changing cells layout. In this paper, the interaction between the microstrip (microsystem) and the floor of the semianechoic chamber (macrosystem) has been evaluated. In Figure 7 the subdivision of the structure in volume conductive cells to evaluate the inductive coupling is represented. Figure 7: Example of discretization in volume cells (3 for the PCB local plane and 1 for the PCB track). The quantity Ltrans=Lplane - Mplane/track, was to be evaluated: it has not been necessary to introduce cells on the floor. Furthermore the current mainly flows parallel to the track, then volume cells were placed only along this direction, and not transversally. Repeated tests leaded to the convergency of Ltrans at a value of 0.9 nH, using 25 volume cells in both conductors. With the analytical formula in [6], Ltrans was equal to 0.89 nH. The evaluation of the capacitive coupling has been done without modeling dielectric: C10 (coupling between the PCB local plane and the floor) was the only quantity of interest, representing the coupling between the micro and the macrosystem. Repeated tests leaded to the convergency of C10 at a value of 5.27 pF, using a grid of 5x5 uniform cells on the track and 7x7 on the local plane and on the floor. A value of 10 pF has been obtained with the 2D approximation in [6]. 3.3.3 Comparison between simulation and measurement The Figure 8 shows the results of simulation of the emissions from the PCB alone, the attached cable, the PCB with the attached cable and the measurement of the emissions from the PCB with the attached cable.
  5. 5. 10 100 1 10 320 10 0 10 20 30 40 50 60 70 80 FrequencyinMHz |E| indBmV/m EN55022 EmissionsfromPCB+ cable: measurement EmissionsfromPCB+ cable: simulation EmissionsfromPCB: simulation Emissions fromthe cable: simulation 8 Figure 8: Microstrip configuration. Simulation of the emissions from the PCB alone, the attached cable, the PCB with the attached cable. Measurement of the emissions from the PCB with the attached cable The comparison of the experimental results of Figure 4 and 8 shows that the cable has a major contribution on the RE from the system PCB + cable mainly below 250 Mhz. In deed, the cable is one meter long and the PCB is only 30 cm long. Until fmax = 230 MHz the maximum frequency for which it is possible to apply the method, the simulated envelope follows reasonably the measured spectrum. But the most significant frequency is shifted and the method underestimates the maximum radiated field. It is probably due to the use of the envelope of the signal injected on the track and the fact that the method does not take into account of the coupling by proximity between the track and the cable. The use of a 3D method to calculate the parasitic elements C10 and Ltrans does not improve drastically the results of the radiated emissions in dB respect to [6] as the inductive effect is predominant and was predicted in a good approximation in [6] (see 3.3.2). The quantity Ltrans does not depend on the presence of the ground floor of the semi-anechoic chamber, it can be so approximatively determined by the use of analytical formula also for a PCB placed in a vertical position. For non terminated lines the capacitive effects have a greater influence. As the PEEC method gives a 3D evaluation of C10, it is worth using it to determine the capacitive coupling when the board is placed in horizontal or vertical position. 4. THE STRIPLINE STRUCTURE 4.1 Emissions of the PCB alone Respect to [6] an other very common PCB structure has been examined in this paper: it is the stripline type that can be found in multilayer PCBs with several ground planes. Due to the nature of the structure, one wire/strip between two ground planes, one could expect that the ground planes act like a shield and no practical emission should be measured. This is true if the planes are tiedly connected together or, in the limit, also attached around their periphery in order to form a perfectly closed container. But, if one plane is floating or badly connected to the other plane, resonances occur and produce emissions higher than those produced by an equivalent microstrip structure [9] In our case of study the two planes are tiedly connected. One must so realise that the structure is quasi-symmetric, the signal current is quasi-equally divided between the two planes and the emission is only due to a small percentage difference between the current of the two ground planes. The current distribution on the ground planes depends on the frequency. But for striplines used in modern high speed digital electronics, the currents on the structure can be calculated by a multiconductor transmission line model taking into account that the return current in the ground plane is not spread more than 4-5 times the distance between the strip and one of the ground plane [10]. The equivalent model consists of three conductors: one for the track and the other two for the planes spaced a distance 2 h as shown in Figure 1. If a signal differential current Isign spreads along the conductor which models the track, a current Isign (1+K)/2 spreads in the conductor which models one metal plane, a current Isign (1-K)/2 spreads in the conductor which models the other plane. For example, the calculated values reported in Figure 9 were obtained by a difference of about 20% between the two return currents on the ground planes (K=0.2) and take into account the phase differences.
  6. 6. 10 100 1 10 320 10 0 10 20 30 40 50 60 70 80 |E| in dBmV/m EN 55022 simulation measurement Frequency in Mhz Figure 9: Stripline configuration: emissions of the PCB alone The simulated envelop follows reasonably the measured spectrum. But it would be necessary to justify better the difference of about 20% between the two return currents on the ground planes. 4.2 Emissions of the PCB with an attached cable 4.2.1 The used model A cable is attached to one of the two planes of the stripline. The same model as in 3.3.1 is used but the equivalent generator Vn is equal to: j w Ltrans Isign /2. The quantities Ltrans and C10 have to be calculated for a stripline structure placed in vertical position above the metal floor of the semi-anechoic chamber. 4.2.2 The use of PEEC method to determine the concentrated desired elements The same approach as in 3.3.2 is used to determine the capacitive coupling between a stripline in vertical position and the metal floor of the chamber. Repeated tests leaded to the convergency of C10 at a value of 10.7 pF, using a grid of 3x3 uniform cells on the track and 9x9 on the local plane and on the floor. As far as inductive coupling is concerned, Ltrans has been evaluated by a closed form formula. 4.2.3 Comparison between simulation and measurement The Figure 10 shows the results of simulation of the emissions from the PCB alone, the attached cable, the PCB with the attached cable and the measurement of the emissions from the PCB with the attached cable. 10 100 1 10 320 10 0 10 20 30 40 50 60 70 80 |E| in dBmV/m Frequency in MHz EN 55022 emissions fromthe PCB + cable: simulation emissions from the PCB + cable: measurement emissions fromthe PCB alone emissions from the cable: simulation Figure 10: Stripline configuration. Simulation of the emissions from the PCB alone, the attached cable, the PCB with the attached cable. Measurement of the emissions from the PCB with the attached cable. With a cable attached to one of the two planes of the stripline structure the same level of emission as those obtained with the microstrip structure is noticed. The same remarks presented for the modelling of the microstrip structure apply for the stripline. CONCLUSION A feasibility study on the modelling of the Radiated Emissions (RE) from simple two-layered PCBs with and without an attached cable has been presented. The validity domain of these models has been evaluated by the means of comparisons with measurements. A numerical approach as the PEEC method is necessary to determine the capacitive coupling between the PCB and the metal floor of the semi- anechoic chamber especially when the board is in vertical position. The inductive coupling can be rapidly determined by the use of close-form formulas but it is more accurately determined using PEEC approach. A major improvement on the presented modelling could be the use of Simulation Program of Response of Integrated Network Transients (SPRINT) [11] simulator to obtain the current on the track instead of using the envelope of the calculated spectrum at the near end of the PCB track. As SPRINT is embedded in an EMC post-layout environment: Post-layout Rapid Exhaustive Simulation and Test of Operation (PRESTO) [11], these models could be inserted in PRESTO in order to complete the PRESTO/EMIR solution for RE prediction.
  7. 7. In the presented examples the cable was responsible for non passing the EN 55022 limits. The cable is a radiating structure which is powered by the PCB ground noise. Reducing the ground noise at the PCB level will reduce the emissions level of the cable. It is so useful to use an EMC environnment for PCB to try to control the PCB ground noise already at Post- layout level. A stripline, of which the two planes are tiedly connected, is a quite symmetrical structure and produces low emissions. But with a cable attached to one of the two ground planes the stripline radiates as a microstrip with an attached cable. ACKNOWLEDGEMENTS The authors thank C. Giachino, A. Giuliano for their help to use the PEEC approach and L. Vitucci, G. Marano for their participation to the measurements in ITALTEL. REFERENCES [1] E. Leroux, F. Canavero, G. Vecchi, "Prediction of radiated electromagnetic emissions from PCB traces based on Green dyadics," Proc. EURO-DAC, Brighton (UK), Sept. 18-22, 1995, pp. 354-359. [2] S. Caniggia, L. Catello, V. Costa, “On the computation of electromagnetic filed components fron a transmission line: theory and application”, IEEE 1990 International Symposium on Electromagnetic Compatibility, Washington USA, 21 - 23 August 1990 [3] T. H. Hubing, T .P. Van Doren, J. L. Drewniak, “Identifying and Quantifying Printed Circuit Board Inductance”, IEEE International Symposium 1994 [4] J. R. Bergervoet, “EMC Measurements and Models Connecting the System Level with the Module Level”, Philips Journal of Research, Vol. 48, pp.63-81, 1994 [5] R. W. Dockey, R. F. German, “New Techniques For Reducing Printed Circuit Board Common-Mode Radiation”, IEEE International Symposium 1993 [6] E. Leroux, F. Canavero, S. Caniggia, B. Demoulin, “Contribution des méthodes analytiques pour la prédiction du rayonnement de cartes et connectique”, International Symposium and exhibition on EMC, 2 - 5 september 1996, Lille France [7] A. E. Ruehli, “Equivalent circuit models for three-dimensional multiconductor systems”, IEEE Trans. on Microwave Theory and Techniques, v. MTT-22, pp. 216-221, march 1974 [8] A. E. Ruehli, P. A. Brennan, “Efficient capacitance calculations for three-dimensional multiconductor systems”, IEEE Trans. on Microwave Theory and Techniques, v. MTT-21, pp. 76-82, february 1973 [9] S. Caniggia, L. Catello, V. Costa, G. Marano, N. O’Riordan, L. Vitucci, “Controlling radiated emission from a physically large telecommunication system”, paper to be published on IEEE Trans. on EMC, August 1996 [10] S. Celozzi, A. Orlandi, “PCB emission analysis in different image planes configurations”, International Wroclaw 1994 symposium on EMC, Wroclaw Poland [11] S. Forno, S. Rochel, "Advanced simulation and modeling techniques for hardware quality verification of digital systems," Proc. EURO-DAC, Grenoble (F), 1994.

×