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Carrier Aggregation for LTE-Advanced: Design Challenges of Terminals

Carrier Aggregation for LTE-Advanced: Design Challenges of Terminals

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Carrier aggregation  lte challenges(devices) Carrier aggregation lte challenges(devices) Document Transcript

  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 76 RADIO COMMUNICATIONS Carrier Aggregation for LTE-Advanced: Design Challenges of Terminals Chester Sungchung Park, Konkuk University Lars Sundström, Anders Wallén, and Ali Khayrallah, Ericsson ABSTRACT Carrier aggregation is a key feature of 3GPP LTE that addresses the support of higher data rates and utilization of fragmented spectrum holdings. In this article, the relevant design challenges of terminals are discussed. The transmitter architectures are reviewed, and the minimum amount of power amplifier back-offs is evaluated. In addition, several receiver architectures are compared from the perspective of design tradeoff. The radio impairments affecting the receiver performance are analyzed and the simulation results are provided. The corresponding silicon implementation is presented together with the measurement results. INTRODUCTION This article addresses the radio frequency (RF) design challenges of terminals for carrier aggregation (CA). CA is one of the key features of the Third Generation Partnership Project (3GPP) Long Term Evolution (LTE), helping to achieve higher data rates by transmitting multiple carriers from or to the same terminal [1]. In Release-10 of LTE, up to five carriers can be aggregated, each with a bandwidth of 1.4, 3, 5, 10, 15, or 20 MHz, thereby allowing for overall bandwidth of up to 100 MHz. Wide bandwidths are very desirable, as they enable mobile broadband services such as high-definition video streaming to take off. Unfortunately, in many areas of the world, there are currently few, if any, contiguous spectrum allocations of 20 MHz or higher. CA is the technical solution to overcome the spectrum fragmentation, and it provides the flexibility needed to adapt to a wide variety of spectrum scenarios. However, with flexibility come RF design challenges of terminals, as will become clear in this article. Carriers may be contiguous or non-contiguous (NC) in frequency within the same band (intra-band CA) or across multiple bands (interband CA), which allows for full utilization of fragmented spectrum. From a digital baseband point of view, there is little difference among these CA types. From an RF implementation point of view, however, the design complexity of the terminal heavily depends on the CA type, 76 0163-6804/13/$25.00 © 2013 IEEE with intra-band contiguous CA being the least complex. Moreover, the RF characteristics of a terminal vary with the aggregated frequency bands. During initial access, a CA-capable terminal behaves similarly to a terminal from earlier releases; that is, there is a single carrier, referred to as a primary component carrier (PCC). Upon successful connection to the network, depending on its own capabilities and the network configuration, a terminal may be configured with additional carriers in the uplink (UL) and downlink (DL), which are referred to as secondary component carriers (SCCs). Therefore, in reality, a terminal may be configured with multiple CCs even though not all of them are currently used. In order to save battery consumption, the network can activate/deactivate the configured carriers for a particular terminal as needed. In Release-10, the number of DL carriers is always greater than or equal to the number of UL carriers (i.e., only DL-heavy asymmetries are supported). This is useful to handle different spectrum allocations, for example, if an operator has more spectrum available for DL than for UL. In addition, this helps to provide a terminal with DL-heavy traffic such as high-definition video streaming. For example, in Fig. 1a, terminal X1 supports two aggregated carriers in DL, whereas it supports a single carrier (i.e., no CA) in UL. To facilitate load balancing across carriers and handle different capability among terminals, the configuration of carriers is terminal-specific. In other words, the configuration of PCC and the set of carriers in each direction may vary from terminal to terminal. For example, in Fig. 1a, the PCC and the set of carriers supported by terminal X2 are different from those supported by terminal X3. Therefore, it readily follows that the number of carriers configured in a network may be different from the number of carriers supported by a terminal. Since each carrier uses the legacy physical layer structure in order to achieve backward compatibility, it is fully accessible to a terminal from earlier releases (e.g., terminal X3). The association between UL and DL carriers is cell-specific and signaled as part of the system information. CA is supported for both frequency-division IEEE Communications Magazine • December 2013
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 77 duplex (FDD) and time-division duplex (TDD) with all carriers using the same duplex scheme. Throughout this article, FDD operation is assumed, since it is generally more challenging from the perspective of RF implementations (e.g., transmitter emission). The readers are referred to [1, 2] for more technical details of CA, and [3] for a general understanding of radio design. In this article, the design challenges of terminals are described, assuming intra-band NC CA (unless otherwise mentioned). The coexistence of different networks is explained, taking into account the unwanted emission requirements of a transmitter and the selectivity and blocking requirements of a receiver. Several transmitter architectures for CA operation are introduced. Considering the nonlinearity of the transmitter (both RF chain and power amplifier [PA]), the minimum PA back-off required to meet the unwanted emission requirements is evaluated. In addition, several receiver architectures for CA operation are introduced, and the radio impairments affecting the receiver performance are discussed together with some simulation results. Specifically, the radio impairments include transmitter emission, phase noise, receiver nonlinearity, local oscillator (LO) coupling, limited harmonic rejection, and limited image rejection. COEXISTENCE ISSUES Component carriers occupied by one operator may be located adjacent to those occupied by another operator. In general, networks belonging to different operators are deployed in an uncoordinated way. In Fig. 1b, base stations X and Y, belonging to operators X and Y, respectively, are placed in different locations. Terminal X2 (connected to the network of operator X) is located closer to base station Y than terminal Y (connected to the network of operator Y). Therefore, an interfering signal from terminal X2 seen by the receiver of base station Y may be much stronger than a wanted signal from terminal Y (since the path loss from terminal X2 to base station Y is much smaller than the path loss from terminal Y to base station Y). Likewise, an interfering signal from base station Y seen by the receiver of terminal X2 tends to be much stronger than a wanted signal from base station X, as illustrated in Fig. 1b. In order to satisfy regulatory requirements and guarantee coexistence with other operators, the transmitter of a terminal should be designed to comply with the unwanted emission requirements, which set limits on all emission outside the channel bandwidth. For example, the unwanted emissions from terminal X2 should be kept reasonably low in order to protect the wanted signal from terminal Y at the receiver of base station Y. The unwanted emissions from the transmitter are divided into out-of-band (OOB) emissions and spurious emissions [4]. The OOB emissions are unwanted emissions immediately outside the assigned channel bandwidth, which are specified in terms of a spectrum emission mask (SEM) and an adjacent channel leakage power ratio (ACLR). The spurious emissions are defined for all frequency ranges other than those IEEE Communications Magazine • December 2013 Downlink X1 Uplink P P X2 P X3 P P P Y P P (a) Base station X Base station Y XYX X Y X Terminal Y X Y X Terminal X2 XYX (b) Figure 1. Examples of CA configuration: a) terminal specific configurations; b) coexistence with uncoordinated operators. covered by the SEM. In the case of intra-band NC CA, the unwanted emission requirements for single-carrier operation are applied to each of the carriers so as to keep the unwanted emission to a maximum of the emission level for single-carrier operation. For example, the SEM and spurious emissions are set to the maximum of those of all carriers [5], as illustrated in Fig. 2. The ACLR is defined on a per-carrier basis and calculated with respect to the total power of aggregated carriers (being used as the numerator) [5]. Similarly, in order to guarantee the reception of a wanted signal in the presence of strong interfering signals (possibly generated by another operator), the terminal receiver should be designed to meet the selectivity and blocking requirements [4]. Note that an interfering signal may be located within the gap between two aggregated carriers. For example, in Fig. 1b, the receiver of terminal X2 should be able to receive two aggregated carriers from base station X in the presence of a strong interfering signal from base station Y inside the gap. TRANSMITTER TRANSMITTER ARCHITECTURES In order to support CA, the baseline transmitter for single-carrier operation can be extended in several ways. The choice of transmitter architecture for CA operation depends on how to combine the aggregated carriers in the transmitter chain. Here we consider three options to illustrate the design trade-offs. Assuming two contiguous or non-contiguous carriers, the left side 77
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 78 -5 PSD [dBm] -10 -15 -20 -25 -30 -35 -40 -30 -20 -10 0 10 Frequency [MHz] 20 30 40 MINIMUM PA BACK-OFF (a) -5 PSD [dBm] -10 -15 -20 -25 -30 -35 -40 -30 -20 -10 0 10 Frequency [MHz] 20 30 40 (b) Figure 2. Unwanted emission requirements: a) single-carrier operation ; b) intra-band NC CA operation . of Fig. 3a illustrates various transmitter architectures depending on where the carriers are combined: at the digital baseband (option 1), at the RF chain before power amplification (option 2), or after power amplification (option 3). Option 1 is characterized by a single transmitter chain. Conversely, option 3 includes two separate transmitter chains, each consisting of a baseband chain, an RF chain, and a PA. The numbers of baseband chains, RF chains, and PAs depend on the transmitter architecture. Furthermore, it is possible to extend option 3 into a multi-antenna architecture by removing the combiner and connecting each PA to an antenna. Option 1 is the least complex architecture, for example, in terms of chip size and power consumption, since it has only a single transmitter chain. This transmitter architecture is generally well suited for contiguous CA. The other extreme, option 3, is the most complex, since each carrier has its own transmitter chain. This transmitter architecture is well suited for interband CA. Option 2, a compromise between these two options, is considered a reasonable candidate for intra-band NC CA. When choosing the transmitter architecture, one of the most important factors to consider is the amount of PA back-off required to satisfy the unwanted emission requirements mentioned earlier. The required PA back-off largely depends on whether the carriers are amplified in a single PA or two different PAs. For example, in options 1 and 2, two carriers are combined first and then amplified with a single PA. The situation is similar to a two-tone test for a PA; that is, a large PA back-off may be required to suppress intermodulation (IM) products (in order to comply with SEM and spurious emis- 78 sion requirements). In contrast, in option 3, each carrier is amplified by a separate PA, and the two carriers are added up in the combiner. This tends to reduce the IM products and thus lower the required PA back-off (e.g., if two PA outputs are sufficiently well isolated from each other). Note that the use of a combiner results in insertion loss of at least 3 dB and therefore doubles the power consumption. For options 1 and 2, this is not an issue since the power levels of combined signals are even lower. In this section, the amount of PA back-off required to satisfy the unwanted emission requirements is evaluated. The minimum PA back-off of a transmitter heavily depends on the air interface of LTE UL, which is characterized by discrete Fourier transform spread orthogonal frequencydivision multiplexing (DFTS-OFDM) [2]. Therefore, the transmitter supports flexible resource allocation in frequency. The frequency domain resource allocation keeps the cubic metric (CM) from being used as an indicator of the required PA back-off, since the IM products do not always fall into the OOB domain. For this reason, the maximum allowable PA back-off of an LTE terminal is generally specified in terms of resource allocation as well as modulation order [4]. The CA operation tends to increase the minimum PA back-off, especially when the gap width is so large that the IM products of carriers occur far apart from the carriers (e.g., in the spurious emission domain). In the case of intra-band NC CA, the required PA back-off is generally maximized when the UL resource is allocated on the outermost edges (since the IM products reach as far as possible). For example, assuming two 5 MHz carriers with 5 MHz gap and option 2, the third order IM products (referred to as IM3 products hereafter) fall into the spurious emission domain with the outermost edge allocation (Fig. 4a), whereas they fall into the SEM domain with the innermost edge allocation (Fig. 4b). (Note that the measurement bandwidth of power spectral density [PSD] is either 1 MHz [red curve] or 30 kHz [blue curve].) Hence, it is clearly shown that the outermost edge allocation leads to even larger minimum PA back-off than the innermost edge allocation for the same resource allocation (0.9 MHz per carrier in this example). However, two aggregated carriers with sufficiently large gap generally require relatively large minimum PA back-off, regardless of whether the UL resource is on the outermost or innermost edges, since (an entire portion of) the IM3 product belongs to the spurious emission domain. In addition, the minimum PA back-off generally decreases with resource allocation, since the IM products spread out more in frequency, lowering the PSD for a given PA back-off. (Note that both the SEM and the spurious emissions are specified in terms of PSD, not the absolute power level [4].) For example, the allocation of 2.7 MHz per carrier (Fig. 4c) leads to smaller minimum PA back-off than the allocation of 0.9 MHz per carrier (Fig. 4a). Assuming random resource allocations, Fig. 4d shows the simulation results of the minimum PA back-off for intra-band NC CA as a function IEEE Communications Magazine • December 2013
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 79 Transmitter Receiver Option 1 LNA PA1 Digital D/A baseband1 LO1 Digital D/A baseband1 LO1 A/D LO1 Digital baseband1 A/D Digital baseband1 A/D Option 2 Digital baseband2 A/D Digital baseband1 A/D Digital baseband2 C BIAS Option 1 CSF Option 2 LNA PA1 LO1 Digital D/A baseband2 Digital D/A baseband1 Option 3 LO2 LO1 LO2 PA1 LO1 Option 3 LNA PA2 Digital D/A baseband2 LO2 LO2 (a) Switch unit 3 bits ADC LNA LO HB ADC ADC IF filter IQ2 7 bits 3 bits IF mixer 3 bits ADC LO DIG IQ3 7 bits ADC CLK LO LB 0.7-1.5 GHz 4x64bits DCXO LO IF DIV SEQ 26 MHz 195-390 MHz +1,2,4,8,16 LDOs LO IF 3 bits 4.75-34.75 MHz Formatter scrambler LO DIG LVDS drivers TESTMUX DIG CTRL SPI CSF ADC CSF ADC LO LB 312 MHz BIAS ADC DCXO ADC LNA IF FILTER TESTMUX IF MIXER 3 bits SEQUENCER 5-20MHz LO HB 1.8-2.7 GHz RFHB RF HB 3 bits DIG CTRL SPI RFHB IQ1 7 bits RF LB (b) Figure 3. Terminal implementation: a) transmitter and receiver architectures; b)implementation example of receiver architecture (option 2). of resource allocation as specified in 3GPP [5]. It is clearly shown that the envelope of minimum PA back-off values decreases linearly with resource allocation. Here the PSD of two carriers is assumed to be equal. In general, the envelope of minimum PA back-off values depends on the PSD ratio [6]. It is also worth mentioning IEEE Communications Magazine • December 2013 that, given the resource allocation and PSD ratio, the minimum PA back-off varies substantially depending on the resource allocation ratio across carriers. The resource allocation ratio that maximizes the required PA back-off is approximately inversely proportional to the PSD ratio. For example, if carrier 1 has twice as high PSD 79
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 80 20 PSD [dBm / 1 MHz] or [dBm / 30 khz] PSD [dBm / 1 MHz] or [dBm / 30 khz] 10 0 -10 -20 -30 -40 -50 -60 10 0 -10 -20 -30 -40 -50 -70 -40 -30 -20 -10 0 10 Frequency [MHz] 20 30 40 -40 -30 -20 (a) -10 0 10 Frequency [MHz] 20 30 40 (b) 14 0 Minimum required PA backoff [dB] PSD [dBm / 1 MHz] or [dBm / 30 khz] 10 -10 -20 -30 -40 -50 -60 -70 -40 -30 -20 -10 0 10 Frequency [MHz] 20 30 40 5 MHz gap 10 MHz gap 20 MHz gap 12 10 8 6 4 2 0 0 (c) 1 2 3 4 5 6 Total resource allocation (Hz) 7 8 9 106 (d) Figure 4. Minimum PA back-off for intra-band NC CA for Option 2: a) outermost edge allocation of 0.9 MHz per carrier (11.0 dB backoff); b) innermost edge allocation of 0.9 MHz per carrier (3.4 dB backoff); c) outermost edge allocation of 2.7 MHz per carrier (8.6 dB backoff); d) PA simulation with random resource allocation with respect of inter-carrier gap. as carrier 2, the required PA back-off is maximized when the resource allocation of carrier 1 is half that of carrier 2. Considering the impact of PA back-off on UL coverage, it can be concluded that the UL scheduling decision of the base station has a significant impact on the UL coverage. For example, it is desirable to avoid the resource allocation on the outermost channel edges if the terminal would be power-limited with the corresponding PA back-off. However, it is not always possible to avoid such resource allocation, for example, since the UL control transmission (PUCCH), if any, is typically small resource allocation scheduled around the edges of UL channel bandwidth [2]. RECEIVER RECEIVER ARCHITECTURE Assuming two contiguous or non-contiguous carriers, the right side of Fig. 3a illustrates various receiver architectures differentiated by where 80 the carriers are separated from each other: at a digital baseband (option 1), at an intermediate frequency (IF) chain (option 2), or at an RF chain (option 3). The numbers of RF chains, IF chains and baseband chains vary with choice of receiver architecture. For example, option 1 consists of a single RF chain, a single IF chain, and multiple digital baseband chains, whereas option 3 consists of two receiver chains, each consisting of an RF chain, an IF chain, and a digital baseband chain. Options 1 and 2 are equipped with a single RF LO whose center frequency may be set to the center of the whole frequency range spanning two carriers that are down-converted to IF simultaneously. An example of option 1 is a single wideband homodyne receiver that processes the down-converted frequency range as a single carrier signal. An example of option 2 is a heterodyne receiver where an IF receiver is dedicated to each carrier as shown in [8], which is more detailed in Fig. 3b. On the other hand, option 3 is equipped with two RF LOs whose center fre- IEEE Communications Magazine • December 2013
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 81 quencies may be set to those of two carriers. Thus, each receiver chain down-converts a carrier to baseband separately. An example of option 3 is a receiver where one homodyne receiver is dedicated to each carrier. Option 1 is the least complex architecture, for example, in terms of chip size and power consumption since it has only a single receiver chain. However, it may have design issues such as gain control of analog circuitry, especially when there is a significant time delay among carriers. This receiver architecture is generally well suited for contiguous CA. The other extreme, option 3, is the most complex, since each carrier has its own receiver chain. This receiver architecture is well suited for inter-band CA. Each of the receiver architectures has its own pros and cons. For instance, option 1 tends to consume less battery power, while it is more susceptible to the adjacent channel interference. In the remainder of this section, the radio impairments affecting the receiver performance will be discussed. Again, for simplicity, intraband NC CA with two carriers is assumed, although most of the arguments are generally applicable to other CA types. Uplink Downlink (a) Uplink Downlink (b) Downlink Uplink LO (c) Uplink Downlink TRANSMITTER EMISSION The unwanted emission from the transmitter may cause non-negligible interference to the receiver of the same terminal. Although this may also be true for single-carrier operation, it is even more likely for CA operation. For instance, considering a scenario with two UL carriers located at a distance of integer multiples of the inter-carrier spacing from a DL carrier, the IM products of two UL carriers fall on top of DL carriers. In particular, when the inter-carrier spacing is half the duplexer distance, the IM3 product falls on top of the close-in DL carrier, possibly causing severe interference to the receiver, as shown in Fig. 5a. The presence of even a single UL carrier may cause severe interference to the receiver because of limited duplexer isolation. If the close-in UL carrier is configured as the PCC by the network (as shown in Fig. 5b), the minimum distance between UL and DL carriers is given as the duplexer distance minus the inter-carrier spacing; thus, it may be much smaller than the duplexer distance. In an extreme case where two UL carriers are located on both ends of the transmit band, the gap between UL and DL carriers is as small as the duplexer gap. In general, the emission due to spectral regrowth from the transmitter increases with the resource allocation in UL. For example, the allocation of 5 MHz (Fig. 6a) creates more emission than the allocation of 1.8 MHz (Fig. 6b). Thus, it is possible to keep the interference to the receiver below a certain level by limiting the resource allocation in UL. However, even small resource allocation may create non-negligible interference at a small frequency offset [9]. For example, when small resource is allocated around the close-in edge of UL channel bandwidth, the IM products of UL transmission, and its IQ image appears at a small frequency offset, as illustrated in Fig. 6c. On the other hand, when small resource is allocated around the far-off edge of IEEE Communications Magazine • December 2013 LO (d) Uplink Downlink (e) Uplink Downlink LO (f) Uplink Downlink (g) Figure 5. Receiver impairments: a) transmitter emission with one UL carrier; b) transmission emission with two UL carriers; c) transmitter phase noise; d) receiver phase noise; e) receiver nonlinearity; f) LO coupling or harmonic rejection; g) image rejection. UL channel bandwidth, the UL transmission creates the counter IM3 (CIM3) onto a DL carrier, which tends to be about 20 dB weaker than the aforementioned IM3 product of UL transmission and its IQ image. The UL data transmission (PUSCH) may be such small resource allocation 81
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 82 In order to avoid severe desensitization, the spur at the IM3 frequency should be about 80 dB smaller than the desired LO signal. It is worth mentioning that the LO coupling may be dependent on the radio frequency planning, e.g., through the VCO coupling. around the edges of UL channel bandwidth, depending on the network scheduling. In addition, the PUCCH transmission, if any, is typically small resource scheduled around the edges of UL channel bandwidth together with frequency hopping over two slots (with each slot spanning 0.5 ms). The impact on the DL data reception (PDSCH) also depends on the network scheduling, since the interference hits only a small fraction of the channel bandwidth. However, the DL control reception (PDCCH) may be most likely interfered, since it generally spans the whole channel bandwidth [2]. Assuming that the close-in UL carrier is configured as the PCC by the network and the resource allocation starts from the nearby edge of UL channel bandwidth, the amount of transmitter emission on the close-in DL carrier is evaluated in Fig. 6d with respect to UL resource allocation. The transmitter emission is measured at the antenna port of terminal. The simulation parameters are set as follows: • Duplexer distance : 80 MHz • Duplexer gap : 15 MHz • Receiver noise figure: 6.5 dB • Insertion loss: 5 dB It is clearly shown that the receiver performance is severely affected by the transmitter emission when the inter-carrier gap is extremely large. PHASE NOISE The ideal implementation of LOs provides the terminal (both transmitter and receiver) with a sinusoidal waveform (i.e., a single tone) whose frequency is set to the center frequency of the wanted signal. However, the desired LO signal is accompanied by a phase noise that decreases with the offset from the desired frequency. In the case of intra-band NC CA, the impact of phase noise becomes more pronounced, since the minimum distance between UL and DL carriers may be far smaller than the duplex distance (depending on the inter-carrier spacing). The phase noise of transmitter LOs causes interference to a DL carrier (referred to as inband noise) after the up-conversion of a transmitter, as illustrated in Fig. 5c. In general, the closer the UL carrier is to the DL carrier (i.e., the smaller the minimum distance between UL and DL carriers), the more interference the phase noise causes to the DL carrier. Similarly, the phase noise of receiver LOs causes interference from a UL carrier (referred to as reciprocal mixing) after the down-conversion of the received signal, as illustrated in Fig. 5d. Again, the interference due to phase noise decreases with the distance between UL and DL carriers. Assuming the phase noise of –143 dBc/Hz at the offset of 20 MHz and the slope of –20 dB per decade [10], the amount of interference due to phase noise is evaluated as a function of intercarrier gap in Fig. 6d. It is shown that the phase noise may dominate the receiver performance loss (e.g., when the inter-carrier gap is between 45 and 55 MHz). on the DL carriers. The interference due to nonlinearity of analog baseband tends to become stronger with a smaller minimum distance between UL and DL carriers, since the channel selection filter provides less attenuation of the UL carrier. However, considering the wide-band characteristics of the RF front-end (e.g., mixer), the interference due to receiver nonlinearity is mostly independent of the distance between UL and DL carriers, as illustrated in Fig. 6d. Here the receiver’s second order intercept point (IIP2) [3] is assumed to be 45 dBm. It should be noted that the receiver nonlinearity often dominates the receiver performance loss (e.g., when the inter-carrier gap is smaller than 45 MHz). LO COUPLING In the case of options 2 and 3, the simultaneous operation of multiple LOs on the same die may cause severe coupling between LOs. More specifically, because of limited isolation between LOs, an LO signal applied to one mixer may also appear, at an attenuated level, at the LO input of the other mixer. Taking option 3 as an example, at the mixer for carrier 1, other than the desired LO signal at the frequency of carrier 1 (LO1), several additional spurs may appear, which include the LO signal at the frequency of carrier 2 (LO2) and the IM products of these two LO signals. For example, the latter spurs may occur when two LO signals pass through nonlinear devices such as an LO buffer. The relative strength of these spurs is subject to the receiver implementation. These spurs appear at the mixer, thereby resulting in unwanted frequency conversion. For example, if the inter-carrier spacing is equal to half the duplexer distance, the unwanted frequency conversion causes interference from a UL carrier to a DL carrier, as illustrated in Fig. 5f. In order to avoid severe desensitization, the spur at the IM3 frequency should be about 80 dB smaller than the desired LO signal. It is worth mentioning that the LO coupling may be dependent on the radio frequency planning (e.g., through the VCO coupling). HARMONIC REJECTION The widely used switched mixer typically leads to non-negligible conversion at harmonics of the LO center frequency. When it comes to an RF LO, such frequency conversion is not so problematic, since most of the interfering signals at the harmonics are easily suppressed by the duplexer (receiver) filter. However, option 2 cannot rely on the selectivity of a duplexer as the harmonic frequencies of the IF LO may be within the DL band. The corresponding frequency conversion is very similar to the frequency conversion due to LO coupling (Fig. 5f) in the sense that a couple of spurs occur around the desired LO signal. Harmonic rejection mixers can be used to mitigate such interference [11]. The measurement results on the left side of Fig. 6e show that it is possible to achieve a harmonic rejection ratio of 85 dB. RECEIVER NONLINEARITY 82 IMAGE REJECTION Receiver nonlinearity may cause further interference from a UL carrier to a DL carrier. In particular, a UL carrier may create an IM2 product The presence of gain and phase imbalance of receiver (quadrature) paths leads to limited image rejection of the receiver. IEEE Communications Magazine • December 2013
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 83 -60 PSD (dBm/7.5 kHz) -40 -60 PSD (dBm/7.5 kHz) -40 -80 -100 -80 -100 -120 -120 -140 -140 0 10 20 30 40 50 60 70 Distance from UL center frequency (MHz) 80 0 10 20 30 40 50 60 70 Distance from UL center frequency (MHz) 80 (b) (a) -50 TX leakage - 1RB TX leakage - 10RBs TX leakage - 25RBs Thermal noise TX phase noise RX IM2 RX phase noise -40 -60 Noise on DL center [dBm] PSD (dBm/7.5 kHz) -60 -80 -100 -120 -70 -80 -90 -100 -110 -120 -140 0 10 20 30 40 50 60 70 Distance from UL center frequency (MHz) 80 -130 0 1 2 3 4 5 Inter-carrier gap (MHz) (c) 6 7 x 107 (d) Carrier 1 0 70 -10 65 -20 Image rejection ratio (dB) Conversion ratio (dB) -30 Carrier 2 -40 85dB -50 -60 -70 -80 -90 60 After calibration 55 50 45 40 -100 Before calibration 35 -110 1800 2000 1900 Frequency (MHz) (e) 2100 2621 2626 2631 2636 Frequency (MHz) 2641 (f) Figure 6. Impact of receiver impairment: a) transmitter emission with 5 MHz allocation; b)transmitter emission with 1.8 MHz allocation; c) transmitter emission with 180 kHz allocation; d) interference as a function of intercarrier gap; e) harmonic rejection (measurements); f) image rejection ratio (measurements). IEEE Communications Magazine • December 2013 83
  • PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 84 The simulation results show that receiver performance is mostly determined by either transmitter emission or phase noise for a large inter-carrier gap, whereas it is dominated by receiver nonlinearity for a small inter-carrier gap. In the case of options 1 and 2, one DL carrier is seen as the image of the other DL carrier with respect to the center frequency of RF LO (e.g., LO1 in Options 1 and 2). If two DL carriers have unequal bandwidth, an unwanted signal adjacent to the narrowband DL carrier may interfere with (a fraction of) the wideband DL carrier, as illustrated in Fig. 5g. Recall that the unwanted signal may be substantially stronger than the DL carriers, since it may come from a different operator, as mentioned earlier. Thus, the image rejection of a conventional receiver may not suffice to protect the wanted DL carriers. Note that this is not a problem with two carriers with equal bandwidth, since the image always comes from the other carrier whose power level is typically similar. It is possible to avoid the interference due to the image by calibrating the gain and phase of analog circuitry, as depicted in [8, 12]. The measurement results on the right side of Fig. 6e show that a digitally calibrated receiver provides more than 55 dB image rejection ratio. (It is worth mentioning that further improvement of image rejection ratio is possible, e.g., by digitally compensating for gain and phase imbalance [13].) In contrast, option 3 does not have such an image issue (regardless of whether two carriers have equal bandwidth or not), since each carrier is down-converted by a dedicated LO. [2] E. Dahlman, S. Parkvall, and J. Sköld, 4G LTE/LTEAdvanced for Mobile Broadband, Elsevier, 2011. [3] B. Razavi, RF Microelectronics, Prentice-Hall, 1988. [4] 3GPP TS 36.101, “Evolved Universal Terrestrial Radio Access (E-UTRA); User Equipment (UE) Radio Transmission and Reception,” v. 11.2.0, Sept. 2012. [5] 3GPP TR 36.823, “Evolved Universal Terrestrial Radio Access (E-UTRA); Carrier Aggregation Enhancement; UE and BS Radio Transmission and Reception,” v0.5.0, Jan. 2013. [6] 3GPP R4-125599, “MPR Simulations for NC Intra-Band CA,” Oct. 2012. [7] C. S. Park, “Dependence of Power Amplifier Backoff on Resource Allocation for Non-Contiguous Carrier Aggregation,” Electronics Letters, vol. 49, no. 15, July 2013, pp. 962–64. [8] L. Sundström et al., “A Receiver for LTE Rel-11 and Beyond Supporting Non-Contiguous Carrier Aggregation,” Proc. IEEE ISSCC ’13, San Francisco, CA, Feb. 2013. [9] 3GPP R4-123306, “Reference Sensitivity for Non-Contiguous Intra-Band CA,” May 2012. [10] D. B. Leeson, “A Simple Model of Feedback Oscillator Noise Spectrum,” Proc. IEEE, vol. 54, Feb. 1966, pp. 329–30. [11] L. Sundström et al., “Complex IF Harmonic Rejection Mixer for Non-Contiguous Dual Carrier Reception in 65 nm CMOS,” to be published (invited paper), IEEE J. Solid-State Circuits. [12] L. R. Wilhelmsson et al., “Design of a Configurable Analog Receiver Front-End Supporting LTE Carrier Aggregation,” Proc. IEEE VTC ’13, Dresden, Germany, June 2013. [13] C. S. Park and F. S. Park, “Digital Compensation of IQ Imbalance for Dual-Carrier Double Conversion Receiver,” IEICE Trans. Commun., vol. E95-B, no. 5, May 2012, pp. 1612–19. BIOGRAPHIES CONCLUSION An overview of design challenges of CA-capable terminals is presented in this article. Different radio architectures are discussed from the perspective of design trade-off. The simulation results show that the support of CA generally leads to significant increase of minimum PA back-off in order to meet the unwanted emission requirements. In addition, the radio impairments affecting the receiver performance are discussed, which include transmitter emission, phase noise, receiver nonlinearity, LO coupling, limited harmonic rejection, and limited image rejection. The simulation results show that receiver performance is mostly determined by either transmitter emission or phase noise for a large inter-carrier gap, whereas it is dominated by receiver nonlinearity for a small inter-carrier gap. The silicon implementation shows significant enhancement of harmonic rejection and image rejection capability. ACKNOWLEDGMENT The authors would like to thank Peter Jakobsson and David Duperray from ST-Ericsson for their insightful comments. REFERENCES [1] K. I. Pedersen et al., “Carrier Aggregation for LTEAdvanced: Functionality and Performance Aspects,” IEEE Commun. Mag., vol. 49, no. 6, June 2011, pp. 89–95. 84 C HESTER S UNGHCUNG P ARK (sungchung.park@gmail.com) received his Ph.D. degree from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, in 2006. After about two years with Samsung Electronics Inc., Giheung, Korea, he joined Ericsson Research, USA, where he worked on 3GPP-LTE digital baseband/frontend and participated in 3GPP-LTE standardization of carrier aggregation and MIMO. Since 2013, he has been with Konkuk University, Seoul, Korea, working on algorithm and system-on-a-chip (SoC) architecture for digital signal processing. L ARS S UNDSTRÖM received his Ph.D. in applied electronics from Lund University, Sweden, in 1995. From 1995 to 2000 he was an associate professor at the Competence Center for Circuit Design at the same university where his research focused on linear radio transmitters and RF ASIC design. In 2000, he joined Ericsson Research where he presently holds the position of senior specialist with interests ranging from RF, analog, and mixedsignal IC design to radio architectures for cellular transceivers. A NDERS W ALLÉN received his Ph.D. in automatic control from Lund University in 2000. Since then, he has been with Ericsson Research, currently holding a master researcher position. He has participated in 3GPP standardization for both HSPA and LTE, and has primarily worked with terminal front-end requirements and physical layer algorithms. ALI KHAYRALLAH received M.S. and Ph.D. degrees from the University of Michigan, Ann Arbor, and his B.E. degree from the American University of Beirut. He is currently director of research at Ericsson in San Jose, California, and has held various research positions with the company since 1995. Previously, he was on the faculty of the Electrical Engineering Department at the University of Delaware. IEEE Communications Magazine • December 2013