The German Copper Institute, DKI, welcomes you to its presentation “New Loads on Old Systems”.
Who are we? DKI is the German Copper Institute, the central information and advisory service in Germany dealing with all aspects of the production, use and purchase of copper and copper alloy. We deal with everything from simple enquiries about proof of delivery documents to detailed queries about, say, material properties that often require us to research the specialist literature relevant to them. DKI is open to anyone and everyone – commercial companies, the trade professionals, the arts and handicrafts sectors, industry, academia, students, school children and the general public. The Institute is supported by its members – two copper fabricators and around 45 companies all of which are semi-manufacturers. Only just recently DKI recruited our first member company from within the electrical engineering sector.
Although DKI has been operational for nearly 75 years and despite the fact that electrical engineering has always been the dominant area of application – accounting for between 50 and 65% of all copper products – it is only recently that the DKI has begun to actively address the problems associated with the use of copper in the electrical and electronics sectors. Historically, there was perhaps the feeling that copper would always be in demand for electro technical products because there were and are so few alternatives. The only other materials suitable for use as electrical conductors are silver and aluminium. Silver is not viable for cost reasons and aluminium has such specific advantages and disadvantages compared to copper that it is functionally competitive in only a very few localized areas of the electrical engineering industry. However, on the last decade it has become more apparent that there may be applications in which better results can be achieved by specifying in general terms greater copper densities. Studies were carried out that identified two major areas of electrical engineering in which significant savings can be made by using more rather than less copper. First, the lower current density that can be achieved by increasing the cross-sectional area of conductors in electric motors, transformers and cables can cause such a significant reduction in power loss that in certain cases the extra investment costs incurred are recoverable in less than a year. Second, modern electronic devices and equipment have load characteristics that are very different to those of conventional loads. As a result, power system configurations that served their purpose well for decades are now unable to cope with the demands now being placed upon them. It is this second area that we wish to address in today’s presentation. Nearly all electronic equipment needs to be supplied with ripple-free direct current. Rectification by simply reversing the polarity of one half-wave of the sinusoidal a. c. supply may well be sufficient to drive a d. c. motor, but it will never be adequate for supplying an electronic circuit. The zero-crossing period is therefore bridged by a capacitor that is itself recharged only during the relatively short time when the line voltage is near its maximum. It is only within this fairly small time window that current is actually drawn from the mains, and the magnitude of these short current bursts is correspondingly large. A sinusoidal supply voltage thus drives a current whose waveform is no longer anything like sinusoidal. This is a relatively new phenomenon and has major repercussions as we shall see.
Despite the rumours that one often hears to the contrary, poor power quality is only very rarely attributable to the effects of higher power use. It is far more likely to be caused by the load itself. But why should that be the case? And what can one do about it?
Originally, the three sine wave voltages of a three-phase system drove three currents in the three phase wires. Whilst these currents could be displaced in time relative to the voltage due to the absorption of reactive or “wattless” power, they were all sinusoidal in form. If three identical single-phase loads were then connected to the phase wires of the power system, the values of the three currents at any one instant would sum to zero. Today, things look quite different.
Let’s now consider the example of one of the older types of electronic ballast designed for a 58 watt fluorescent lamp. The first thing the ballast does is to rectify the (yellow) incoming a. c. voltage and then smooth it by means of an electrolytic capacitor. This smoothing or “reservoir” capacitor charges up to almost the peak value of the line voltage – as shown by the blue curve. About 180 milliamps of direct current must be drawn from the capacitor in order to generate sufficient power for the lamp. However, current from the supply network can only flow during the short periods in which the magnitude of the (yellow) rectified a. c. voltage is greater than the (blue) residual voltage on the capacitor. Using the power supply parameters shown here, the input current in the circuit reaches a peak value that is 15 times the mean average value – far more than the square root that we would typically expect for purely sinusoidal waveforms! On the other hand, the current has a very small conduction angle, that is, the circuit draws current from the line only during a relatively small portion of the period. For most of the semi-wave there is absolutely no current flow from the supply to the circuit. Measurements of real-life applications confirm that the example just discussed is not unrealistic. In fact it is a close reflection of what is observed in practice. If a conventional linear load such as an incandescent light bulb is connected to a sinusoidal voltage, the resulting current waveform will also be sinusoidal. If three such identical bulbs are connected to the three outgoing lines of a three-phase four-wire system we would have a balanced system in which the current in the neutral conductor would be practically zero.
There is a lot of things that could be said about electronic ballasts questioning their supposed advantages and which nobody says, but one thing that is always quoted as their drawback is actually not true: Everywhere everybody mentions electronic ballasts as the first example of loads that do not behave in such a linear way. Yes, this used to be so during the first days of electronic ballasts but today’s standards prohibit this. Only in the range of small models rated less than 25 W, including compact fluorescent lamps (CFLs or energy saver lamps), the standards are very generous and allow for such great current distortion. Like them, an increasing number of other electronic devices legally do this, even if their power intakes are substantially higher. But let us stick to the old electronic ballast as an example for these comparisons, since here one and the same load, a fluorescent lamp, can be operated in either the conventional or the electronic way.
So if instead of three incandescent bulbs we connect three fluorescent energy saver lamps to the three-phase supply (one on each line), there will be no overlap between the current peaks in one line and the troughs in another. As a result there is no cancellation – not even partial cancellation – of the return currents in the neutral conductor. How could there be? If the situation never arises in which one current flows back just as another is flowing in the opposite direction, then there will never be any mutual cancellation of currents. In fact what is observed is that the two peaks per period in each phase give rise to six peaks per period in the neutral conductor. The neutral current is thus nothing like zero and its average magnitude is three times that in one of the phase conductors! Even expressed in terms of the root mean square value, the neutral current is still greater by a factor of square root of 3. That’s probably the reason why the VDE regulation 0100 continues to allow N and PEN conductors with reduced cross-sections. Of course, a cynic would point out that large neutral currents are not a problem as after all, the PEN conductor isn’t fused! The clue to the origin of this large neutral current lies in its frequency of 150 hertz. If it was due to an unbalanced load, then it would have a frequency of 50 hertz and would also be much smaller in magnitude. This has six different and quite general consequences, and in TN-C or TN-C-S earthing systems there are four additional pitfalls that one must be aware of. In what follows we shall address each of these in turn.
Such large deviations in the current waveform from a purely sinusoidal profile will of course interact with the power system impedance to create a highly distorted voltage profile. Calculations indicate that a mere twenty of the old 58-watt electronic ballasts mentioned earlier will, when connected to a phase conductor of a typical power distribution system, deform the voltage waveform in the manner shown here. Today’s electronic ballasts no longer work on this principle. Instead the input current is switched at a high-frequency allowing the sinusoidal waveform to be approximately recreated by a step-like function. Nevertheless, the technique of the older-style electronic ballasts is still state-of-the-art in many other electronic devices such as the compact fluorescent energy saver lamps, already mentioned, that have power ratings of up to 25 W, or the switched-mode power supplies used in personal computers and their monitors. In what follows we will continue to use the older-style electronic ballast to illustrate the effects of non-linear loads. One good reason for doing this is that it allows a direct comparison with conventional technology, as the same fluorescent tube can be driven by a standard magnetic ballast or by an improved low-loss ballast.
Once again, measurements of real supply networks demonstrate that these computed voltage distortions are not alarmist but point to the existence of a very real problem. At the top of this chart is shown the current in an outgoing feeder from the main distribution board to a floor distribution panel board within a large storage, dispatch and administrative building. About one half of the total current is clearly made up of currents in such non-linear, distorting loads, while the other half arises from currents in conventional linear loads. And at the bottom you see what the voltage profile looks like. The distortion in the voltage waveform is considerably greater than one would assume from the current distortion. This tells us that the distortion is due in part to the upstream medium-voltage system – but more about that later.
Before that let us take a look at the input current of a small television set. The input power is a mere 40 watts, which implies a mean current magnitude of about 170 milliamps. For a purely resistive load, we would therefore expect a peak current of around 250 milliamps. The actual peak value though is 4 amps! We now have concrete confirmation of our earlier theoretical calculation, namely that one cannot downsize the smoothing capacitor when designing a TV set or similar device, because the peak current drawn by such loads actually surpasses the factor of 15. We get the same picture of course for a three-phase load which is made up in the same way as three single-phase loads. In the example shown here, we see the input side of a variable speed drive. Typically though a three-phase power converter is not connected to the N-conductor so the problems encountered tend to be somewhat different. And here is the harmonic spectrum of the neutral current. What can we learn from this graph?
The French mathematician Jean Baptiste Fourier discovered that any non-sinusoidal periodic waveform can be written as the sum of sinusoidal waves whose frequencies are whole multiples of the fundamental frequency. The fundamental frequency referred to here is understood to be that of the synthesized non-sinusoidal waveform. One should also note that the even multiples of the fundamental frequency – the so-called even harmonics – are not present in most applications of technical interest, and that the amplitudes of the odd or prime number harmonics lower as the frequency increases. The word “harmonic” might seem something of a misnomer when one realises that they are responsible for causing a very great deal of disharmony in power distribution systems. So let’s first look at how a triangular wave can be synthesized from its fundamental frequency and its harmonic components. At first you see the fundamental or first harmonic. Now let’s add the third harmonic. The second harmonic is not part of the triangular waveform. The fourth is also missing as are all the other even harmonics. The thick blue line represents the sum of the components and the more harmonics that are included, the closer their sum corresponds to the target triangular waveform. The contribution of the higher harmonics becomes smaller the higher the harmonic order as can be seen by the decreasing amplitudes or rms values, here rounded to the nearest percent. The negative signs simply indicate that the respective harmonic begins with a negative semi-wave. Summing the squares of the individual rms values for each component harmonic and then taking the square root of the sum computes the overall rms value. And we will expand on this later. The fundamental frequency and the third to the seventeenth harmonic have now been included in the Fourier synthesis. As you can see the synthesized waveform really does look quite triangular. In theory the series would carry on indefinitely, but for most practical applications one can stop at this point as the fraction of the total current due to higher order harmonics is essentially negligible.
But is that always the case? Let’s now look at the Fourier synthesis of a square wave. The component harmonics are qualitatively the same as those in the triangular waveform. However, closer inspection reveals that the fundamental is somewhat smaller than that in the triangular wave. Once again the even harmonics are not needed in the synthesis and the thick blue line again represents the sum of all the contributing harmonics. But if you focus on the amplitudes of the components, you’ll notice that each is significantly larger than its counterpart in the Fourier synthesis of the triangular waveform. And that is not surprising when you consider the fact that, optically, a square wave deviates much more than a triangular wave from a purely sinusoidal curve. But it is just that difference between the target curve being simulated and its fundamental sine curve that has to be compensated for by the higher order harmonics. The amplitudes also clearly reflect the much greater role played by the higher harmonics. The representation of the square wave achieved by summing the fundamental and higher harmonics is less convincing than in the case of the triangular waveform even though all odd harmonics up to the seventeenth have now been included. In this case it would have been advisable to incorporate higher order harmonics. In general, it is not possible to say how many harmonics should be included in the summation, as this will clearly depend on the waveform being synthesized and on the required degree of accuracy.
For periodic curves whose mathematical form can be described relatively simply, the Fourier coefficients – the peak amplitudes of the harmonic components – can be taken directly from a formula glossary. However, for waveforms that can only, if at all, be described by highly complex mathematical functions, no such tables of Fourier coefficients exist. Fortunately there is a simpler method we can use in such cases. To illustrate the approach, we shall return to today’s leading source of power system contamination – the PC and its monitor. Both devices are equipped with the same type of switched-mode power supply unit. Taken together, the two devices will have a smoothing capacitance of around 330 µF – that’s about one hundred times the capacitance found in a compact energy saver lamp. The power input for the PC and monitor is approximately 100 W which corresponds to between 4 and 15 energy saver lamps (depending on the wattage). The power supply has a typical source resistance of 0.5 and an inductance of 0.9 mH. The current that flows from the rectifier to the electrolytic capacitor has a very small conduction angle (that is, the current flows for only a very short period of time). The magnitude of the current is correspondingly large, with current spikes as we saw earlier more than ten times the rms current. On the a. c. side – that is, on the supply side – the current profile looks like the top diagram. Here, as so often in practical situations, we are dealing with a current waveform that cannot be described by a simple mathematical function and whose Fourier coefficients cannot therefore be found in a formula glossary. In this case, however, it is immediately apparent that the current profile is very similar to a triangular waveform with a pulse duty cycle of 1:7 and a peak magnitude of 2.8 A, as shown at the bottom. Using this simplification, we can now synthesize the current profile using standard tables.
If we do this and include harmonics up to order 17, with the bold blue line being the fundamental, we get the following result. As we did not include the higher harmonics, the apex of the triangle is not perfectly sharp. But that has the advantage of providing an even better simulation of the original profile than the approximation of a perfect triangle. We have here one of those rare cases in which the imperfect simulation provides a closer match to reality than the mathematically more precise approach – pure fluke though it is in this case.
Harmonics are key to understanding what is going on. Current harmonics are created within non-linear loads and are able to propagate upstream from the load, that is counter to the direction of the energy flow. Now, utility companies put a great deal of effort into generating what is effectively a pure sinusoidal voltage and any distortion that the voltage subsequently suffers has been inflicted on it by the load. This phenomenon is unique to electrical power. No other commodity is affected in this way and so it is quite understandable that many customers are unaware of the true source of their difficulties. Harmonic currents can therefore flow from the low-voltage to the medium-voltage power system. However, if the next distribution transformer can effectively block these higher frequency components, then the harmonic currents will flow into other attached loads. It is important to bear in mind that the voltage drop across the transformer is essentially inductive in nature. Thus a transformer with a rated short-circuit voltage of 4%, has a short-circuit voltage of almost 12% at a frequency of 150 hertz. At 250 hertz, this figure has risen to nearly 20%. This effect can be enhanced but can also be considerably ameliorated by selecting the right type of transformer vector group – a design aspect we will be returning to in the third presentation of this series. The point to remember at present is that the less the current harmonics are able to propagate, the greater the resulting distortion of the mains voltage from its ideal sinusoidal form.
The abbreviation THD stands for “total harmonic distortion” and is usually expressed as a percentage of the total rms value or the fundamental, but it is not uncommon to see it quoted in volts or A. Now it turns out that a voltage or current with a relative fundamental content of, say, 71% contains an additional 70% of harmonics. That clearly needs some explaining! So let’s take a brief look at how the THD value is calculated. As already mentioned, the root mean square value of a non sinusoidal waveform is computed by adding the squares of the individual sinusoidal components, as displayed here in the third column, and then taking the square root of the sum. The THD is calculated in exactly the same way except that the fundamental wave is not included in the sum, as displayed here in the fourth column. Now there are two conflicting understandings of relative THD values. THDr is related to the TRMS value, THDf relates only to the fundamental. Unfortunately the THDf has become the more popular value. In this way the share of something contained in something else may easily exceed 100%. Also in this example the content of distortion is substantially greater than the fundamental wave. This approach of rating may be quite misleading, for which reason THD will always be meant to stand for THDr from now on in these presentations. Fortunately the metering instrument can be set to show which value is displayed with setting visible in the display.
When analysing harmonics it is not just their relative magnitudes that are of interest. The flattened voltage profiles that are typical of power distribution systems containing a large fraction of switched-mode power supply loads can also be found in residential areas during the peak TV viewing hours on Saturday evenings. Both of these should actually be pure sine curves. The rms values indicated on the plots correspond very closely to the rated values, though the peaks appear to be about the same size. If one takes into account the scaling used for the plots (100 V per division and 200 V per division respectively), then the peak voltage ratio is almost 1-to-2 rather than the 1-to-“root 3” theoretically required. Note also that the degree of distortion in the line-to-neutral voltage is very close to that found in the voltage between phases. This is because the rms values of the component harmonics – here dominated by the fifth order mode – are of similar size. At first sight, this may seem incompatible with the clear differences between the two measured voltage profiles. However, the harmonic spectra also include the corresponding phase angles, and these are mutually shifted by 180 degrees. To illustrate what is happening, look at the theoretical simulations on the right. Both simulations represent the sum of the fundamental mode and the fifth harmonic, whose amplitude is 5% of that of the fundamental. In the one case, the fundamental and harmonic are in phase, that is there is a phase shift of zero. In the other case, the two component waves are phase shifted by 180 degrees. These phase shifts are similar to those recorded in the measurements shown on the left. Once again, the thick blue line represents the resulting sum curve and shows that phase shifts are the key to understanding why the two voltage profiles are so different.
Here you can see the extent of such voltage deformations in real power systems. These profiles, recorded by Professor Fender of Wiesbaden University of Applied Sciences, were not recorded at peak viewing hours at home, but during core working hours in Frankfurt’s banking district.
It was not in a banking area and not in an industrial but in a domestic precinct where these curves were recorded. At first sight it becomes evident that the stability of mains voltage (blue) is excellent, and the distorting loads obviously do not represent the major part of the entire load, otherwise a negative correlation between the line voltage and the degree of distortion THDU (red) had probably become evident. Rather, the two curves have an arbitrary or no relationship with each other. Now they are parallel, now they diverge. Apart from the technical issues, the measurement also allows for some sociological conclusions: The principal TV and computing day of the Germans is obviously neither Friday nor Saturday, as could have been expected, but Sunday.
Therefore let us have a clear look at that shortly, but let’s begin with Saturday. From the steep drop of the THDU just before the daily news on the second channel of German public TV, ZDF, at 7pm, sociologists might conclude a complete lack of interest the audience has in political matters, while media scientists will immediately point out that the steep increase at exactly 8pm, when the Daily Review on the first channel, ARD, illustrates the complete opposite. On top of this, ZDF offsets this viewing loss during its Today magazine that is screened at 9:45pm.
But in fact these findings are meaningless as on the following Sunday the trends are both the same and different. Both Today and Daily Review come up with a dip here. So let us leave this field to the social scientists, and let’s have a look at the technical details. These alone provide us with enough riddles to solve: In this picture the mains voltage and the THD were omitted, and instead the voltage harmonics of order numbers 2, 3, 5, 7, 9, 11, 13 and 15 are displayed individually. Surprisingly enough, their courses are anything but proportional to each other. E. g. at 9:40pm they all leap up, probably stemming from one common cause, that is same source type, in this case rectifier loads. Shortly afterwards, however, the harmonics number 3, 5 and 9 decrease steeply, while the eleventh stays about stable and the seventh continues to rise! This may be due to several causes: a different sort of consumer being activated, a change in load unbalance, network resonances and so forth. Drawing the similar graph for other days of that same week, would reveal more phenomena like this. For example, the third and fifth harmonics, which are very far apart throughout this Sunday, had about equal levels for several hours on the Saturday before. This is typical of a substantial unforeseen unbalance that occurs across the phases of rectifier loads for this period of time.
The second effect caused by electronic devices is the immense inrush currents when these non-linear devices are energized. At the moment of switching on, when the reservoir capacitor is uncharged, there is a virtual system short-circuit. Now you may be lucky and happen to switch on just as the supply voltage is passing through zero. In that case, the current pulse will look like the dark red curve. The peak is “only” about 35 A. The voltage on the capacitor – shown here in dark blue – overshoots somewhat due to the inductance of the power system. But on the other hand you may not be so lucky and you could switch on just as the supply voltage is at its peak. In that case, the inrush current will look like the bright red pulse. The peak current is now an astonishing 145 A! And the capacitor voltage is now calculated to overshoot to almost 550 V! An overshoot of that magnitude would, of course, not occur in practice, because the capacitor would be simply unable to block a voltage of that magnitude and would start to conduct, acting as a surge protection device. And any electrical technician will know just how good that is for the capacitor.
Inrush currents are indeed a real problem in power distribution systems. Here you can see in the middle plot the inrush current generated by a compact energy saver lamp with a nominal power of a mere 13 W. The peak voltage exceeds 25 A. In the magnified plot below with the expanded region up to 2 A, one sees that the current oscillates. This is a resonance phenomenon between the inductance of the power system and the reservoir capacitor. The upper plot shows the corresponding voltage. The voltage dip at switch-on is clearly evident. The flattened voltage profile also indicates that this electronic energy saver lamp is clearly not the only one being driven by this power system.
We now turn to the third problem in our list and to the question of how to measure alternating currents correctly. The root mean square value of an alternating or pulsating current is the value that a smooth (pure) direct current would have to have in order to cause the same heating effect in a fixed resistive load. As the heating effect is proportional to the square of the current averaged over one cycle of the waveform, we can state that for two currents with the same average magnitude, the heating effect will be larger in that current with the greater harmonic distortion and we will return to this point later and illustrate it with an example. In a moving-iron current meter, the pointer is driven by the repelling force of two homopolar magnetized soft iron strips. The repulsive force is proportional to the product of the magnetization of each iron strip. Because both strips are magnetized by the same current, the repulsive force is proportional to the square of the current. If the instrument has sufficient mechanical inertia, the time-averaged position of the pointer will be proportional to the rms current. A moving-coil instrument on the other hand does not have these properties. The force acting on the pointer is in this case linearly dependent on the current, that is, on its arithmetic mean. If the meter is fed via a bridge rectifier it will indicate the mean average magnitude of the current. Now there is not a great deal of difference between the price of a moving-coil meter and that of a moving-iron instrument. True you might say, but nowadays who uses analogue meters anyway? However, simply replacing an analogue instrument by a digital one is not as straightforward as it might seem. For a digital meter to measure the root mean square value it must be able to pick out the curve, square the instantaneous values obtained, sum the squares, take the square root of the sum and then divide this value by the number of discrete values – and that is not cheap. A digital meter that can do that will still cost you a lot more than what you’d pay for a simple average-value instrument – though prices for these digital meters have fallen substantially – but, of course, the price is nothing like the cost of the damage that can occur if measured current values are significantly wrong. And the inexpensive analogue meter can also call itself an “rms instrument” provided that the form factor was included in the calibration procedure. In the case of a purely sinusoidal waveform, the form factor reflects the fact that the root mean square value is 1.11 times greater than the mean value. Now whilst the voltages in our power systems are still roughly sinusoidal in appearance, the current profiles look wholly different – as you have already seen and will frequently see again. The apparent rms values of these non-sinusoidal waveforms are as a rule considerably smaller than the true rms values and we will give a quantitative example of this later on. If you want to measure a true rms value, you have to use a “true rms instrument”. The situation has got a lot in common with buying flour at a supermarket. You can purchase coarse-grain, medium-grain or fine-grain flour, though it’s not sold as “coarse”, “medium” or “fine” grade flour, but as “fine”, “extra fine” and “superfine”. You grab the fine one and get the coarse one. In this way, a meter that offers “rms response” will not display the rms value. Only a meter that’s marked as a “true rms” instrument will indicate the true rms value. Surprising and illogical it may be, but it’s important to be aware of this fact when assessing power quality – otherwise you might end up just measuring “flour power”.
Returning once again to our old-style 58-watt-electronic lamp ballast, you may recall that we calculated that a mean d. c. current of 180 mA was required in order to power the lamp. If we plug in the device characteristics and the power supply parameters given earlier into the iterative simulation, the program yields a mean current magnitude of 179 mA. The accuracy of the calculation is clearly adequate. Now physics dictates that the mean currents immediately in front of and immediately behind the reservoir capacitor must be identical, because averaged over a wave period the same number of electrons must flow onto the capacitor from the feeding side as are drawn off on the consumer side. If this wasn’t the case, the voltage on the capacitor would not be the same after each full wave. For a purely sinusoidal waveform we would therefore expect an rms current on each side of the capacitor of 1.11 times 179 milliamps, that is a value of nearly 200 mA. However, the value measured is more than three times larger! Measuring the “rms value” rather than the “true rms value” will therefore lead to an error not of a few percent, but of several hundred percent! The peak current is 2.7 amps, fifteen times the mean value! The apparent power supplied by the system is thus 141 VA but the active power flow is only 58 W. The rest is just reactive power due to distortion. In this case, the form factor of the power system current, which is defined as the ratio of the true rms to the mean, is 3.4. The crest (or peak) factor of the power system current, that is the ratio of the peak value to the true rms value, is about 4.4. For comparison, the form factor of a purely sinusoidal waveform is pi divided by twice the square root of 2, approximately 1.1. This is the factor with which the pseudo rms instruments multiply the mean current magnitude, presenting the result as the “rms current value”. In this case, an analogue or cheap digital multimeter would have served up a putative rms value of 200 millamps – far below the true rms value. The crest factor of a pure sinusoidal current is the well-known factor of “root 2” or 1.41, about one third of the value associated with the distorted current profile shown here.
Now let us compare an RMS to a True RMS meter. Here both of them are measuring the same current, one which is feeding the power supply unit of a laptop PC. Yes, what you see is true: The TRMS value on the red meter is four and a half times as high as the reprocessed mean on the grey meter! Oh dear, what a mean value! While there is really just a minor trick in getting the value that high: Usually the line voltage looks like what we see at the top of the PC screen, namely already flattened by the rectifier loads. This alone attenuates the distorting effect, making it appear less severe than would be the case with a pure sine wave. In contrast, a wave shape with excess peak that may also show up in existing networks, was used in this case to feed the power supply. This can be seen at the bottom of the PC screen and on the system analyzer on the left. Yet this scarcely exaggerates the effect any more than the first case disexaggerates it, compared to the pure sine wave. If we multiply this reprocessed absolute mean value by the 230 V line voltage, we merely reach 19 W, VA or whatever, by the way. This even falls far below the active power intake! Obviously this instrument with this waveform displays even a lot less than the absolute average mean value, as becomes apparent when attenuating the distortion simply by connecting a resistor in series. Now not only does the TRMS value drop by well over 20%, but apparently the average value leaps up to double the previous magnitude, while the active power (of course) remains the same!
How is it possible that the heating effect of two apparently identical currents can be so different? After all, a current of one ampere means that in one second 6.25 times 10 to the power of 18 electrons are flowing past every point within the conductor. But that is not necessarily the case when you’re dealing with the rms current. A simple example will help to illustrate why this is so: The continuous square wave current with a frequency of 50 hertz and a peak amplitude of 1 ampere – in so far as one can speak of the peak of a square wave – flows through a 1 ohm resistor. The voltage drop across the resistor is 1 volt and the power dissipated as heat is of course 1 watt. The electron flux per cycle is equal to 2 times one ampere times 10 milliseconds, that is 20 milliampere seconds per period. The thermal energy released is thus 1 watt times 20 milliseconds, that is, 20 milliwatt seconds per period. The same “amount of current”, by which we mean the same number of electrons, can be transported not only continuously but also in intermittent bundles. A very simple example would be transporting twice the number of electrons in half the time and zero electron flux during the remaining time. The current, when flowing, has a magnitude of 2 A and the voltage drop across the same 1 ohm resistor is now 2 V. The thermal energy released is now 2 times 4 W times 5 milliseconds, that is, 40 milliwatt seconds per period. The heating effect is therefore double what it was even though the mean electron flux is the same. A d. c. current that produces the same heating effect as that in the lower example would have to have a magnitude of 1.414 (that is, root 2) A. The voltage drop would then be root 2 V and the power would be 2 W. In the lower example, the a. c. current therefore has an rms value of 1.414 A and the rms voltage across the resistance is 1.414 V, because these are the values which in the case of a d. c. current would cause the same heating effect. What does this imply for the measurement of currents and voltages? Whilst the rms current is the critical value when, as is often the case, one is interested in heating effects in conductors, it is clearly not sufficient to restrict consideration to this one averaged parameter. The chemical and magnetic effects of an electric current grow linearly, not quadratically, with the electron flux. Using a bridge rectifier, the two currents shown on the left might be used to supply a charging current of 1 ampere to a rechargeable battery. If in the lower case you charge for one hour at 1.414 A – as measured by a true-rms ammeter – you will still only end up with a charge of one ampere-hour in your battery. The same situation arises with capacitors, although one is more likely to be dealing with milliampere-seconds of charge rather than ampere-hours despite the fact that it is the capacitor and not the rectifier that is responsible for the current distortion that one observes during the process of power supply rectification. And while we’re talking about rectification it’s worth noting that if these two currents are rectified so as to drive a d. c. motor, both will, when averaged over a complete cycle, generate the same torque and the same speed, i.e. the same shaft output. However, in the lower case on this slide, the ohmic losses in the windings are twice that measured in the upper case! Which measuring system one should use clearly depends on what it is you want to measure.
Back to our old favourite – the 58-watt fluorescent lamp – and to the reason why we keep using this as our example load. The reason is that a fluorescent lamp is a load that we can connect to the power supply in both a conventional and a modern electronic way. A fluorescent lamp is a gas-discharge path and demonstrates all the typical properties of such a system. A d. c. measurement will illustrate what we mean: The lamp has what one can call a negative resistance. As the current increases, the voltage across the lamp falls – just the opposite of what Ohm’s law dictates. That’s the reason why you shouldn’t connect these tubes directly to the power system. If you do, they tend to go bang! Now an attempt was made to model the observed behaviour using an empirical formula, and the approximation appears to be quite successful. Using this empirical formula, one can now try and simulate mathematically what happens when the fluorescent tube is connected to a magnetic ballast – provided the system voltage is still sinusoidal. The highly antinon-linear characteristic of the fluorescent tube means that the voltage across the tube (shown in blue on the plot) is very highly distorted and not even remotely sinusoidal in shape. However, as the discharge tube has a very large inductance – approaching 1 henry in this case – and is connected in series, the massively distorted voltage has only a minor effect on the current (here shown in red) and thus on power quality: The current profile is almost sinusoidal apart from the slight kink that is apparent every time it passes through zero. The current peak lags considerably behind the peak of the voltage curve because, as is well known, running a florescent lamp with a magnetic ballast results in a large reactive power loss – the standard, well-known, fundamental frequency reactive power that is nearly always compensated for in industrial applications. This phenomenon, too, cannot only be derived theoretically but also measured in practice.
As used in electrical engineering, the word “compensation” is a somewhat vague term. On the one hand we speak of “reactive power compensation” or “power factor compensation” and mean the fundamental reactive power that arises when voltage and current are both sinusoidal but are not in phase. Within any one period there are two ranges in which the voltage and the current have the same sign and in which the power – the product of voltage and current – is therefore positive and can conventionally be equated with energy consumption. There are, however, also two ranges in which voltage and current have opposite signs and in which the power is, from the point of view of the load, negative. During these times, the load does not act as a consumer but as a producer of electrical energy. Energy that was previously stored is now fed back to the power distribution system. Sometimes we also talk about “harmonic power compensation”. The principle is the same. If one wishes to record the power profile due to a single current harmonic, then one has to multiply that current component with the voltage present. If, for example, one wishes to compute the power associated with the third current harmonic and if one assumes that the line voltage is still an ideal sine wave, then the resulting power profile will be that shown by the black line and the sum of the areas under this curve over a single line period will be the energy transferred during that time. The sum of the areas equals zero, which means that no energy is transferred on average and that energy simply oscillates between the source and the load – exactly what we mean when we talk about “reactive” or “wattless” power.
Let’s take another look at the currents in three fluorescent lamps, each of which is now connected to one of the three phase conductors of a three-phase system. We shall distinguish between two cases: in the first, the lamps are driven by magnetic ballasts, in the second, they are driven by old-style electronic ballasts. As already mentioned, nobody actually builds ballasts of this type anymore, but they are of interest to us because two of these older type ballasts are equivalent to a PC and its monitor, or to two or three TV sets, or to between five and fifteen compact energy saver lamps – all devices that still use this type of technology. When the lamps are driven by magnetic ballasts, the currents in the three phase conductors all have very large inductive components – which would, of course, be compensated for in most industrial or commercial applications – but they are all approximately sinusoidal in form. If you sum the three currents over time, you see that they pretty much cancel each other out. The resultant current in the neutral conductor has an approximately triangular profile with an amplitude of about 35% of the current flowing in any one of the phase conductors and with the typical frequency of 150 hertz. The situation is quite different when the lamps are driven by the older electronic ballasts. The currents now have very small conduction angles, which means that for most of the time no current flows at all. The reason, if you recall, is that the reservoir capacitor can only recharge when the rectified line voltage is greater than the residual voltage across the capacitor. As there is clearly no overlap between these peaks there can be no mutual cancellation between them. The three pairs of current peaks on the phase conductors all appear in undiminished form as six current pulses on the neutral conductor. Once again we are not dealing with theoretical possibilities here, but with real phenomena observable in present-day power systems.
What we’ve just observed is open to analysis. If we break down the three identical currents in the phase conductors into their Fourier components, we will of course get three fundamental modes with the same amplitude and phase angle which cancel each other at the neutral point. However, for the third harmonic, with its frequency of 150 hertz, the phase shift of 120 degrees has the same effect that a phase shift of 360 degrees would have for the fundamental mode. The three harmonics present on the three phase conductors are thus all exactly in phase with one another and sum to form a neutral current with three times the peak amplitude, three times the mean average and three times the rms value. The same situation will arise for all other harmonics whose order is divisible by three – the so-called triple-N harmonics – but the largest distortion is due to the third harmonic and only this harmonic can, when tripled in the manner shown, yield neutral currents that exceed the overall rms currents in the phase conductor.
Harmonic effects of this type can influence the behaviour of other loads. As an example let’s consider what is normally regarded as a very robust device – the three-phase asynchronous induction motor. If the motor was fed only by the third harmonic, it would be the same as connecting all three line connectors to a single phase conductor. If the motor has a delta winding or if the neutral point is not connected, nothing will happen … all is quiet. But if the neutral point is connected, some particularly unpleasant humming results, followed a little while later by a rather unpleasant smell – but no torque. This is what is known as a zero phase-sequence system. In contrast, the seventh order harmonic – a positive phase-sequence system – is trying to drive the motor forward, but at seven times the rated speed. That this doesn’t work is due to the fact that the fundamental voltage component still dominates and forces the motor to rotate at the rated speed. From the point of view of the seventh harmonic, it looks as if the harmonic is trying to feed a motor that is running at only one seventh of the synchronous speed. To the seventh-order voltage component, the motor is approximately stationary and the harmonic is continuously attempting to supply current to a motor that is effectively at rest! The situation is different again with the fifth harmonic. Unlike the seventh order mode, the fifth harmonic is attempting to drive the motor at “only” five times the synchronous speed. Unlike the seventh-order harmonic, it’s also trying to turn the motor backwards. This sort of situation is known as a “negative phase-sequence system”. The fifth voltage harmonic is therefore continuously feeding a motor that is running backwards! Because the associated current is predominantly inductive, the impedance is higher at higher frequencies. However, the fifth harmonic current also gives rise to a permanent and, because of the opposite sense of rotation, larger starting current that results in increased copper losses in both the stator and rotor and a reduction in motor torque. If under these conditions start-up fails and if no motor circuit-breaker has been fitted, then the motor has pretty much had it. If, on the other hand, the motor does actually start, then the “only” problems will be the shortening of its service life due to the unacceptably high temperatures generated and the extra cost for the additional heat loss.
We now know that a transformer, and in particular its leakage reactance, can influence the behaviour of non-linear loads. In the upper figure we have a relatively clean power system and a transformer that is powering a smallish load – a compact energy saver lamp. The lamp has a power rating of 11 W and – so far – an apparent power input of 20 VA. However, if the same transformer is made to carry its rated current by powering a load that consists purely of lamps of this type, the power supply characteristics will change as shown in the lower figure. While there is less relative distortion of the current profile and the apparent power drops to 15 VA, the distortion of the line voltage is substantially larger. Even though the harmonic spectrum of the current is attenuated, it still has an effect on the transformer! Unfortunately, this experiment is rather complicated to set up in the laboratory whilst being practically impossible to perform in the field. For this reason, the measurements shown below were made using only one energy saver lamp and the resistance and the leakage reactance of the transformer windings were simulated. This nevertheless produces the same conditions as those when, for example, a 15 kVA transformer powers 1000 of such 15-VA lamps (and nothing else). Extrapolating from the single-lamp simulation to the 1000-lamp configuration then simply involves multiplying the measured current values by a thousand, i.e. reading “ampere” for “milliampere” and (inversely in this case) “microhenry” for “millihenry”. In the simulation, the transformer carried its rated current and should therefore exhibit the rated temperature rise. However in reality, the eddy-current losses that arise in components made of ferromagnetic materials and, most importantly, in the conductors themselves, grow with the square of the current and with the square of the frequency. This quadratic dependence makes sense when one considers the fact that the stray magnetic fields of the load current induce what one might call an “eddy voltage” that drives the eddy currents perpendicular to the actual current direction. As this “eddy voltage” is proportional to the rate of change of current, so too is the (ohmic) eddy current – the eddy-current power loss, which is computed as voltage times current, is therefore proportional to the square of the rate of current change.
This is how we learned to quantify a transformer’s losses: The steel loss is constant, and the copper loss increases by the square of the load current. Now it is justified to assume that the input voltage equals the rating and that the steel or no-load losses therefore are constant. But a transformer’s current rating always refers to ohmic load and rated frequency. To be precise, the formula then looks as shown here, for if the load current includes any higher frequencies, the supplementary losses increase by the square of the current and by the square of the frequency!
Therefore let’s now try and quantify this effect by taking a look at a simplified example. Nothing could be easier. After all the harmonisation document HD 428 provides two very plain and simple formulae for this purpose. The only catch with these equations is that you’ve got to enter numbers that are simply not normally available. In such a situation it’s probably best to calculate a made-up extreme example, since there may hardly be any worse polluters around than such 1000 lamp load, and then use the results to compute an adequate safety factor for practical applications: Typically, about 5 to 10% of load losses in a transformer are due to eddy-current losses – assuming a 50 hertz sinusoidal current. Here in our worst case scenario, the losses are 10%. The table lists the voltages and currents associated with the various component harmonics as measured at the output of the transformer. The table can be understood as shown here: With our 1000-lamp “rated load”, the fundamental frequency of the load current generates eddy-current losses that are 5.6% of the copper loss we have under the rated operating conditions. The third and fifth harmonics, though smaller in magnitude than the fundamental, generate eddy-current losses that are 29.5% and 24.5% respectively of the copper losses. If one adds all these eddy-current losses up to the 51st harmonic, it becomes apparent that when powering this particular load, the eddy-current losses are not 10% but 81.4% of the copper losses. Thus, if one measures the true rms rated current, the additional power loss in the transformer due to eddy currents is eight times greater than the value specified for a 50 hertz operation.
If we had simply just extrapolated from one compact fluorescent lamp, with the transformer’s influence being negligible, to the case with 1000 lamps, we would have calculated the influence of the load upon the transformer without taking the influence of the transformer upon the load into account. Then the “supplementary additional losses” would have turned out nearly 9 times as high as they really are!
Auch hierfür gibt es ein Werkzeug: Den K Factor Calculator des britischen Kupferinstituts. Laden Sie sich das Programm von www.cda.org.uk herunter.
Given the quadratic dependence of the copper losses including eddy-current losses on the current, the load current has to be reduced by 35% or a transformer selected that is 35% larger if overheating is to be avoided. As the example calculation was a worst-case analysis, this figure of 35% will certainly be sufficient to cover all practical situations. This of course does not touch any additional reserves that may need to be taken into account for any other reasons. It scarcely ever has any disadvantages; rather, as a rule, a more advantageous point of operation is reached, since a transformer, depending on design, always has its best efficiency between 25 and 50% of rated load. A formula can be a wonderful thing, but if the harmonic spectrum of the load has to be known before the formula can be used, it’s not terribly helpful – no matter how accurate the formula itself may be.
Power plant generators usually have quite high stray impedances that lead to short circuit voltages of around 40%, which in turn result in the voltage induced in the winding being 40% higher than that measured across the generator terminals at rated load. The internal voltage drop is high enough to offset this and, accordingly, the short circuit current is relatively low – basically a welcome advantage. However: During load variances the induced voltage has to be adapted quickly (by varying the excitation current) but this is impossible because the inductance of the exciter winding is tremendously high. A significant change of excitation current takes several seconds. Voltage dips during sudden increases and overvoltages during sudden decreases of load current are thereby inevitable. Now non-linear loads are such whose impedances are not constant but which change quite substantially at least two times each cycle! Consequently, the terminal voltage is no longer sinusoidal, however clean may be the internal induced voltage, also called electro-motoric force EMF by physicists. Neither of these consequences affects a common electricity system, since a large number of generators are always working in parallel and their short circuit powers add up – more or less, depending on distances. But what happens when the feeding of a system converts undetected from the public grid to one single emergency diesel generator? The inner impedance of the generator is several times greater than that of the grid, and all of a sudden you receive tremendous distortions of the voltage, although its TRMS value matches the rating. A bicycle dynamo may serve as an extreme example of this, here a 28-pole hub dynamo that runs at the speed of the front wheel. The usual 8-pole machines provide an output frequency of about 250 Hertz at a travelling speed of 25 kilometres per hour, but the electrical properties, other than the unacceptable mechanical friction losses, are largely the same: The no-load voltage is several times the rated voltage even at a moderate speed. The frequency and along with it the inner reactance both increase proportionally with speed, and therefore the current remains largely constant above a certain speed. In physicists’ terminology the dynamo is not a voltage source but a current source. Accounting for by lack of precision in the design and production of such machines, the no-load voltage is anything but sinusoidal. Yet, under short circuit conditions or with ohmic load the current becomes a clean sine wave almost without any distortion. With ohmic loads, the same applies also for the voltage, which still looked quite bad even before any load was applied. The effect is the same when feeding a bridge rectifier in sequence with an accumulator (charging the accumulator): The voltage shows up as an alternated image of the accumulator’s DC, but the current sine wave is retained – just the other way round than we usually expect and desire from usual voltage supply sources. The higher the inner impedance of a source, especially the inductive share, the more of this usually undesired characteristic any power supply will adopt. This may cause quite a number of strange effects, for instance additional zero crossings. Electronic control equipment, e. g. power electronic regulators, that use the zero crossings as a trigger signal, may react in an unpredictable way.
Alternatively, we can take a look at a capacitor – a robust and simple component. At high frequencies, the impedance is, as expected, lower and the capacitor allows correspondingly more current to flow. Higher-frequency voltage components superimposed on the mains voltage, that are not discernible on the voltage curves, become evident in the current profiles. Here you can see what happened when an 11-watt fluorescent lamp was driven by a magnetic ballast in an office building. The first thing that’s apparent is that the ballast was not particularly good, as the total active power is over 17 W. But that is not what we want to focus on here. The second is the extreme magnitude of fundamental reactive power that should have been compensated for in a commercial building. On analysis, the following problem is revealed: The current that was previously nearly sinusoidal in character now contains a large number of higher frequency components. As the only change has been the introduction of the compensation capacitor in parallel, this must be the cause and the capacitor must be conducting these higher frequency currents. This would also account for the fact that complete compensation has not been achieved; despite a correctly dimensioned capacitor, almost 10 VAr of wattless power remain. A further indication of the role played by the capacitor is the renewed rise in the reactive power to 23 VAr without any other change to the circuit having taken place. All that happened in this case was that the converter-fed lift within the building started up. Evidence for this is the 1-volt drop in the voltage and the even greater distortion now evident in the total current waveform. This distortion and the sudden increase in reactive power can only be explained by the superimposition of higher frequency components on the system voltage. Whilst too small to be seen in the voltage profiles shown on the display, they nevertheless drive high-frequency currents through a capacitor that is directly connected to the power system. Consequently the capacitor has to handle a far greater load than it would if connected to a pure 50-hertz voltage. As the additional current flows through nothing but a capacitor, it is obvious that we are dealing here with reactive current, that is, with the distortion-related reactive power mentioned earlier. Nowadays such effects must be taken into account when dimensioning capacitors and reactive-power compensation equipment. In contrast to the current trend, configurations involving series lead-lag compensation are preferable.
A number of these problems can be shown on the DKI’s demonstration panel which we regularly display at trade shows and exhibitions. In the case shown here, we can connect either three incandescent light bulbs with or without dimmers or three compact energy saver lamps to the three phase wires of the power system. The currents flowing in each of the four conductors (L1, L2, L3 and N, the neutral conductor) are measured by a moving-iron instrument and by a moving-coil meter with rectification. In addition, the neutral current flows through a loudspeaker.
In conventional fluorescent lamps, the currents in the phase wires are nearly sinusoidal; the difference between the arithmetic mean displayed by the moving-coil meters and the rms values displayed by the moving-iron instruments is slight. With only 1 load in operation nothing much unexpected happens. The loudspeaker produces a low dull 50 hertz hum. Turning the second speaker on leaves the sound unchanged. The curve of the return conductor current changes, but its rms magnitude hardly does. If all three lamps are on there is practically no current in the neutral wire. All one hears now from the loudspeaker is a quiet 150-hertz tone; the 50 hertz components have cancelled each other out because the three loads are balanced, that is, they are distributed symmetrically.
However, the situation looks (and sounds) very different when the three compact energy saver lamps are on. Already with just 1 of them in operation, the arithmetic mean values shown by the moving-coil instruments are now only about half the rms values displayed on the moving-iron meters. Turning on the second lamp adds substantially to the return conductor current, which now significantly exceeds the amplitude of the individual lamp’s current. With 3 lamps in operation, the neutral current nearly reaches its theoretical maximum of root 3 times the phase conductor current, despite the fact that the loads are balanced and the 50-hertz fundamentals still cancel one another out, with the loudspeaker humming loudly in protest.
Conditions such as those just described mean that it is now very common to find working currents present in the structural metalwork of buildings fitted with TN-C, TN-C-S and “unclean” TN-S systems, as well as in systems with badly designed or ill-configured earthing. A typical example was a house built by a company in 1962 as a luxury residence for one of its managers. Originally power was supplied by the company. Later the house had to be connected to the public electricity supply. A telecom subscriber connection and an equipotentail bonding busbar were installed. This current was found flowing from the telecom subscriber connection to the equipotential bonding busbar. Pay attention to the continuously varying rms value and the continuously varying current profile! Where this current comes from and where it’s flowing to is unclear. Equally unclear is the damage that it could do within the phone company’s network or anywhere else for that matter.
Although it used to be perfectly adequate for many purposes, the TN-C system is no longer able to meet modern power quality needs. On the one hand, increasingly large working currents are now found on the return conductor and this same wire also has to carry any leakage and fault currents. On the other hand, more and more pieces of electrical equipment are being installed which require a definite signal reference voltage to be provided by the protective conductor. A voltage drop of only 1 volt between two earthed points caused by the presence of permanent working currents in the earthing system can seriously inhibit data flow. And although the maximum permissible fault voltage of 50 V may never have killed a person, it has certainly finished off a fair number of network interface cards. That’s why today the return currents and the fault currents should use separate paths. In practice that means that if the transformer is in the building, a TN-S system is required from the transformer onwards. If not, a TN-S system is needed at the building services entrance point. A TN-S system from the transformer onwards would be the best solution, as trials have shown. But it’s hardly realistic to expect the power supply company to come and replace the earth cable. In new installations, however, consideration should certainly be given to using a four-core cable whose screen can be used as the PE conductor.
In some cases, it may be expedient to completely suppress the third harmonic in the neutral wire by means of a parallel resonant circuit that is tuned to reject 150 hertz oscillations. A device of this type is made in Finland, and can be supplied in four sizes for current ratings of 25, 63, 125 and 160 A. For higher power applications, several of these filter units can be connected in parallel. However, the rated current will only ever be reached if one of the phase lines were to fail. The device is also available as the “THX” model in Germany. In this variant, the capacitance is inductively coupled. The leakage inductance is so high in the booster transformer that the transformer also functions as a substitute choke. The benefit of the THX model is that the voltage level for the capacitors is freely selectable and that one can therefore use readily available standard components instead of having to employ the special version for 120-volt continuous duty and 250-volt short-time operation if the Finnish THF is used.
The filter response can be seen on this graph. At 50 hertz the impedance is very low. We don’t therefore expect any problems with the shift in the neutral point if one of the phase lines were to fail, or with the tripping conditions in the event of a short circuit occurring. Nevertheless, these issues should be checked by computation before using these filters in practice.
Though useful, these filters are certainly not an excuse to go back to using neutral conductors with a lower cross-sectional area. The filter is equipped with a bypass disconnector and can therefore be shorted if measurements need to be made or if problems arise, as was the case at a TV transmitter in England. The improvement in the current profile is at the expense of a deterioration in the voltage profile within the installation – as can be seen from these measurements taken in an office building. In the absence of the filter, harmonic distortion totals 2%. If the filter is now connected, the THD in the power supply falls to 1.3%, but on the installation side, the third current harmonic “piles-up” or accumulates in front of the filter which leads to major distortion of the voltage which has now leapt up to 10.8%! Notice that the voltage profile has become much more rectangular in form. One might reasonably expect that the supply company would buy the filter and install it and that the customers would be up in arms to prevent it. But that’s not what you find in practice. In most cases, the filter is sold to those installing and operating the electrical equipment. That’s because these devices are not sold to improve voltage quality, but to reduce losses. And loss reduction is something they do very well, by significantly reducing the loading in the neutral conductor and by avoiding circulating currents from the zero sequence system in the high-voltage delta winding of the transformer. In one case, the oil temperature of a distribution transformer could be reduced from 100 to 70 degrees Celsius. The example just mentioned which corresponds to surroundings similar to those at the television transmitter in England is not an optimum environment in which to use the THF. The neutral current is only reduced from 96 amps to about 50 amps, depending on when one makes the measurement and at the price of a substantial increase in voltage distortion. Here is another example. A nursery in Finland that has plant lamps with a total wattage of 500 kW supplied by a remarkably clean power distribution system with a voltage distortion level of only 0.3%. The phase currents in the not perfectly balanced loads were between 900 and 1100 A. The current in the neutral conductor was 866 amps at 150 hertz and was reduced to 112 amps at 50 hertz after the filter was switched on. The main portion of this residual current of 112 A must be due to the remaining load imbalance as the fundamental frequency is now 50 hertz and the waveform almost sinusoidal. The 150 hertz component has practically disappeared and the total harmonic distortion within the installation has only risen to a relatively harmless 3.2%.
And now let’s turn to the problem of stray alternating magnetic fields in TN-C systems which arise when one can no longer control where the neutral currents are flowing. The larger the distance between the phase and the return conductors, the greater of course the space into which these fields can propagate. In a clean TN-S system the forward and return currents all flow within the same cable and so stray fields do not arise. In TN-C and TN-C-S systems, by contrast, one may have to cope with flickering monitors and other effects caused by alternating magnetic fields. Occasionally, disturbances due to magnetic fields will also arise in TN-S systems. For example, when a transformer station is located in the basement immediately below an office. However, the transformer may not always be the source of the problem and one should bear in mind that the stray fields generated by transformers are frequently overestimated. In many cases, the problem stems from the use of single-core rather than five-core low-voltage cable. The Swiss have developed an EMC transformer for just this sort of situation. The transformer differs from a conventional distribution transformer in that the bushings are not arranged in a row on the lid but in a rectangular pattern near the base of the unit. The result is a reduction in the magnetic flux density of about 60% when measured 2 metres above the transformer’s lid. In an otherwise businesslike, reasonable advert the company came up with the intriguing pronouncement: “When we build transformers, the first thing we focus on is complying with our customers’ requirements and only then on complying with standards. Why? Because it is customer needs and not standards that offer real scope for product innovation.” Unfortunately, there are all too many electrical engineers and building installation planners who simply reject customer preferences or requirements with a withering “ … and where does it say that?” response. This “by the book” approach is not actually very good for business …or would you go to a car dealer offering cars that are only equipped to comply with what’s laid down in the motor vehicle traffic regulations? I suspect not.
In a TN-C system one almost inevitably ends up with a mixture of data currents and working currents – a situation in which faults are almost pre-programmed! Indeed one German company specifically draws its customers’ attention to this very problem. But why? After all, they manufacture high-frequency interference suppression devices and they are not going to be much good at solving the problem of mixed currents. The problem is that unexpected currents in the earthing system can render the interference suppression equipment itself effectively useless. Uncompensated currents (in a cable for example) can cause magnetic saturation in the ferrite cores of the suppressors leaving them unable to function correctly. Perhaps, all too often, the real source of the problem went unrecognized, leaving the company to face the wrath of angry customers convinced they were experiencing an HF problem. To put things straight, the company presumably decided to let customers know that the root of the problem may well lie in the power distribution system rather than in their products. One of the factors contributing to this problem is that the shielding on the data transmission lines, which are earthed at both ends, also helps to carry the return currents in the PEN conductor. Indeed it has been reported that in extreme cases, the shielding itself can become warm – what a cynic might call a real “hotline” problem. These people are not the only ones to have recognized the problem. Others include the property insurers who have to bear the costs of any resulting damage, and the technical experts who compile weighty technical reports along with an invoice. Nevertheless, preventive measures taken during the planning phase and aimed at avoiding faults and preventing damage do not cost a great deal. Retrofitting, on the other hand, proves to be a very expensive undertaking. The copper industry has estimated that the cost ratio is about 1 to 10. Though this doesn’t of course include the fees demanded by our friends the inspectors and surveyors!
For decades the received wisdom has been that electrochemically induced corrosion is only possible with a d. c. current source. However, over the last few years, there has been a noticeable rise in the number of voices claiming that this is not perhaps totally the case. Take a look at this particular example. What you see is a steel strip that served for perhaps ten years as part of the earth electrode in the TN-C-S system at Paderborn-Lippstadt airport. The strip shown was located only a few metres away from a transforming station. The side that faced the transformer has become razor sharp as a result of material removal; the side facing away from the transformer appears much rounder and is essentially unchanged.
One would not imagine that d. c. current had flowed in this case but the station’s outgoing cable, as measured across all three phase conductors and the PEN conductor, indicated a shortfall of some 25 amps which instead of flowing to earth through the earthing system must have flowed back to the neutral point via the transformer tank. A clean TN-S system has now been installed in which the neutral point is no longer earthed by being connected to the tank. So far none of the previous damage including corrosion of the earth electrodes has reappeared.
With lightning protection of this type you shouldn’t have a lot to worry about. But before the new lightning conductors were fitted, every time one of the frequent lightning storms struck between ten and twenty runway lights would blow – an expensive business given the special materials used in these lamps. Just as working currents have to be kept away from protective earth and equipotential bonding systems, so too must interference currents, in particular lightning stroke currents, be kept well clear of active conductors. In a TN-C system it’s pretty easy for lightning to cause problems because when it strikes the building it is usually only a short way to the next piece of electrical equipment. In a TN-S system on the other hand the route is considerably longer, leading back initially to the central earthing point, which is the only connection between the neutral and the protective earth wires, and then back again. In that time the lightning current will have lost much of its power, unless of course it was some massive freak strike that would anyway have broken down the insulation and caused a flashover.
The TT system, just like the introduction of the PEN conductor, was an invention stemming from times when copper was short. Unfortunately, today either or both systems have established themselves as viable options. This will remain so as long as, when reading the readers’ inquiries in the trade journals, all questions touching this area centre around the question whether that short piece of wire from the star point to the connection of the N and PE conductors shall be marked blue or green-yellow. Note that physics doesn’t care about colour codes. The real problem about the TT system is that it comes in a blurred multitude of variants. It may look like this … …or like this … …or like this … …or like this. The more earthing points, the better it is, of course, that’s right!
But before you really become aware, you have created one of these dangerous mergers which a colleague in a standardisation committee erratically called a TNT system – sure a Freudian error of course but as such with a lot of truth! in it! Imagine the earthed structures were storage buildings for instance for tri-nitro-toluene! Now if someone who is not certain about which configuration was selected at the feeding side adds a connection at the consumer side that should not be there, the whole thing may become hazardous!
Protective earth, operational earth, functional earth, your earth, my earth … so how many earths are there? There is of course only one earth and only one reference potential. Unless of course you manage to earth one part of the installation, to “Mars” a second, to “Saturn” a third and so on.
Just what do we mean when we talk about a TN-S system? It’s fairly common both in journals and in reality to come across a system that looks like or claims to be a TN-S system but turns out to be just something like a TN-S system in name alone. As soon as there is more than one connection between the N and PE conductors, the effect of the TN-S system is largely wiped out. Let’s assume that the prerequisite mentioned here is fulfilled and that left-hand transformer is on and is driving a single-phase load between L1 and N. The return current flows left through the neutral conductor to the transformer’s neutral point, but that’s not quite the whole story. Part of the neutral current flows towards the right to the neutral point of the inactive transformer and from there via the PE conductor to the neutral point of the feeding transformer! One possible solution in the present case could be the use of four-pole switches, but this only does it in cases like this where simultaneous feeding of several sources is undesired and avoided. Another one would be to disconnect the PE wire from the neutral point on one side, but this might result in the impedance between the conductors being too great to meet the required tripping conditions. A better solution would be to disconnect both connections and to create a new one at the load centre. Now both of the transformers could equally feed simultaneously, and the TN-S system would still be real.
Now some experts with an elaborated sense of precision say a multiply fed system could never ever be a pure TN-S system but at best a TN-C-S system, since the interconnection of two star points was always to be seen as a PEN conductor. Regardless of this, however, the IEC standard shown here claims: If, from any point of the installation, the neutral and protective functions are provided by separate conductors, it is not permitted to connect the neutral conductor to any other earthed part of the installation. Since PE and PEN conductors are earthed conductors by definition, there shall be only one connection between the N and PE conductors within one installation. Hardly any statement in the standards is older, and hardly any statement is violated more often. It is a tradition to equip each and every new switchgear cubicle, even if supplied only partly prewired, at least with a bridge between N and PE. Any well intended TN-S system is sabotaged this way. This leaves two possibilities: 1. The system is IEC compliant, and the little bridgelet between the N and PE conductors is therefore the only one of its kind within the plant. All lightning and earth fault currents (that are usually given in kilo-amperes!) have to use it. 2. The bridge is not the only one within the system. Then about half of the return current flows through it, and this may be 10 amps, 100 amps or 1,000 amps, depending on the system’s size and the types of loads it supplies!
So, that there is no “bang” is really a miracle.
In future, service cabinets will (hopefully!) all look like this one at Paderborn Airport. There is a neutral bar close to the phase conductor bars, to limit magnetic stray fields, and a protective earth bar remote from the others, in order to reduce inductive coupling, and the two are completely separate with no connection between them. What you see is a thick single-core cable, with a blue identifier strip on one end, that is attached to the neutral bar and a separate thick single-core cable with yellow-and-green colour coding at one end which is attached to the PE bar.
The equipotential bonding bar is located right next to the service cabinet – and if you look carefully, what do you see? A thick single-core cable with a blue identifier strip at one end and a thick single-core cable with yellow-and-green colour coding at one end! Both cables are earthed once and once only at this point.
And exactly that was the configuration laid down in this European standard and published in the draft version. Unfortunately, when the final version came out in September 2001, it contained an older but incorrect drawing!
Luckily a number of people in the standardization committees noticed the error fairly quickly – though not quickly enough, as the version containing the erroneous drawing had already been published. Although an erratum was published and distributed soon afterwards, there is no doubt that there are nevertheless quite a few people out there who have the version with the wrong drawing in it and not the correction. And the moral of the story is …? Well, the more engineers and technicians there are to keep a watchful eye on the compilation of a new standard the better. It’s no good complaining about errors, omissions and inconsistencies when it’s all over – far better to get involved right from the start to make sure that any mistakes or inaccuracies get ironed out early on in the compilation process.
Whilst some truths may well be eternal, others have a very definite and in some cases rather short half-life! In a highly respected electrotechnical trade journal, one could read the following ancient truth: “The PEN conductor plays a dual role in TN-C systems. Its primary role is as a protective conductor, its secondary function is that of a neutral or return wire.” Now this statement is certainly still valid. Protection against electric shock and fire remains of course the number one priority. What is worrying is what one learns by reading between the lines … The passage just quoted suggests that a PEN conductor is something perfectly normal, ubiquitous and completely ok, and that the return currents flowing along it are not worth mentioning. Statements like these – which may well have been true years ago – are no longer a fair reflection of current practice. From today’s point of view, the description of a PEN conductor quoted above should have been reformulated or supplemented by additional remarks to prevent the reader from gaining a false impression and a false sense of security.
At this point, the anxious question is usually raised: “But where do I buy five-core cable? I’ve never seen any offered by our normal supplier!” This authentic photograph is proof that five-core cable is commercially available. For more information we recommend contacting International Cablemakers Federation. It would probably be quite instructive if cable manufacturers were to calculate how much they really do save by using antiquated three-and-a-half core cable. Not only is a single PEN conductor used, but the cross-section of the PEN wire is smaller too and it’s circular, unlike the segment-shaped phase conductors. Given that these design differences complicate production logistics and raise storage costs, it would be interesting to learn just how much the manufacturers save by using this cable. And how much extra insulant is needed to fill the gaps created within the cable? Yes, copper is expensive, but trying to save money by economizing on cables often backfires making it necessary to carry out costly retrofitting work.
Some company logos are logical indeed, as seen here at a booth at Hanover Fair 2003, while it is only quite recently that one company even appears to emphasize the importance of the 5th conductor in their logo.
Standards are not always the best way to solve technical problems. Occasionally one’s left with the feeling that there is more politics than science in some standards. To compile a technical standard you need a host of engineers willing and able to draft a document that reflects real physical conditions, to keep it up-to-date and, wherever possible, to phrase it in a way that is as simple yet clear as possible. That’s a mammoth task, but one that is certainly accomplishable. Gradually, the standardization authorities are realizing that four wires are simply not enough for distributing three-phase alternating current in modern commercial or industrial buildings. For example, the version of this European standard now states that: “ …it shall be considered that a PEN conductor through which the unbalanced currents and the accumulating of harmonic currents and other disturbances are transmitted cannot provide an appropriate earthing. It shall also be considered that the TT and IT mains systems need more corrective measures in particular against overvoltage; …”. Well it may not be perfect, but it’s certainly a nod in the right direction. Unfortunately the standard then continues: “ … there should be no PEN within the building …” and “ … wherever possible, the TN-S system should be used.” Fine as far as it goes, but it does raise the question whether a word like “should” should really be used in a standard.
Oder sehen Sie in die VDE 0100 Teil 444 von Oktober 1999. Dort heißt es: »In Gebäuden, die in bedeutendem Umfang Betriebsmittel der Informationstechnik aufweisen oder von denen dies für die Zukunft zu erwarten ist, muss ab dem Gebäudeeintritt der Stromversorgung die Anwendung des TN-S-Systems, d.h. die Verlegung getrennter Schutzleiter (PE) und Neutralleiter (N),… –« Sehen Sie, nun haben wir endlich das »muss« drin! Ja, was muss sie denn nun? In Betracht gezogen werden muss sie! Ist das nicht schön? Jawohl, das ist nicht schön! Wenn Sie also nach dem TN-C-System installieren und nachher vor einem unzufriedenen Kunden oder schlimmstenfalls vor dem Kadi stehen, müssen Sie nur nachweisen, wie auch immer Sie das anstellen wollen, dass Sie die Anwendung des TN-S-Systems in Betracht gezogen haben! Mit viel Glück stimmt Ihnen das den Richter gnädig; den Kunden aber garantiert nicht.
Initial enthusiasm was great among utility people when in March 2000 a standard fixing the quality of power supply was finally available. Fixing? Well, let’s wait and see. First of all it says there: “The voltage magnitude on the low voltage level is 230 V plus/minus 10%, measured between the phase and neutral conductor in 4-conductor systems and between phase conductors in 3-conductor systems.” This at least is a clear statement, whenever 5-conductor systems appear not to exist. But beyond there are only statements to be found like: “Under normal operating conditions the number of voltage dips lies between some tens and several thousands per year. They usually last less than 1 second and have a retained voltage of over 40%.” “Under normal operating conditions rapid voltage changes usually remain below 5% of the rated voltage, but deviations of up to 10% may under certain circumstances occur several times a day.” What is to be seen as “normal operating conditions” and when “certain circumstances” are given, is unfortunately not specified. “Short interruptions of up to 3 minutes occur some tens up to several hundred times a year. Up to 70% of these may last for less than 1 second.” What about the other 30%? Are they allowed to do whatever they like? “Switching transients usually do not exceed 1.5 kV, surges commonly stay below 6 kV. In individual cases, however, they may be higher than that.” How satisfying! Unbalance: “95% of all 10-minute mean values must have an inverse system of less than 2% the direct system. But where there are many single- and two-phase loads in operation, it may as well give rise up to 3%.” And what about the rest of the 10-minute mean values? What happens within the 10 minutes? If the inverse system rises to 20% for one out of every 10 minutes, then this is in order? – it does not in fact make any difference, since where by nature unbalance occurs, the limit is simply adapted to be higher.
“The frequency should be between 49.5 Hertz and 50.5 Hertz for at least 99.5% of a given year.” Let us compare these “limits” to what happens in the real existing UCTE mains when in Spain, on account of a failure, 900 MW vanish unforecast from the network. This graph was published in “etz” in 1998. It does require some high-end metering equipment to begin to identify the effects, since the continental grid is so stable. Instantaneously the frequency dropped from 50.005 Hertz to 49.960 Hertz, overshot to 50.015 Hertz after another 2 seconds and back again to 49.965 Hertz before stabilizing at 49.975 Hertz. So when do the limits of 49.5 Hertz and 50.5 Hertz ever become relevant? Islands you mean? – No, island networks not running synchronous to the UCTE mains are exempted. So this does not only refer to the Shetlands but already starts with the British isles, since the British system is connected to the UCTE network via HV-AC link and therefore does not run synchronous.
Indeed, when you browse the internet for a map of the UCTE grid, you find that the British Isles are not part of it. But let us have a look at a “real” island – here it comes. It is so small that we have to move it and offset it in a different colour to be able to pick it out. Malta has a population of 395,000 people and a totally isolated electricity supply system of 547 MW.
Still, the frequency did not vary any more than from 49.8 hertz to 50.13 hertz over several hours. A general remark adds with reference to all of these statements: “These limits refer to normal operating conditions only, not to fault conditions.” So does the obligation for supply also drop out when power drops out?
Subsequently this German standard states: “The features given in DIN EN 50160 represent extreme situations but do not describe the usual situation in the mains. For planning electrical installations with a normal usage it is sufficient to consider the most likely situation in the mains at the corresponding point of common coupling.”
Or take a look at the 2001 edition of this standard. Here we are told that: “For buildings which have, or are likely to have, significant information technology equipment installed, …” – which pretty much covers every building nowadays – “consideration shall be given to the use of separate protective conductors (PE) and neutral conductors (N) …”. Wonderful isn’t it? Well, no actually it isn’t. It means that if you install a TN-C system and later find yourself facing a dissatisfied customer or, worse still, being hauled up in front of the courts, all you have to do is demonstrate (though quite how is not clear) that you did indeed consider installing a TN-S system. With any luck the courts will be lenient with you, though the customer is likely to see things rather differently! Fortunately, the 2002 edition now emphasises doing something rather than just thinking about doing it. Unfortunately, the word “shall” has been replaced with “should”.
It is however perfectly possible to formulate the provisions clearly and correctly. For example, as of September 2000, paragraph 6.3 of this European standard reads: “The AC distribution system inside a building shall conform to the requirements of the TN-S system. This requires that there shall be no PEN conductor inside the building, …” No room for ambiguity there! One should perhaps mention however that by including this passage the authors of the EN 50310 standard are essentially making life easier for themselves. After all, EN 50310 is the European variant of the VDE regulation 0800 on “Information Technology Installations in Buildings”. The telecommunications engineers have clearly realized that many of their faults are power-system based and by formulating the standard in this way they don’t then have to worry about planning, implementing or paying for remedial measures.
Another approach attempts to solve the problem at source, by focusing on the individual devices that generate current harmonics. Whilst this approach is certainly very commendable, it also proves to be the least economical way forward, because concentrating on lots of small units, means that material and labour costs will always exceed those from implementing a central solution. We can illustrate what we mean by looking at the electricity supply network in Germany. The German grid is composed of huge power stations each generating over one gigawatt of electrical power. However, this structure is not the result of a nation having caved in to the demands of power industry lobbyists, nor is it there to help fuel the arguments of certain political groups, it is quite simply the cheapest solution available in terms of both construction and operating costs. Despite the long power transmission distances demanded by this sort of network, it is still more economical than a more decentralized system consisting of a larger number of smaller generators. And the same applies to harmonics in power networks – it makes more sense both economically and ecologically to seek a central solution to the problem. This European standard has, nevertheless, come into force. So it’s worth taking a look at the approach that they’ve adopted, for example, with respect to so-called class D equipment. Fortunately, the rather weak and vague formulation “with a special wave shape” that was still being proposed in 2000 as a definition of Class D devices has now been abandoned. We now have a situation in which, for example, a personal computer is included in this category not because of the presence of some particular current profile but by definition: personal computers are Class D products. But let’s take a look first at the requisite test conditions.
The voltage used for the test measurements must meet very tight tolerances – and for good reason. The 2% tolerance on the rms value appears rather generous and indeed it is, but this only reflects the fact that the rms value does not have a great effect on the spectrum of current harmonics. On the other hand, the upper limit of 0.4% that specifies the maximum permissible fifth harmonic content will be difficult to comply with. These limits must be tight because the harmonic content of the voltage has an extremely strong influence on that of the current, as the following simple experiment demonstrates. The input current of a small power pack for a laptop computer was recorded. For the purposes of the measurement the power pack drove a small constant resistive load. The rated output power was 3 A multiplied by 15 V, the input power was thus less than the threshold of 75 W above which this standard would apply. The power pack was not therefore supplied with any integrated harmonic suppression circuits. The mains voltage already had a THD of 4.3%, as you see in the third picture at the top. As we’ve already seen, the harmonic content of voltages in power systems that drive single-phase rectifier loads is pretty much the same irrespective of whether one measures between a phase and a neutral conductor or between two phase conductors. If one compares the dominant fifth harmonics in these two cases, one finds that they are shifted relative to one another by 180 degrees, i.e. they have opposite polarity. As the distortion of the phase-to-neutral voltage (i.e. the one with the flat-top waveform) has the effect of reducing harmonic currents emissions, one would expect that reversing the polarity of the fifth voltage harmonic would result in an increase in the absolute magnitudes of the component current harmonics. And this is exactly what one observes in practice. To check this, we use a transformer to generate a 230-volt equivalent of the 400-volt phase-to-phase voltage and feed this into the same test circuit, as shown in the middle of this slide. If we now subtract the fifth harmonic content from one another in these two cases – which is the correct thing to do as they are of opposite polarity – we arrive at a difference of 8% between the fifth harmonics of the two voltages. With the following consequences: The active power is the same, but the total rms current increases from 321 mA to 545 mA. The peak current trebles from 1 ampere to 2.5 A. Compare the current curves of the second pictures in the top and middle rows. The reactive and apparent power also increase accordingly. Clearly the effect the voltage profile has on the current waveform is immense. This is why the tolerances for the test voltage have to be so tight. If they weren’t, the values measured when using a typical mains voltage would give a highly falsified picture. Now it might be possible to create a voltage that, whilst not quite conforming to the standard, would nevertheless be clean enough for the purposes of some initial pre-testing. The idea here is to try and achieve a compromise between the two waveforms discussed above and we do this by inserting a choke between the transformer and the power pack. The fifth current harmonic flowing in the transformer creates a voltage drop at the choke with the opposite polarity to the fifth voltage harmonic coming from the mains. If the choke inductance is adjusted to match the load, the two voltages should cancel one another out. And this is indeed what is observed! The fifth harmonic component in the voltage across the load has now dropped from 8.1 V to 0.8 V, as you can see comparing the measurements in the third pictures of the middle and bottom rows, respectively. Unfortunately the third-order current harmonic – the largest of the current harmonics – now generates a substantial voltage drop across the choke. In addition, there is also a resonance phenomenon probably between the choke and a suppression capacitor in the power pack, as becomes evident in the first measurement screen at the bottom of the slide. Instead of falling, the THD of the voltage across the load has now risen to 5.8% in the third bottom screen measurement. Oh well, it was a nice idea while it lasted. So there’s no getting round it, we need to use a special clean test voltage source and that’s not going to come cheap, because as we’ve emphasized the tolerances for the test voltage allowed by the EN 61000-3-2 standard are very tight.
Unfortunately, the same cannot be said of the harmonic emission levels that the standard stipulates for the equipment under test. As currently specified, these limits really can’t be said to harm anyone – but they also can’t be said to do much good either. As a result, there are no fewer than four simple, almost trivial methods that one can use to modify a computer workstation so that it conforms to the provisions of the 61000-3-2 standard. For Class D equipment, the EN 61000-3-2 standard stipulates limits on harmonic emissions relative to the device’s nominal power. For example, in the case of the third current harmonic, a Class D unit must draw no more than 3.4 milliamps per watt of input power.
The input power of a Pentium 166 PC was measured to be 116 W. According to the new standard, this means that the third current harmonic must not exceed 395 mA. The value measured, however, was 411 mA – at least that was the value recorded on this power system with its particular internal impedance value. The lower the internal impedance of the mains, the less susceptible the line voltage will be to distortion by the load current. Unfortunately, a lower internal impedance also means increased emission of current harmonics from the load! Be that as it may, in the present case we have a fairly old computer that was built well before the new standard came into force and in which the third harmonic current exceeds – though not dramatically – the maximum acceptable value specified by the standard. As already mentioned four different methods are available to bring the computer back into line with the standard. One way is to quote a nominal power that is 4% higher than the measured input power. The standard actually allows a tolerance of 5%. But what is actually meant by the “nominal power” of a personal computer? You won’t find it quoted on the rating plate because it doesn’t make much sense to specify a nominal power input for a piece of equipment that is frequently modified by adding extra components at some later stage. That’s the reason why the power supply units in PCs are always massively over dimensioned. The nominal power of the supply unit is of course always quoted; in this case it’s about 220 W. But if we took this value as our yardstick, we wouldn’t need to worry about compliance with the standard at all. And if the power supply unit is tested alone and at full load, then the measured values are not going to be a very useful guide to the behaviour expected when driving only a fraction of the full load – which is nearly always the situation we’re dealing with. Alternatively we can view the PC workstation not as a single unit but as a collection of separate devices, which is after all what it is. Remember the application range of Class D. If we do that, we find that the monitor draws about 60 W, the computer unit about 40 W and the remainder is adsorbed by the peripheral equipment. The standard, however, applies only to Class D products with an input power of between 75 and 600 W, so in this case none of our component devices would be affected by the new standard! A third solution would be to connect an ohmic resistance in series with the load and thus artificially degrade the power system. “Whack in a light bulb, problem solved!”, as a cynic once put it.
Probably more by accident than design, an anonymous humorist has come up with a fourth solution to our problem, namely to connect a resistive load in parallel rather than in series. The standard states that a Class D device may emit a maximum third harmonic current of 3.4 milliamps per watt of fundamental power into the mains. With a 50-watt switched-mode power supply load in parallel with a 1000 watt ohmic load there shouldn’t be too much of a problem keeping 3rd harmonic current emissions below 3.4 amps.
But non-linear loads are not restricted to switched-mode power supplies and there are a large number of other devices that will also cause some radical reshaping of the current profile. One such device is the conventional lighting dimmer. If you connect three 60-watt incandescent light bulbs with phase-control dimmers to the three phase conductors of a three-phase power distribution system, and you set the dimmers to full power, you will effectively have three identical balanced ohmic loads. As a result, the currents in the neutral conductor will cancel each other out and the combined neutral current will be zero. But as soon as you dim the lights by, for example, reducing each of the conduction angles by, say, 60 degrees, the lamp currents, though still identical, will no longer be ohmic. And in that case, the three phase currents no longer sum to zero along the neutral conductor. Reducing the conduction angle means that a portion of the phase current is “cut off” and it is just these cut-off portions that then appear on the neutral conductor with a fundamental frequency of 150 hertz and a whole range of harmonic components. Now it’s clear that one is rarely going to have a situation in which all three dimmers are set to the same level. If, as would be usual, each is set to a different level, then the three currents are no longer equal neither in terms of their rms values, nor in terms of their profiles or harmonic spectra. As a result, the neutral conductor now has a 50 hertz component.
A further, though often overrated, way of deforming current waveforms is to exploit the magnetizing current in transformers, since the no-load current is severely distorted, having a highly non-sinusoidal form. Yet, the influence of such distortions on overall power quality is slight. Whilst the no-load current in a small transformer may reach 30% of the rated current, this ratio decreases markedly with increasing transformer size, and in a good distribution transformer the no-load current will be less than 1% of the rated current. The measurement shown here was taken during excitation from the 420 V low voltage side, so the no-load current is in the range of 0.1% of the rated current! Since the no-load power loss is within the same range, the transformer picks up hardly any magnetizing current at all. Now those who claim that transformers cause a worsening of the mains power factor by introducing inductive reactive power are quite correct, though the deterioration in the power factor is due to the leakage reactance of the windings and has nothing whatsoever to do with the core. That’s why the leakage reactances are perfectly linear, why the reactive currents or inductive voltage drops are sinusoidal and why power factor correction is simple to achieve by using conventional capacitor banks – if only these are controlled, for with the mains running without load the stray reactive currents would be zero.
Another means of loading and possibly overloading the neutral conductor is to feed d. c. currents into the three-phase a. c. system. Doesn’t happen? Well, don’t be so sure! The following method was proposed in the German electrotechnical magazine ETZ as a means of connecting a greater number of small d. c. motors in a weaving mill to the power system. According to the article, the benefit of the method lay in the fact that the applied d. c. voltage of “root 2” times 230 V was more suitable for running small motors than the “root 2” times 400 V that arises in the normal B6 circuit. That’s quite correct. The article then went on to argue that the return d. c. currents on the positive bridge and those on the negative bridge will cancel. That too is quite right, if one assumes that both motors are switched on and loaded at the same time. Whether a situation like that will ever actually arise in practice remains rather doubtful. In addition the circuit is unlikely to look like the one shown in ETZ. It’s more likely that each converter has its own connection to the return conductor and that between them there a countless meters of conductor cable. It is in this intermediate section – and only in this section – that a compensating current will flow in the neutral conductor. If of course a PEN rather than an N conductor was used, then this current will flow in the earthing system and in all other components that are earthed via the PEN conductor. The currents from the phase wires once again sum to give a d. c. current equal to the cube root times the rms current flowing in each of the phase conductors. Nevertheless, it is highly unlikely that this will be the cause of a fire. It’s far more probable that the d. c. current will start to cause problems long before the installation reaches its thermal rating. So for those who still believe that you need a d. c. current for electrochemically induced corrosion, here you have it – d. c. current in an a. c. system. As we have seen, installing a TN-S system enables you to avoid a great many disadvantages including that of having d. c. current in your earthing system. The one disadvantage with the TN-S system is that the earthing system can no longer help to conduct return currents, thereby easing the load on the PEN conductor. In fact this ability to conduct return currents is now presented as an advantage of the TN-C system – which it certainly is. Nevertheless, it does nothing to ameliorate the many disadvantages inherent in the TN-C system. And if a full-cross-section neutral conductor is fitted in a TN-S system, then there is really very little risk of overloading the neutral. In extreme cases, a reinforced or a double neutral conductor can be fitted. However, the d. c. current still causes an unsymmetrical voltage drop in the power system so that there is a small d. c. voltage component superimposed on the mains voltage. Doesn’t matter? Well, that depends …
If you run a toroidal-core transformer on a more or less “normal” power system you will be aware of the favourable properties this type of transformer offers. The no-load power loss and the no-load current are both very low, and the magnetizing reactive power is almost zero. In the transformer shown here, which is rated at 200 volt-amps, the no-load current is a mere 10 milliamps while the no-load losses are below 2 W. And things stay like that until you decide to switch on a conventional hair dryer at half-power. Typically, the half-power mode involves inserting a diode in front of the heater and fan so that the dryer is driven by the half-wave rectified voltage. The result is the slight asymmetry in the mains voltage mentioned above. It arises because the drop in the mains voltage is perhaps half a volt or at the most one volt greater on the one halfwave than on the other. The effect of this minimal asymmetry on a toroidal-core transformer in no-load mode is however enormous. The no-load current is now 1.26 A with the peak current on one side almost 6 amps! The no-load power rises to an astonishing 36 W and leaves no further scope for loading the transformer as the transformer is too hot even under these no-load conditions. If you now switch the hairdryer to full power, everything is fine again and the no-load current in the toroidal-core transformer falls back to its normal level. If one looks carefully at what is going on, this phenomenon is not perhaps as astonishing as it seems at first. With a no-load current in the toroidal core of 10 milliamps, we can assume that magnetic saturation of the core will set in at current levels slightly above the peak value at around say 15 milliamps. As the ohmic resistance of the primary winding is about 20 , all we need is a d. c. voltage of 400 mV to generate a current of 20 milliamps in the primary coil which saturates the core.
At least this standard has managed to put an end to this approach. As you see here, such awkward power consumers count among the few that produce harmonics of even orders. One-way rectification is a typical example of this. Here you see that this hair dryer already produces 1.46 amps of second harmonic current, while the new limit is 1.05 A. Again, the required degree of reduction is not terrific, but here it leads to a basic change of technology: The power is now no longer halved by means of a semiconductor, but new hair dryers switch off half of the heating wires, since this happens to be about as cheap. So in this special case the new limit effects a 100% reduction, even though the standard would have been satisfied with 28%, but that’s just pure fluke.
Can one drive a standard small transformer, especially one with a toroidal core, on a conventional phase-control dimmer? Based on what we’ve just seen, you’d probably say it’s better not to try. If the dimmer exhibits even the smallest degree of imbalance, the no-load losses in the transformer would increase dramatically – right? Well … no actually! Operating the transformer with a dimmer cannot be compared to the imbalance arising from ohmic voltage drops in the mains. After all, in the former example the transformer was permanently connected to the mains and exposed to come what may from there. Now, however, we have a semiconductor switch in between, which is somehow switched by the transformer. An extreme example should help to make the point: A conventional small transformer typically has a no-load current of about 54 milliamps and a power loss under no-load conditions of about 1 watt. If we connect the transformer to a variable a. c. voltage supply and insert a diode between the a. c. supply and the transformer, then the transformer will be fed only the rectified half-wave. What is remarkable here is that one has to turn up the voltage to 64 V, that is to over 25% of the rated voltage, before the no-load current reaches its normal value. And the voltage has to be increased to 80 V before the no-load power attains its normal level! The transformer is being fed by a d. c. voltage – albeit a pulsating one – with an amplitude of nearly 30% of the rated voltage. Right? … Wrong! The current in the primary winding is a pulsating direct current, which the diode will only block when the current in the conducting direction stops flowing. However, due to inductance this current continues to flow for quite some time after the voltage has already changed polarity. As a result the current flows for certain periods counter to the applied mains voltage. And of course that’s all that reactive power is, in this case the reactive power due to magnetization. So the diode only blocks – and thus separates the transformer – when the voltage of opposite polarity has “slowed” the current to zero. The larger the current, the longer it takes for it to reach zero. However, during this time there is already a voltage at the transformer terminals which does not correspond to the polarity of the current – the inductance has therefore almost symmetrized the voltage at the terminals. So what have we learned? A small transformer can be safely driven by a dimmer even if the dimmer is significantly asymmetrical. Nevertheless, when dimming inductive loads there are still a few aspects that we need to address.
Right at the beginning, we stated that all electronic circuits need a smoothed d. c. voltage for operation. But as they say, there is no rule without an exception. An electronic transformer for halogen lamps functions by first rectifying the input a. c. voltage and then rapidly chopping the rectified voltage to yield an a. c. voltage of much higher frequency which can then be transformed by a transformer that is much smaller than would be needed if transforming a 50-hertz voltage. An electronic transformer has no need for a reservoir capacitor, so, unlike the compact energy-saver lamp, it can’t be accused of distorting the current drawn from the power distribution system. The transformer simply stops working when the line voltage passes through zero, but that’s not a problem as a halogen lamp is an incandescent lamp and can therefore cope with an interruption to the supply voltage irrespective of whether driven directly by a 50 hertz supply or by a frequency of 50 hertz modulated on the transformer’s high-frequency output, which can be seen in the middle of the bottom row of this slide. The fact that electronic transformers have a high-frequency output should be taken into account when selecting a suitable transformer for driving halogen lamps. A standard that limits the maximum length of the cables to be used, but says nothing about the distance between cables is not particularly well suited to avoiding interference with other electronic devices. Though to be honest, most DIY enthusiasts who set about installing a set of suspended halogen spots are not generally aware of the existence of any relevant standards or, if they are, are usually quite happy to ignore them. The problem is made worse by the square waveform of the output current, which in theory comprises an infinite spectrum of harmonics and in practice means that disturbances can be caused by harmonic components well up in the megahertz range.
It also seems that the authors of the relevant standards and the manufacturers of electronic transformers for halogen lamps have forgotten to include the inductance in their calculations. Why? Well, take a look at these lamps with their elaborately wound connectors designed for a cable-mounted system. The inductance of these connectors is so high that it effectively swamps that of the cables. Now at 50 hertz that doesn’t have any effect but for an electronic transformer with an output frequency of say 40 kilohertz one shouldn’t be too surprised when the voltage across the actual lamp terminals is somewhat less than the voltage across the cables. Examples like this help to give you an impression of the extent to which alternating magnetic fields have on electrical equipment and – as some believe – on people too. Now the makers of electronic transformers claim that the output voltage of the transformer is controlled electronically. But where? At the output terminals or at the last lamp in the row? It makes all the difference! The voltage is of course controlled at the terminals, but just what voltage is left for the last lamp no one can say – especially when the lamps are kitted out with these fancy spiral connectors. The voltage drop across a conventional transformer, that is the difference between partial-load and full-load conditions, can be effectively minimized by appropriate design and anyway falls off the larger the transformer gets. In summary, electronic transformers for halogen lamps do make sense when weight, size or a specific shape is important, and in such cases they have undisputed benefits. In general, however, you’re likely to have fewer problems by sticking to conventional transformers made of copper and steel.
Most of the phenomena that have been presented can be readily understood by building a model building. Old printed circuit boards, bolts from the local DIY store, original wiring accessories, car lamps as loads and all fed by a 12-volt “distribution transformer station”. With these components you can now put together a wonderful TN-C system – warts and all! In the model, the PEN conductor at the “transformer station” is “earthed” by connecting it to the protective conductor on the mains plug. The model building is also equipped with a “foundation earth electrode”. Now 10.6 A flow in the phase wire of this single-phase model, but only 9.6 A return via the PEN conductor. So where’s the rest gone? Aha! Gone to ground obviously!
And so stray electric currents are flowing through the building’s structural metalwork – just what we don’t want!
Now here’s another method we can use to stop harmonics getting into the power distribution system. In the spirit of the European standard EN 61000-3-2, one could connect a series resonant circuit in front of each switched-mode power supply and tune it to filter out everything other than 50 hertz signals. This would force the devices to accept only sinusoidal current. Now it wouldn’t be possible to do this with every type of device, but it certainly works with the classic contaminators: Computers, television sets and compact fluorescent lamps. Consider an inductive and a capacitive reactance in series. At resonance, the inductive reactance (shown here in blue) and the capacitive reactance (in yellow) are of equal magnitude and, because the phase angle between them is 180 degrees, the resulting reactance (shown in red) is zero. Thus at 50 hertz, the only factor contributing to the overall impedance is the ohmic resistance of the reactor winding.
Tests were conducted using various combinations of inductors and capacitors. The inductors were conventional magnetic ballasts for fluorescent lamps and the capacitors were selected to match the inductance. As you can see, the results are, in principle, always the same: the power loss at the ballast’s rated current is relatively small. If a short-circuit occurs in the load, the voltage across the capacitor will not increase to dangerous levels because at high currents the core material in the inductor will become magnetically saturated, the inductance will drop, the resonant frequency will no longer match the line frequency, and this will automatically limit any further rise in the short-circuit current. The short-circuit current in a filter with a rated current of 0.67 ampere does not go above 1.7 amps.
The efficacy of filtering is very clear. The current has readopted an approximately sinusoidal waveform and the peak value has fallen from 4 A to 1 A.
When any new regulation is issued, it’s usually accompanied by a lot of moaning by those who’ll have to bear the costs or who think or claim they’ll have to bear the costs. In the case of the EN 61000-3-2 standard, the manufacturers kept up their resistance until they were sure the final standard was ineffectual. The possibility of achieving “conformity” by exploiting the compliance tolerances written into the standard has not been used. Instead, the decision was made to introduce half of the filter just described in order to eliminate the remaining few percent of harmonic emission. Placing this small inductor in series with the power pack on a typical mains supply with a THDU value of 3.1% is sufficient to lower the THDI value, i.e. the total harmonic current distortion of the input current, from 78.5% to 65.8% which is enough to ensure compliance with the standard At the same time as having the effect of making the computer inductive or at least a little more inductive than it was previously. Considered alone (i.e. without an added inductor), a switched-mode power supply is a slightly capacitive load, so the compensating effect of a series inductance is to be welcomed. However, if the EN 61000-3-2 standard is tightened up in future – which is certainly going to be necessary if the standard is to have any real impact – then a larger inductor is going to be needed. And a larger inductor means that a capacitor will also have to be added. If a compensation capacitor is not included, the voltage drop across the inductor will be too great, even at the 50 hertz fundamental frequency, and the voltage across the reservoir capacitor will simply collapse. The new monitor that came with our new PC was a flat panel monitor. The input power of these new devices is only about 20 W and that fact alone represents real progress in monitor technology. At these low power ratings, flat panel monitors obviously don’t need to concern themselves with the specifications in the 61000-3-2 standard, though the manual proudly informed us that the monitor was indeed compliant. Well it certainly sounds reassuring even though the idea of complying with a standard that isn’t actually relevant is a little strange. It’s rather like sticking a label on a shoe box that reads “Contents comply with food laws” – possibly true, but not terribly helpful! And the next PC, which was about ten times faster than the previous one, actually consumes less power. Only when it’s being driven hard does the power input exceed that of the older device – but this is where the inductor kicks in and lowers the THDI. If the power pack ever had to supply power at its rated power output of 300 W, the absolute level of harmonic emission would of course be greater, but the relative THDI would be less. The power pack was subjected to these conditions in the lab during testing but these are not conditions that are ever likely to be experienced in practice.
As mentioned earlier, a solution that is based on modifying each individual device is not the most cost-effective approach. A better solution would be one in which the various loads are allowed to emit their harmonic spectra and the low-voltage power system would be designed so that it can cope with these current components. Given that in its present form the EN 61000-3-2 standard does little to prevent harmonic emission into the mains, the need for a line-side solution remains. Professor Manfred Fender of the Wiesbaden University of Applied Sciences has come up with some interesting suggestions. According to Professor Fender, every other single-phase load that emits harmonics should be converted into a two-phase load. These two-phase loads would not draw current from the mains at the same time as the single-phase loads, but 30 degrees before and 30 degrees after. As a result, the current peaks would be more broadly distributed across each halfwave. The only problem with this idea is its practical implementation as it would require three-phase a. c. power sockets in our offices and every second computer would have to be designed to handle 400 V.
A more feasible idea proposed by Professor Fender is that we should no longer restrict ourselves to using only distribution transformers of the Dyn5 vector group, which introduce an angular displacement of 150 degrees between the input and output voltages. Fender suggests that every second transformer should be wound as either a Dzn6 or Yzn6 vector group, which are characterized by a shift of 180 degrees between the input and output sides. Once again, the result would be to shift half the current peaks on the medium-voltage side by 30 degrees thereby distributing the current flow more evenly and restoring a much more sinusoidal current waveform. An additional aspect of this arrangement is that for a zero phase-sequence system such as the third, ninth and fifteenth harmonics, a Dyn5 transformer still exhibits a short-circuit voltage of around 60% of the rated value, making this type of transformer relatively impermeable to these harmonic components. As a consequence, these currents tend to “pile up” or accumulate in front of the blockage leading to the well-known voltage distortions. In contrast, transformers belonging to the Dzn6 and Yzn6 vector groups exhibit a short-circuit voltage that is only between 5 and 10% of the rated value, that is the value for three-phase current. The triple-N harmonics can therefore pass through almost unobstructed, apart from the fact the voltage drop in the transformer, which determines its short-circuit voltage, is predominantly inductive and therefore rises with increasing frequency. Clearly the output voltages from transformers belonging to different vector groups can no longer be connected in parallel. The statement “every second transformer” shouldn’t therefore be taken literally. What is meant is that within a single installation or building – anywhere in fact where transformers could theoretically be connected in parallel – one should use transformers that are in the same vector group. A different vector group should then be selected in the next installation. Varying the vector groups in this way is sufficient to mitigate the distortion in the medium-voltage supply. The pictures we saw earlier of the grossly distorted mains voltage in the Frankfurt business district would then be a thing of the past as would the propagation of the distortion from one low-voltage supply to the next.
If none of these solutions proves feasible, it’s normally sufficient to filter out those harmonics with the largest amplitudes. These are typically the third, fifth, seventh and, possibly, the ninth-order harmonics and the filters used are suitably dimensioned series resonant circuits, or acceptor circuits as they’re sometimes known – and strictly speaking a subject in their own right. In principle, these circuits are nothing more than inductor-capacitor units whose resonant frequency can be tuned to accept current at the individual harmonic frequencies. Previously, this sort of circuit was consciously avoided as a tuned acceptor circuit effectively represents a short-circuit at its resonant frequency. If currents at the resonant frequency are present, they will, of course, prefer to take the path of least resistance irrespective of whether these currents were generated in the installation being protected or whether the filter circuit is “accepting” currents of this frequency from outside. Of course, attracting external currents would not occur if all electrical installations were cleansed of their harmonic currents. The problem is that whoever starts cleaning up their own system ends up cleaning up everyone else’s as well. In order to prevent any overloading, the filter has to be dimensioned large enough to cope with any pollution from the neighbours. Quite apart from the costs involved and the irritation of cleaning up somebody else’s mess, it could also be that you end up paying for the imported reactive power as if it were reactive power stemming from your installation. The reason is that the meter cannot distinguish between the two. Harmonics generate reactive power and reactive power is not associated with any specific direction. That’s why one normally avoids tuning inductive VAR compensators to match specific harmonic frequencies. Although desirable from an engineering viewpoint, the problem with this solution is how to set the ball rolling. A start could be made by using this fairly low-price model in a relatively well isolated power distribution system. The idea involves tuning the individual filter circuits in a conventional inductor-capacitor compensation unit of the sort normally used to compensate for the reactive power generated by the fundamental current mode to the different harmonic frequencies. At full load, all harmonics, from the third to the ninth, are absorbed by the filters. If the need for compensation falls at lower loads, then the filter for the highest order harmonic is switched off. This process is continued as necessary until only the filter for the third order current harmonic is active. Such an approach seems plausible because when driving partial loads it seems reasonable to suppose that the absolute level of harmonic emission will be lower and filtering the higher-order harmonics is less important. The base load and the third harmonic are always filtered and it’s astonishing to see just how much that can help. But in order to delve deeper into this technique please refer to our other presentations.
Let’s now summarize the key things we’ve learned so far: We need to construct low-voltage power systems in which the harmonic currents can be left to their own devices. These power systems must not use cables which have neutral or protective earth conductors of reduced-cross-section. In certain cases, a PE conductor with a smaller cross-sectional area may be acceptable, but the neutral conductor should, if anything be reinforced, or even, in the most extreme cases, be configured as a double-neutral. The neutral and the protective earth conductors must be separate wires and must only be connected to one another at one point and one point only, even with multiple incoming feeder lines! Quite generally, the cross-sectional area of a conductor must be generously dimensioned. The larger the cross-section, the smaller the voltage drop and the lower the effect that distorted currents can have on the system voltage and thus on energy loss. A further advantage of using conductors whose cross-sections are big enough to handle more than just present loads is that they provide a useful safety buffer for coping with any future increase in demand. When using meters and measuring instruments, it is essential to pay attention to true root mean square values (TRMS). However, this doesn’t necessarily mean that the rms value is always the decisive one. For example, when assessing chemical processes such as the level of charge in a battery or when measuring the magnetic forces acting in a motor, it’s the mean current magnitude and not the rms value that is relevant. But when assessing the thermal loading capacity of a conductor, the crucial quantity is indeed the rms value! And all would be well in the medium-voltage power systems if a variety of distribution transformers belonging to different vector groups were used. If necessary one could also filter out the first few harmonics using suitably tuned acceptor circuits. But that, as we mentioned earlier, is a rather different kettle of fish.
And what has the Leonardo logo been good for all of the time? It wants to point out that our project has been supported by the European Union for three and a half years. But not only this. In December 2004 it was awarded as one of the best 3 projects of the year. Please see this on the background that the EU supports about 4000 projects annually! So we really think we are at least worth a click!
Transcript
1.
New Loads on
Old Systems
Stefan Fassbinder
DKI
German Copper Institute
Am Bonneshof 5
D-40474 Düsseldorf
Tel.: +49 211 4796-323
Fax: +49 211 4796-310
sfassbinder@kupferinstitut.de
www.kupferinstitut.de
2.
The German Copper Institute,
DKI, is the central information
and advisory service dealing
with all uses of copper and its
alloysWe offer our services to:
Commercial companies
The skilled trades
Industry
Research institutes
Universities
Artists and craftsmen
Students
General Public
We can be contacted by:
post
phone
fax
e-mail
internet
online database, or
personally
3.
Standards and Power Quality
– a major problem in public
and commercial buildings
The substantial growth in the number of electronic devices
being used in recent years has resulted in a significant
change in the types of load being driven by today’s power
distribution systems.
Because these devices are equipped with rectifiers and
smoothing capacitors, the current drawn from the power
system is significantly distorted from the sinusoidal
waveform provided by the utility companies.
This has serious consequences for power quality...
4.
Power quality
problems
Turn off the mixer
love, the monitor’s
flickering again!
are
usually of
terrestrial
origin.
5.
The good
old days:
Ideal
three-phase
system voltage
Three balanced
ohmic-inductive
single-phase
loads on the three
phase mains
-150A
-100A
-50A
0A
50A
100A
150A
0ms 5ms 10ms 15ms 20ms
t
i
i1 (L1)
i1 (L2)
i1 (L3)
-350V
-250V
-150V
-50V
50V
150V
250V
350V
0ms 5ms 10ms 15ms 20ms
t
u
u1 (L1)
u1 (L2)
u1 (L3 )
6.
0V
50V
100V
150V
200V
250V
300V
350V
0ms 5ms 10ms 15ms 20ms
t
u
0A
1A
2A
3A
i
0V
50V
100V
150V
200V
250V
300V
350V
0ms 5ms 10ms 15ms 20ms
t
u
0A
1A
2A
3A
i
0V
50V
100V
150V
200V
250V
300V
350V
0ms 5ms 10ms 15ms 20ms
t
u
0A
1A
2A
3A
i
gleichgerichtete
Netzspannung
Kondensator-
spannung
gleichgerichteter
Netzstrom
The situation
today:
Computed current and voltage profiles when a
230 V, 58 W fluorescent lamp with an old-style
electronic ballast is driven on the phase wire of
a typical power distribution system...
System voltage: 230 V
System frequency: 50 Hz
Source resistance: 500 mΩ
Longitudinal induction: 904 µH
System impedance: 575 mΩ
Mean d.c. current: 180 mA
Smoothing capacitance: 220 µF
7.
And what about electronic ballasts that are rated over
25W? Introduce electronic power factor correction (PFC)
0V
20V
40V
60V
80V
100V
120V
140V
160V
180V
200V
220V
0° 15° 30° 45°ϕ
u
0,0A
0,3A
0,6A
0,9A
1,2A
i
8.
Though today’s ballasts are
better if rated >25 W
Only the
small
ones
continue
to pollute,
just like
PCs, TV
sets, ...
do
9.
This leads to a number of
previously unknown effects:
1.Deformed voltage curves
2.Huge inrush current peaks
3.Instrument dependent measured parameters
4.Increased loading of phase conductors
5.Loading and overloading of the neutral conductor
6.Overheating and start-up problems of induction motors
7.Additional stray losses in transformers
8.Repercussions of generators upon the mains
9.Influence of capacitors, repercussions upon the mains
... and four additional complications in TN-C systems:
10.Stray currents in equipotential bonding system: magnetic fields
11. Contamination of data streams by stray currents
12.Stray currents in the earthing system: corrosion damage
13.Lightning currents in devices and equipment
10.
1. Deformed voltage curves
in theory ...
-350V
-300V
-250V
-200V
-150V
-100V
-50V
0V
50V
100V
150V
200V
250V
300V
350V
0 5 10 15 20
t / ms
u
System voltage profile when driving
a single-phase load comprising
twenty 58 W energy-saver lamps in
parallel on a typical 230 V power
system
11.
... and in practice ...
An actual
current profile
and
a voltage profile
recorded in a large
storage, dispatch and
administrative building
12.
... at home or in industrial
environments
Television, 40 W
200 V/div., 2 A/div.,
5 ms/div.
Harmonic
spectrum
Frequency
converter
50 A/div.
Phase currents:
I1=I2=I3=16.0 A
IN=29.5 A
5 ms/div.
13.
Every periodic waveform can be written
as the infinite sum of sinusoidal waves ...
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Sum of squares: ΣI² = 9855 mA²
Root of sum = RMS value: I = 99 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
i3
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Sum of squares: ΣI² = 9977 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
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-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
i3
i5
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Sum of squares: ΣI² = 9993 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
i3
i5
i7
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Cont. 7 th harmonic: I = -2 mA I² = 4 mA²
Sum of squares: ΣI² = 9997 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Cont. 7 th harmonic: I = -2 mA I² = 4 mA²
Cont. 9 th harmonic: I = 1 mA I² = 2 mA²
Sum of squares: ΣI² = 9998 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Cont. 7 th harmonic: I = -2 mA I² = 4 mA²
Cont. 9 th harmonic: I = 1 mA I² = 2 mA²
Cont. 11 th harmonic: I = -1 mA I² = 1 mA²
Sum of squares: ΣI² = 9999 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11 i13
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Cont. 7 th harmonic: I = -2 mA I² = 4 mA²
Cont. 9 th harmonic: I = 1 mA I² = 2 mA²
Cont. 11 th harmonic: I = -1 mA I² = 1 mA²
Cont. 13 th harmonic: I = 1 mA I² = 0 mA²
Sum of squares: ΣI² = 9999 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11 i13
i15
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Cont. 7 th harmonic: I = -2 mA I² = 4 mA²
Cont. 9 th harmonic: I = 1 mA I² = 2 mA²
Cont. 11 th harmonic: I = -1 mA I² = 1 mA²
Cont. 13 th harmonic: I = 1 mA I² = 0 mA²
Cont. 15 th harmonic: I = 0 mA I² = 0 mA²
Sum of squares: ΣI² = 10000 mA²
Root of sum = RMS value: I = 100 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11 i13
i15 i17
Synthesis of a triangular current profile 100mA
Content fundamental: I = 99 mA I² = 9855 mA²
Cont. 3 rd harmonic: I = -11 mA I² = 122 mA²
Cont. 5 th harmonic: I = 4 mA I² = 16 mA²
Cont. 7 th harmonic: I = -2 mA I² = 4 mA²
Cont. 9 th harmonic: I = 1 mA I² = 2 mA²
Cont. 11 th harmonic: I = -1 mA I² = 1 mA²
Cont. 13 th harmonic: I = 1 mA I² = 0 mA²
Cont. 15 th harmonic: I = 0 mA I² = 0 mA²
Cont. 17 th harmonic: I = 0 mA I² = 0 mA²
Sum of squares: ΣI² = 10000 mA²
Root of sum = RMS value: I = 100 mA
14.
... whose frequencies, called harmonics, are
integer multiples of the fundamental
frequency.
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Sum of squares: ΣI² = 8106 mA²
Root of sum = RMS value: I = 90 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
i3
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Sum of squares: ΣI² = 9006 mA²
Root of sum = RMS value: I = 95 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
i3
i5
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Sum of squares: ΣI² = 9331 mA²
Root of sum = RMS value: I = 97 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot
i1
i3
i5
i7
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Cont. 7 th harmonic: I = 13 mA I² = 165 mA²
Sum of squares: ΣI² = 9496 mA²
Root of sum = RMS value: I = 97 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Cont. 7 th harmonic: I = 13 mA I² = 165 mA²
Cont. 9 th harmonic: I = 10 mA I² = 100 mA²
Sum of squares: ΣI² = 9596 mA²
Root of sum = RMS value: I = 98 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Cont. 7 th harmonic: I = 13 mA I² = 165 mA²
Cont. 9 th harmonic: I = 10 mA I² = 100 mA²
Cont. 11 th harmonic: I = 8 mA I² = 67 mA²
Sum of squares: ΣI² = 9663 mA²
Root of sum = RMS value: I = 98 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11 i13
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Cont. 7 th harmonic: I = 13 mA I² = 165 mA²
Cont. 9 th harmonic: I = 10 mA I² = 100 mA²
Cont. 11 th harmonic: I = 8 mA I² = 67 mA²
Cont. 13 th harmonic: I = 7 mA I² = 48 mA²
Sum of squares: ΣI² = 9711 mA²
Root of sum = RMS value: I = 99 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11 i13
i15
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Cont. 7 th harmonic: I = 13 mA I² = 165 mA²
Cont. 9 th harmonic: I = 10 mA I² = 100 mA²
Cont. 11 th harmonic: I = 8 mA I² = 67 mA²
Cont. 13 th harmonic: I = 7 mA I² = 48 mA²
Cont. 15 th harmonic: I = 6 mA I² = 36 mA²
Sum of squares: ΣI² = 9747 mA²
Root of sum = RMS value: I = 99 mA
-175
-150
-125
-100
-75
-50
-25
0
25
50
75
100
125
150
175
0 5 10 15 20
t / ms
i/mA
itot i1
i3 i5
i7 i9
i11 i13
i15 i17
Synthesis of a rectangular current profile 100mA
Content fundamental: I = 90 mA I² = 8106 mA²
Cont. 3 rd harmonic: I = 30 mA I² = 901 mA²
Cont. 5 th harmonic: I = 18 mA I² = 324 mA²
Cont. 7 th harmonic: I = 13 mA I² = 165 mA²
Cont. 9 th harmonic: I = 10 mA I² = 100 mA²
Cont. 11 th harmonic: I = 8 mA I² = 67 mA²
Cont. 13 th harmonic: I = 7 mA I² = 48 mA²
Cont. 15 th harmonic: I = 6 mA I² = 36 mA²
Cont. 17 th harmonic: I = 5 mA I² = 28 mA²
Sum of squares: ΣI² = 9775 mA²
Root of sum = RMS value: I = 99 mA
15.
Even far better simulations are
available free of charge at:
www.powerstandards.com/McEachern
16.
-4A
-3A
-2A
-1A
0A
1A
2A
3A
4A
0ms 5ms 10ms 15ms 20ms
t
i
-4A
-3A
-2A
-1A
0A
1A
2A
3A
4A
0ms 5ms 10ms 15ms 20ms
t
i
Applying the Fourier
analysis to a real PC
current does not work
directly ...
... but it can be
modelled by a similar
triangular curve in
which not all of the
higher harmonics
have been included
Triangular current profile of
the same amplitude and a
pulse duty cycle of 1:7
Input current of a PC
with monitor
18.
M
3
Harmonics
Energy
(active
power)
L1
L2
L3
N
PE
Important:
Harmonics are created within the loads
themselves and can flow “upstream” to
contaminate the power system!
19.
-6A
-5A
-4A
-3A
-2A
-1A
0A
1A
2A
3A
4A
5A
6A
0ms 5ms 10ms 15ms 20ms
t
i
Squared values
Without
Fundamental: 502 mA 251986 mA² the fundamental
3rd harmonic: -479 mA 229000 mA² 229000 mA²
5th harmonic : 434 mA 188595 mA² 188595 mA²
7th harmonic : -374 mA 139903 mA² 139903 mA²
9th harmonic : 304 mA 92527 mA² 92527 mA²
11th harmonic : -232 mA 53671 mA² 53671 mA²
13th harmonic : 163 mA 26600 mA² 26600 mA²
15th harmonic : -104 mA 10780 mA² 10780 mA²
17th harmonic : 57 mA 3299 mA² 3299 mA²
Sum of the squares: 996362 mA² 744377 mA²
Root thereof: 998 mAeff THD = 863 mA
What actually is THD?
Example shown here: triangular current
profile with an r.m.s. current of 1000 mA
and a pulse duty cycle of 1:7
THDr (root mean square) = 863mA/1000mA = 86 %
THDf (fundamental) = 863mA/502mA = 172 %
20.
Effect on phase-to-neutral
and phase-to-phase voltages
Recorded on
June 30,
2002,
2:30 p. m.
What was
the matter
then?
The soccer
final:
Germany vs.
Brazil!
21.
The triple-N harmonics
drive a circulating current in
the delta winding of a
distribution transformer ...
... but the
voltage harmonics
propagate into the next low-
voltage power distribution
system!
22.
229V
230V
230V
231V
231V
232V
232V
233V
233V
234V
234V
Freitag,
23.8.020:00
Samstag,
24.8.020:00
Sonntag,
25.8.020:00
Montag,
26.8.020:00
Dienstag,
27.8.020:00
Mittwoch,
28.8.020:00
Donnerstag,
29.8.020:00
Freitag,
30.8.020:00
t
U
0%
1%
2%
3%
4%
5%
THDU
U(RMS)
THDr
Harmonics in the course of a week
25.
0V
50V
100V
150V
200V
250V
300V
350V
400V
450V
500V
550V
0 10 20 30 40 50
t / ms
u
0A
10A
20A
30A
40A
50A
60A
70A
80A
90A
100A
110A
120A
130A
140A
150A
i
Switching on just as supply voltage passes through zero
0V
50V
100V
150V
200V
250V
300V
350V
400V
450V
500V
550V
0 10 20 30 40 50
t / ms
u
0A
10A
20A
30A
40A
50A
60A
70A
80A
90A
100A
110A
120A
130A
140A
150A
i
Switching on just as supply voltage is at its peak
2. Huge inrush current peaks
– in theory ...
26.
... and in practice
Inrush current of a
compact energy-
saver lamp and its
effect on the system
voltage (recorded at
the Technical
University of
Budapest)
27.
3. Measured parameters are
instrument dependent
The root mean square value of an alternating or pulsating current is
the value that a smooth (pure) direct current would have to have in
order to cause the same heating effect in a fixed resistive load.
Analogue measurement systems: No significant difference
in price between average-reading and rms-reading a.c.
meters – but no longer in common use.
Digital meter:
Much more expensive if true rms value is really displayed!
Moving-iron meter:
Displays rms value of current.
Moving-coil meter:
Displays average magnitude of current if used in
conjunction with a bridge rectifier.
28.
4. Increased conductor loading
in theory – (e. g. from older style
electronic ballasts):
Line current – average magnitude 179 mA
Line current – rms value 615 mA
Line current – peak value 2712 mA
Apparent power 141 VA
D.c. power 58 W
Line current – form factor 3.436
Line current – crest factor 4.410
Comparison with values for a sinusoidal current:
Form factor 1.1107
Crest factor 1.4142
29.
...and in practice
– e. g. power supply
for a laptop PC
Now let us compare an RMS to a
True RMS meter:
30.
blind
blind
All currents are equal ...
î =1A
R=1Ω
ûR=R*î=1V
pR= ûR*î= ûR²/R=1W
q=2*10ms*1A=20mAs
WR=20ms*1W=20mWs
UR=1V, I=1A
î =2A
R=1Ω
ûR=R*î=2V
pR= ûR*î= ûR²/R=4W
q=2*5ms*2A=20mAs
WR=2*5ms*4W=40mJ
UR=1.414V, I=1.414A
...but RMS currents are less equal than others!
31.
... enables us to infer what
happens when run by a
normal three-phase supply
The same fluorescent lamp
with a magnetic ballast:
The behaviour measured
when connected to
a d.c. supply ...
Behaviour of a 58 W fluorescent lamp connected to a d.c.
supply
0
20
40
60
80
100
120
140
160
180
200
0 200 400 600 800 1000 1200
i / mA
u/V
Measurement
Behaviour of a 58 W fluorescent lamp connected to a d.c.
supply
0
20
40
60
80
100
120
140
160
180
200
0 200 400 600 800 1000 1200
i / mA
u/V
Measurement
Calculation
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
200
250
300
350
0 5 10 15 20
t / ms
u/V
-1,0
-0,8
-0,6
-0,4
-0,2
0,0
0,2
0,4
0,6
0,8
1,0
i/A
Systems voltage
Current
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
200
250
300
350
0 5 10 15 20
t / ms
u/V
-1,0
-0,8
-0,6
-0,4
-0,2
0,0
0,2
0,4
0,6
0,8
1,0
i/A
Systems voltage
Lamp voltage
Current
32.
Warning: “Compensation” is not
always what you think it is!
Distinguish
between:
Reactive power
compensation
a.k.a. power factor
correction
Remedial
measure:
Parallel (or
sometimes series)
compensation
using capacitors
Filter circuits
tuned to the
individual
harmonic
frequencies
Harmonic
compensation
33.
5. (Over)loading the neutral conductor
Magnetic ballast: Old-style electronic ballast:
-2,5A
-2,0A
-1,5A
-1,0A
-0,5A
0,0A
0,5A
1,0A
1,5A
2,0A
2,5A
0 5 10 15 20
t / ms i
i(t) N
-2,5A
-2,0A
-1,5A
-1,0A
-0,5A
0,0A
0,5A
1,0A
1,5A
2,0A
2,5A
0 5 10 15 20
t / ms
i
i(t) N
-2,5A
-2,0A
-1,5A
-1,0A
-0,5A
0,0A
0,5A
1,0A
1,5A
2,0A
2,5A
0 5 10 15 20
t / ms
i
i(t) L1 i(t) L2 i(t) L3
-2,5A
-2,0A
-1,5A
-1,0A
-0,5A
0,0A
0,5A
1,0A
1,5A
2,0A
2,5A
0 5 10 15 20
t / ms
i
i(t) L1
i(t) L2
i(t) L3
34.
Adding the 3rd
harmonics
in the
neutral
wire
-150
-100
-50
0
50
100
150
0° 30° 60° 90° 120° 150° 180° 210° 240° 270° 300° 330° 360°
L1
-150
-100
-50
0
50
100
150
0° 30° 60° 90° 120° 150° 180° 210° 240° 270° 300° 330° 360°
L2
-150
-100
-50
0
50
100
150
0° 30° 60° 90° 120° 150° 180° 210° 240° 270° 300° 330° 360°
L3
-450
-400
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
200
250
300
350
400
450
0° 30° 60° 90° 120° 150° 180° 210° 240° 270° 300° 330° 360°
f
i/î/%
N
Physics
dictates that at
any moment in
time the phase
and neutral
currents must
sum to zero
35.
Effect on a three-phase a.c. motor
appears almost identical:
The same motor is being
driven ...
... simultaneously in
these three modes!
homopolar system,
zero-sequence system
direct system,
positive-sequence system
inverse system,
negative-sequence system
37.
... but the load also
effects the transformer!
Total transformer loss is:
2
)()(
+=
nom
nomCunomFeLoss
I
I
PPP
While total transformer loss really is:
2
)(
2
)(
2
)( *
+
+
=
nomnom
nomad
nom
nomCu
nom
nomFeLoss
I
I
f
f
P
I
I
P
U
U
PP
38.
6. “supplementary” additional
losses in transformers
can be calculated rapidly
using the following two
simple formulae:
5,0
2
2
1
2
1
1
+
+= ∑
=
+
Nn
n
nqh
I
I
n
I
I
e
e
K
( )
5,0
1
2
1
1
5,0
1
2
=
= ∑∑
=
=
=
=
Nn
n
n
Nn
n
n
I
I
III
where
Oh well,
perhaps a practical
example is clearer:
1000 compact 11W
(15VA) energy-saver
lamps powered by a
15kVA transformer,
uSC=4%, Pa=0.1PCu
Harmonics of an Osram Dulux 11W
CFL and a serial impedance of
R =29.1Ω & X L=113Ω
U U² I L I L² P a/P Cu
n V V² mA mA²
1 230.2 52992.0 48.5 2352.3 5.6%
3 8.3 68.9 37.1 1376.4 29.5%
5 10.7 114.5 20.3 412.1 24.5%
7 4.3 18.5 5.3 28.1 3.3%
9 1.1 1.2 3.0 9.0 1.7%
11 2.3 5.3 3.8 14.4 4.2%
13 1.0 1.0 1.5 2.3 0.9%
15 0.6 0.4 1.5 2.3 1.2%
17 1.1 1.2 1.5 2.3 1.5%
19 0.5 0.3 0.9 0.8 0.7%
21 0.5 0.3 1.3 1.7 1.8%
23 0.6 0.4 0.8 0.6 0.8%
25 0.4 0.2 0.6 0.4 0.5%
27 0.6 0.4 0.8 0.6 1.1%
29 0.4 0.2 0.5 0.3 0.5%
31 0.3 0.1 0.5 0.3 0.6%
33 0.3 0.1 0.5 0.3 0.6%
35 0.3 0.1 0.4 0.2 0.5%
37 0.3 0.1 0.4 0.2 0.5%
39 0.3 0.1 0.3 0.1 0.3%
41 0.1 0.0 0.3 0.1 0.4%
43 0.2 0.0 0.2 0.0 0.2%
45 0.1 0.0 0.2 0.0 0.2%
47 0.1 0.0 0.2 0.0 0.2%
49 0.1 0.0 0.1 0.0 0.1%
51 0.1 0.0 0.1 0.0 0.1%
P a/P Cu = 81.4%81.4%
etc.
etc.
39.
Harmonics measurement on an
Osram Dulux 11W compact
fluorescent lamp
Harmonics of an Osram Dulux 11W
CFL and a serial impedance of
R =29.1Ω & X L=113Ω
U U² I L I L² P add/P Cu
n V V² mA mA²
1 232.7 54149.3 48.9 2391.2 3.7%
3 0.6 0.4 39.1 1528.8 21.5%
5 4.4 19.4 26.4 697.0 27.3%
7 2.3 5.3 20.0 400.0 30.7%
9 0.1 0.0 19.2 368.6 46.7%
11 0.1 0.0 16.6 275.6 52.2%
13 0.1 0.0 12.7 161.3 42.7%
15 0.1 0.0 11.0 121.0 42.6%
17 0.1 0.0 10.2 104.0 47.1%
19 0.1 0.0 8.7 75.7 42.8%
21 0.1 0.0 7.7 59.3 40.9%
23 0.1 0.0 7.3 53.3 44.1%
25 0.1 0.0 6.1 37.2 36.4%
27 0.1 0.0 4.9 24.0 27.4%
29 0.1 0.0 4.2 17.6 23.2%
31 0.1 0.0 3.6 13.0 19.5%
33 0.1 0.0 3.0 9.0 15.3%
35 0.1 0.0 3.3 10.9 20.9%
37 0.1 0.0 3.1 9.6 20.6%
39 0.1 0.0 2.5 6.3 14.9%
41 0.1 0.0 2.5 6.3 16.4%
43 0.1 0.0 2.5 6.3 18.1%
45 0.1 0.0 1.9 3.6 11.4%
47 0.1 0.0 1.8 3.2 11.2%
49 0.1 0.0 1.9 3.6 13.6%
51 0.1 0.0 1.6 2.6 10.4%
P a/P Cu = 701.7%
To some extent
the transformer
protects itself...
Always remember to
consider it!
If the influence of the
transformer upon the load did
not exist, then the influence
of the load upon the
transformer would be nearly
9 times as high!
701.7%
etc.
etc.
40.
For this, too, a tool is
provided:
K Factor
Calculator by
www.cda.org.uk
www.cda.org.uk/frontend/pubs.htm#ELECTRICAL/ENERGY%20EFFICIENCY
41.
Rule of thumb: Select transformers
35% larger than specification by
apparent power would require!
This suffices to err
on the safe side
even in the worst
case ... and can
hardly ever be
wrong, since
maximum efficiency
always lies between
25% and 50% of
rated load.
Efficiency of a 1 MVA transformer plotted against loading
98,5%
98,6%
98,7%
98,8%
98,9%
99,0%
99,1%
99,2%
99,3%
99,4%
0% 25% 50% 75% 100% 125%
Relative load
η
Design with max. Cu loss and min. Fe loss
Design with min. Cu loss and max. Fe loss
1 MVA oil transformer according to HD 428
42.
Generator:
uSC≈15%...40%
Extreme example
bicycle dynamo:
uSC≈500%!
The generator also
impacts the load:
Transformer:
uSC=4% / 6%
43.
Measured response of an 11 W
fluorescent lamp operated with a
magnetic ballast
44.
We are
dealing with
a complex
problem
These
effects are
not isolated
but are
mutually
inter-
dependent.
Effects 2, 3 and 5 can be demonstrated
on DKI’s display panel
45.
A conventional, approximately
resistive-inductive load
mA
NL1 L2 L3
Two conventional, approximately
resistive-inductive loads
Three conventional, approximately
resistive-inductive loads
46.
A modern electronic load
mA
N
Two modern electronic loadsThree modern electronic loads
L1 L2 L3
47.
Discoveries on a
refurbished
junction box
System
voltage
(as trigger
signal)
Current in
the earthing
conductor
between the
consumer
unit
and the
bonding
busbar
48.
The TN-C system, that was perfectly
adequate some years ago, is unable to
meet present-day requirements
-6
-4
-2
0
2
4
6
0 5 10 15 20
t / ms
i/A
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
200
250
300
350
0 5 10 15 20
t / ms
u/V
49.
“THF”
(third harmonic
filter) made in
Finland
In certain situations, this affordable
filter can be of help
German
version “THX”
50.
0Ω
5Ω
10Ω
15Ω
20Ω
25Ω
30Ω
35Ω
40Ω
0Hz 50Hz 100Hz 150Hz 200Hz 250Hz 300Hz
f
Z
-90°
-75°
-60°
-45°
-30°
-15°
0°
15°
30°
45°
60°
75°
90°
φ
Reactor
reactance
Capacitor
reactance
Parallel
impedace
Phase angle
Data and frequency
response of the filter
1320 µF
875
14 mΩ µH
51.
The filter in use:
In a nursery running sodium vapour lamps (500 kW)In a typical office and administrative building
52.
The EMC Transformer from Switzerland:7. The effects of stray magnetic fields
in TN-C systems“When we build transformers, the first thing we focus on is complying with our
customers’ requirements and only then on complying with standards. Why? Because it
is customer needs and not standards that offer real scope for product innovation.”
53.
8.. In a TN-C system, problems arise from
data streams and working currents mixing!
Others who have
drawn attention to
this source of data
transmission errors
include
engineering ...
... and insurance
companies, experts,
consultants and
many others besides
54.
9. Increased corrosive damage
This (once) galvanized steel strip – the earth electrode ofThis (once) galvanized steel strip – the earth electrode of
a TN-C-S system – was located closea TN-C-S system – was located close
to the transformer stationto the transformer station
Side facing
transformer
station
Side facing away
from transformer
station
55.
WRONG!
RIGHT!
Working currents have no
place in earthing systems
and protective conductors
57.
Yet another skimpers’ network:
The TT system...
58.
...easily turns into an explosive
»TNT system!«
Storage
room
Storage
room
e. g. for tri nitro tolulene
59.
Some experts already claim
for the »TN-S-S system«
Experience from a radio station:
N current 150 Hz:150 A PE current:32 A
N current 450 Hz:14 A PE current:12 A
And this even in a clean TN-S system
with a CEP!?
60.
Some experts already claim
for the »TN-S-S system«
L1
L2
L3
Fi
N
FE
PE
61.
There is only
one earth
on earth
Don’t you believe it!
Protective earth
Operational earth
Functional earth
Power-system earth
IT earth
My earth
Your earth
62.
So what actually is a TN-S system?
As shown here, this is in
the best case a
»TN-S system h. c.«!
But now it is an
»academic« one.
63.
Which bridge to
cross, which bridge
to burn?
Sometimes you just don‘t
believe what you see!
This one will soon
look rather charred!
IEC 60364-5-54:
543.4.3 If, from any point of the installation, the neutral and protective
functions are provided by separate conductors, it is not permitted to
connect the neutral conductor to any other earthed part of the
installation.
64.
Well, then the rest is no longer aWell, then the rest is no longer a
miracle!miracle!
65.
This has, at last, been addressed in EN 50174-2
Disturbing
equipment
Sensitive
equipment
Disturbing
equipment
Sensitive
equipment
Disturbing
equipment
Sensitive
equipment
Not recommended
Transformer
Better Excellent
in the most
recent edition
from
Sept. 2001
– the only
problem is ...
it’s wrong!
66.
Though they’d already got it right
in version 2:2000!
67.
“The PEN conductor plays a dual
role in TN-C systems. Its primary role is as
a protective conductor, its secondary
function is that of a neutral or return wire.”
(Volker Schulze:
“Vorgefertigte Klemmenblöcke für PEN-Leiter-
Verlegung”
[“Prefabricated terminal blocks for PEN conductor
installation work”], in “de” 13/1999, p. 1050)
Well, we wouldn’t mind if a PEN con-
ductor generally just looked like this...
Yesterday’s truths ...
68.
Cables appropriate for
modern electrical
installation work are
available
Still too much of this around:
Standard four-core
for the miserly and
the short-sighted
Contact International Cablemakers Federation
Graben 30
A-1014 Wien
Phone: + 43 1 532 9640
Fax: +43 1 532 9769
http://www.icf.at
69.
And has already integrated it
into its Logo:
4-core good, 5-core better!
One fabricator
says it quite
clearly right
from the start:
Turn your
brains on!
70.
Take a look at EN 50174 from
Feb 2000, subclause 6.4.3:
“…it shall be considered that a PEN conductor through which
the unbalanced currents and the accumulating of harmonic
currents and other disturbances are transmitted cannot
provide an appropriate earthing. It shall also be considered
that the TT and IT mains systems need more corrective
measures in particular against over voltage; therefore:
there should be no PEN within the building, i.e. the
respective option in 546.2.1 of HD 384.5.54 S1:1988
should not be used.
wherever possible, the TN-S system should be used.”
71.
Have a look at CENELEC Guide
R064-004:1999-10, where it says:
For buildings which have, or are likely to
have, sigificant information technology equip-
ment installed, consideration shall be given to
the use of separate protective conductors
(PE) and neutral conductors (N) beyond the
incoming supply point in order to minimize the
possibility of electromagnetic problems due to
the diversion of neutral current through signal
cables causing damage or interference
72.
Or at EN 50160:
Under normal operating conditions rapid voltage changes usually remain below 5% of
the rated voltage, but deviations of up to 10% may under certain circumstances occur
several times a day.
Under normal operating conditions the number of voltage dips lies between some tens
and several thousands per year.
They usually last less than 1 s and have a retained voltage of over 40%.
Short interruptions of up to 3 minutes occur some tens up to several hundred times a
year. Up to 70% of these may last for less than 1 s.
Unbalance: 95% of all 10-minute mean values must have an inverse system of less
than 2% the direct system. But where there are many singe- and two-phase loads in
operation, it may as well give rise up to 3%.
Switching transients usually do not exceed 1.5 kV, other transients commonly stay
below 6 kV. In individual cases, however, they may be higher than that.
The voltage magnitude on the low voltage level is 230 V ±10%,
measured between phase and neutral conductor in 4-conductor systems
and between phase conductors in 3-conductor systems.
5-conductor systems obviously do not exist!
73.
Or at EN 50160:
The frequency should be between 49.5 Hz
and 50.5 Hz for at least 99.5% of a given
year.
Island networks not running synchronous to
the UCTE mains are exempted.
So this already starts with the British isles,
doesn‘t it?
74.
Or at EN 50160:
Indeed...
...But let us
have a look
at a »real«
island –
here it
comes:
75.
Observations on Malta:
Frequencies between
49.80 Hz and 50.13 Hz
While all of this discussion (and the
measurement) apparently proves to be a
waste, since:
Or at EN 50160:
These limits refer to normal operating
conditions only, not to fault conditions.
So does the responsibility for supply
drop out when power supply drops out?
76.
Hopefully we will never ever get
a supply according to EN 50160!
May be you better have a look at VDE 0100
section 100 of August 2002 first of all:
“The features given in DIN EN 50160:2000-
03 represent extreme situations but do not
describe the usual situation in the mains. For
planning electrical installations with a normal
usage it is sufficient to consider the most
likely situation in the mains at the point of
common coupling.”
77.
Or how about
IEC 60364-4-44 from 2001:
“For buildings which have, or are likely to have, significant
information technology equipment installed, consideration
shall be given to the use of separate protective conductors
(PE) and neutral conductors (N) beyond the incoming
supply point, in order to minimize the possibility of
electromagnetic problems due to the diversion of neutral
current through signal cables causing damage or
interference.”
Fortunately, the 2002 edition now emphasises doing
something rather than just thinking about doing it.
Unfortunately, the word “shall” has been replaced
with “should”.
78.
The recent amendment to
EN 50310, section 6.3
from Sept. 2000 appears to have got
exactly the right approach at last:
“The AC distribution system inside a
building shall conform to the requirements
of the TN-S system. This requires that there
shall be no PEN conductor inside the
building, i.e. the option in 546.2.1 of HD
384.5.54 S1:1980 shall not be used.”
79.
So why all the fuss about interference
suppression? After all, the EN 61000-3-2
standard has been in force since 1 Jan. 2001
– so interference is a thing of the past!
Right ...?
Class B: Portable power tools
Class A: Balanced three-phase equipment and all other
equipment not classified below
Class C: Lighting equipment including lighting controls
(except dimmers up to 1000 W)
Class D: Personal computers, PC monitors and televisions
with an input power range from 75 to 600 W.
We now have limits on harmonic emissions for ...
80.
The tolerances that the mains
voltage has to meet are very tight:
2.0% max. permissible deviation from rated value,
0.9% max. permissible content of 3rd harmonic,
0.4% max. permissible content of 5th harmonic,
0.3% max. permissible content of 7th harmonic,
0.2% max. permissible content of 9th harmonic,
0.2% max. permissible content of even-order harmonics
0.1% max. content of harmonics of the orders 11 to 40
during testing!
It’s important that these tolerances are so tight because the
effect of voltage distortions is huge ...
81.
And what exactly are the specified limits?
e.g. for Class D equipment
“For the 3rd harmonic, a Class D unit must draw no more
than 3.4 mA per watt of input power.”
And how do we ensure compliance ...
No problem ...
But ... are we talking about the rated or measured input
power ?
And what about the power system parameters (e.g.
resistance, reactance, voltage profile) ?
... in the case of, say, a conventional PC ?
82.
Emission limit (3rd harm.): 116 W x 3.4 mA/W =395 mA
Measured emission (3rd harmonic): 411 mA
Solution 1: Specify rated power 4 % above the measured
value
Solution 2: Consider each device separately
(PC: 40 W, Monitor: 60 W, Peripherals: 16 W)
Solution 3: Add an ohmic resistance in series
Difference to be “measured away”: 16 mA
Class D: Personal computers, PC monitors and tele-
visions with an input power range from 75 W to 600 W
83.
Solution 4:
The perfect PC
for all editorial staff...
Jack the power up to over 600
W by connecting a large
resistive load in parallel.
Class D:
Personal computers, PC
monitors and televisions with
an input power range
from 75 W to 600 W
84.
Another means of
deforming the current
profile:
The phase-control
dimmer
L1-N L2-N L3-N
φ 0° 0° 0°
u AV 207,1 207,1 207,1 V
UEff 230,6 230,6 230,6 V
L1 L2 L3 N
i AV 0,235 0,235 0,235 0,000 A
IEff 0,261 0,261 0,261 0,000 A
IEff / i AV 1,114 1,114 1,114 ---
î / i Eff 1,410 1,410 1,410 --- -0,4A
-0,3A
-0,2A
-0,1A
0,0A
0,1A
0,2A
0,3A
0,4A
0 5 10 15 20
t / ms i
-350V
-250V
-150V
-50V
50V
150V
250V
350V
0 5 10 15 20
t / ms
U
L1-N L2-N L3-N
φ 60° 60° 60°
u AV 153,7 153,7 153,7 V
UEff 205,8 205,8 205,8 V
L1 L2 L3 N
i AV 0,174 0,174 0,174 0,181 A
IEff 0,233 0,233 0,233 0,204 A
IEff / i AV 1,339 1,339 1,339 1,128
î / i Eff 1,581 1,581 1,581 1,562 -0,4A
-0,3A
-0,2A
-0,1A
0,0A
0,1A
0,2A
0,3A
0,4A
0 5 10 15 20
t / ms i
-350V
-250V
-150V
-50V
50V
150V
250V
350V
0 5 10 15 20
t / ms
U
L1-N L2-N L3-N
φ 45° 90° 135°
u AV 176,7 101,7 30,3 V
UEff 219,9 161,3 69,5 V
L1 L2 L3 N
i AV 0,200 0,115 0,034 0,155 A
IEff 0,249 0,183 0,079 0,188 A
IEff / i AV 1,244 1,585 2,293 1,213
î / i Eff 1,479 2,016 3,251 1,702 -0,4A
-0,3A
-0,2A
-0,1A
0,0A
0,1A
0,2A
0,3A
0,4A
0 5 10 15 20
t / ms i
-350V
-250V
-150V
-50V
50V
150V
250V
350V
0 5 10 15 20
t / ms
U
85.
No-load currents in transformers:
a further – though overrated – source of
current distortion
No-load current in a 630 kVA
distribution transformer, excitated
from the 420 V LV side!
86.
Another means of (over)loading the
neutral conductor:
d.c. currents ...
Original “Fake”
87.
... and their
effects
U = 228.5 V
I = 9.2 mA
P = 1.82 W
S = 2.09 VA
Q = 1.03 VAr
LF = 0.87
cos φ = 0.99
U = 224.3 V
I = 1.26 A
P = 38.0 W
S = 286.0 VA
Q = 283.0 VAr
No-load current in a
toroidal-core
transformer
200 VA
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
200
250
300
350
0 10 20 30 40
t / ms
u/V
-12
-10
-8
-6
-4
-2
0
2
4
6
8
10
12
i/mA
Voltage
Current
No-load current in a toroidal-core
transformer 200 VA when
running a 1500 W
hairdryer at
half-power
in parallel
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
200
250
300
350
0 5 10 15 20 25 30 35 40
t / ms
u/V
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
6
i/A
Voltage
Current
88.
The later models don‘t do it any more
The EN 61000-3-2 has at least managed
to put an end to this approach.
Domestic appliances: Max. 1.05 A of 2nd harmonic
89.
An electrical engineering wonder:
d.c. current generates a.c. voltage
90.
Spicing things up with a pinch of HF:
electronic halogen lamp transformers
92.
Using
simple
materials to ...
... model the
electrical
installation in
a building
93.
A power system that
doesn’t get dirty
doesn’t need to be
cleaned:
0
50
100
150
200
250
300
350
400
450
500
20 30 40 50 60 70 80 90 100 110 120
f / Hz
Z/Ω
-90°
-60°
-30°
0°
30°
60°
90°
φ
XL
XC
ZS
ϕ
94.
Current and power loss as a function of voltage
0,0
0,2
0,4
0,6
0,8
1,0
1,2
1,4
1,6
1,8
0 40 80 120 160 200 240
U / V
I/A
0
20
40
60
80
100
120
PV/W
I [A]
Pv [W]
Power loss, capacitor voltage
and total voltage as a function of current
0
50
100
150
200
250
300
350
400
450
0,00 0,25 0,50 0,75 1,00 1,25 1,50
I / A
U/V
0
5
10
15
20
25
30
35
40
45
50
PV/W
Utot [V] Eff
UR [V] Eff
Pv [W]
The passive
harmonic filter
made of readily
available
components ...
... is both
reliable ...
96.
Half the solution has already been implemented:
97.
Professor Manfred Fender
Wiesbaden University of Applied Sciences:
We should use more two- and three-phase rectifier loads!
L3
L1
L2
N
i
u
L1
t
N
i
prU
t
u
L3
L2
t
Usek
t
u
i
B2B6 B2 B2
i
t
u
i
u
Bild d Bild c Bild b Bild a
Bild e
98.
We should use
transformers
with different
vector groups!
usc for zero-seq. system:
60% of rated value
usc for zero-seq. system:
5...10% of rated value
99.
And if that’s not enough ...
Passive filter circuits will do it
100.
So what do we need to do?
In low-voltage power distribution systems:
Do not use cables in which the neutral or protective earth conductor
has a reduced cross-section.
Use only 5-core cable. Do not install TN-C- or TN-C-S systems.
Be generous when dimensioning conductor cross-sections. This helps
to reduce line voltage drops and thus reduces the effect of current
distortions on the voltage, as well as lowering energy losses (see VDE
0298 Part 100). Generously dimensioned conductors will have no
problem coping with any future increase in demand.
Only use measuring instruments that display the true root mean square
value (TRMS meters).
In medium-voltage power distribution systems:
Use a varied mix of distribution transformers with different vector
groups.
101.
the European Union has been providing a total of three million euros over a three year
period to enable experts from across Europe to co-operate in the development of the
definitive internet site covering all aspects of power quality!
To follow the latest developments visit
www.lpqi.org
and take a look at the growing body of information that has been made available by the
Leonardo Power Quality Initiative.
Our aim is to develop and disseminate teaching materials in 13 languages dealing with
the detection, mitigation and management of EMC problems.
Target groups include electrical technicians, engineers, those in the skilled trades,
building system engineers, architects, planners as well as apprentice technicians and
students and their teachers.
At present the Power Quality Initiative has 106 members from commercial companies,
institutions, universities and trade associations.
We openly encourage other industrial and academic partners to participate in this project
and welcome contributions at any time.
Just log on!
www.lpqi.org
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