1.
A novel transimpedance amplifier with variable gain Pietro Monsurrò, Alessandro Trifiletti Trond YtterdalDipartimento di Ingegneria dell’Informazione, Elettronica e Department of Electronics and Telecommunications Telecomunicazioni Norwegian University of Science and Technology Università di Roma “Sapienza” Trondheim, Norway Roma, Italy trond.ytterdal@iet.ntnu.no monsurro@die.uniroma1.it; trifiletti@die.uniroma1.itAbstract—In this paper we propose a variable-gain Transistor M1 is biased in the triode region, whereas the othertransimpedance amplifier suitable for low-power applications. Its two are biased in saturation. VCTRL controls the transimpedancenoise, bandwidth and input impedance performance are similar gain, and VBIAS determines the bias current.to a more conventional regulated-cascode common-gatetransimpedance with resistive load, with the same powerconsumption and gain performance. The proposed amplifier has,however, variable gain, which can be easily changed by setting acontrol voltage. Besides, it uses no passive components and can Vbias M3thus occupy less space in the layout, a feature of interest inapplications which require the use of many sensors. With 30µWdissipation, it achieves 800MHz performance with 50fF input and Voutoutput loads, in a 65nm CMOS technology. The transimpedancegain is 68dB, and the input impedance is 180Ω. Vctrl M2 Keywords-transimpedance, variable gain, front-end Iin I. INTRODUCTION M1 Transimpedance amplifiers are widely used in applicationswhere the signal from a current-mode sensor needs to be turnedinto a voltage. Transimpedance amplifiers are used in opticalcommunications [1], because the photodiode sensor can be Figure 1. Proposed transimpedance topology.modeled as a current source, and in ultrasound imaging [2]. Inthis last kind of application, it is important to have variablegain in order to equalize the amplitude of the echoes. Besides, B. Frequency responsein a beam forming network with thousands of sensors, it is Under the hypotheses that each transistor has infinite outputpreferable to use very simple topologies with just a few active impedance (except M1, which works in the triode region, anddevices and no passives. whose output conductance is Go1), that the gate-to-drain In this paper we propose a novel transimpedance topology capacitance is negligible, and that the input and output loadwith variable gain, good linearity, noise and bandwidth capacitors Ci and Co (not shown in Fig. 1) are much larger thanperformance. It only uses three active devices and no resistors the gate-to-source capacitors of the devices, theare required. Simulations have been performed using the transimpedance gain of the stage is given by:STMicroelectronics CMOS 65nm low-power process,employing 1.2V devices. gm2 Zm = (1) II. THE PROPOSED TRANSIMPEDANCE g m1 g m 2 + sCo ( g m 2 + Go1 ) + s 2 Ci CoA. Topology Fig. 1 shows the proposed transimpedance amplifier. where gm1 and gm2 are the transconductance gain of M1 and The circuit has only one current branch, it is based on the M2, respectively.flipped voltage follower (FVF) topology [3], and it exploits The low-frequency transimpedance is thus equal to thefeedback to reduce both its input and output impedances. inverse transconductance of M1, and a flat frequency response 978-1-4244-8971-8/10$26.00 c 2010 IEEE
2.
can be easily achieved by increasing Co, Go1 or g2 with respect Assuming ideal MOS devices, the output voltage is relatedto Ci or g1. to the input current by these equations: On the other hand, the input impedance is: VI VO + vO = Z m I B + + VT + Z m iS (4) sCo 2 Z in = (2) 1 g m1 g m 2 + sCo ( g m 2 + Go1 ) + s 2 Ci Co Zm = (5) 2 K1VI IB Due to the assumption of infinite open-loop gain, the input VI = VCTRL − VT − (6) K2impedance at low frequencies is zero, although in reality finitegain effects increase the minimum achievable impedance. Inthe 65nm process, the loop gain is about 20dB, and thus the where K1 and K2 are the non-linear transconductance of M1low-frequency input impedance is about one tenth of the and M2, respectively, and iS is the input signal. VO is the biasinverse of the transconductance of g2. point at the output and vO is the output signal around the bias point. It can be noted that the output voltage vO and the inputC. Signal-to-noise ratio current iS are linearly related through the transimpedance gain The transimpedance amplifier is often used to sense very Zm, and that this gain can be varied by changing the voltage onsmall currents arising from sensors such as photodiodes. The the drain of M1 (VI), by changing VCTRL. With ideal MOSsignal-to-noise ratio is thus a key specification for a devices, the transimpedance should be linear.transimpedance amplifier. Assuming ideal MOS devices, with white drain noise III. SIMULATIONScurrent equal to in1, in2, in3, respectively, and an input currentsource iS, we have that the signal-to-noise ratio (SNR) is: A. Benchmark The proposed transimpedance has been compared with a more standard solution, a common-gate amplifier with resistive load RO, and regulated cascode feedback to improve the input is2, rms impedance [4], as shown in Fig. 2. The same power SNR = (3) 2 2 in1 + in 3 consumption, supply voltage, and transimpedance gain have been used to design this amplifier, so as to make the comparison as significant as possible. However, the proposed amplifier has also gain control, whereas the benchmark where the noise due to devices M1 and M3 needs to be transimpedance has fixed gain, being it equal to RO (althoughintegrated over the relevant signal bandwidth. Noise due to M2 this resistor could be realized using active devices to create agives no contribution because of the infinite gain of the variable resistor, this would impact linearity).cascode structure M1–M2. Under realistic conditions, its noisecontribution is divided by the voltage gain of a MOS device, The topology in Fig. 2 uses one resistor, and thus it is likelyi.e., by a factor of about 10. to occupy a larger space in the layout than the proposed topology, which only uses active devices.D. Linearity and gain control Calculations show that the two topologies have roughly If M1 is biased in the saturation region, the circuit is heavily similar input impedance, noise and bandwidth, for the samenon-linear because the transconductance of M1 changes with transimpedance and power consumption.the output voltage. On the other hand, at least with ideal MOSdevices, when M1 is in the triode region there are twoimportant consequences: the transimpedance becomes roughlylinear and the voltage VCTRL determines the transimpedance Rogain (although it also affects the output bias point), whichmoreover depends linearly on the control voltage. Vbp M4 Vout Linearity is due to the constancy of the input voltage, whichcauses the non-linear term in the triode current equation to be M1constant. This is the result of assuming infinite loop gain, andthus zero input impedance. The non-linear behavior in the M2 Iintriode region depends on the square of the input voltage,whereas in the saturation region it is due to the square of the Vbn M3output voltage, thus linearity improves in the triode region evenwith finite gain. Figure 2. Conventional transimpedance amplifier.
3.
B. Simulation results -120 Total power consumption is 30µW for both amplifiers, andthe supply voltage is 1.5V. With a bias current of 20µA, the -125input current swing of 10µA used in the simulations represents -130 Noise power density spectrum (dB)about one half of the dynamic range. The input and output -135nodes are loaded with 50fF capacitors. -140 Fig. 3 shows the transimpedance gain of the proposed(solid) and conventional (dashed) amplifiers, Fig. 4 shows the -145input impedance, and Fig. 5 the output noise spectrum. -150 -155 74 -160 72 -165 -170 70 -3 -2 -1 0 1 2 3 Transimpedance (dBOhm) 10 10 10 10 10 10 10 Freq (MHz) 68 Figure 5. Output noise spectrum (solid: proposed, dashed: conventional) 66 Fig. 6 shows the transimpedance of the amplifier for 64 varying values of the control voltage VCTRL. At very low values for this voltage, the transimpedance increases, but the 62 bandwidth is reduced. Beyond a certain value, M1 goes in the saturation region and the transimpedance becomes independent 60 on the control voltage. Fig. 7 shows the transimpedance and the bandwidth of the proposed transimpedance amplifier as a -3 -2 -1 0 1 2 3 10 10 10 10 10 10 10 Freq (MHz) function of the control voltage. The transimpedance gain can be varied by about 10dB by changing VCTRL. A higher range of Figure 3. Transimpedance gain (solid: proposed, dashed: conventional) programmability can be achieved for higher current densities (higher gate-to-source voltages) because M1 remains in the 75 triode region for a wider range of VCTRL values. The maximum transimpedance gain is set by bandwidth requirements, because 70 g2 and GO1 shrinks with VCTRL, limiting the bandwidth of the complex poles (or even creating two real separated poles). The 65 minimum gain, on the other hand, is set by linearity Input Impedance (dBOhm) requirements, because linearity quickly worsens when M1 60 moves toward its saturation region. 55 84 50 82 80 45 Transimpedance (dBOhm) 78 40 -3 -2 -1 0 1 2 3 10 10 10 10 10 10 10 76 Freq (MHz) 74 Figure 4. Input impedance (solid: proposed, dashed: conventional) 72 The two amplifiers have roughly the same performance interms of noise, bandwidth and input impedance. The cascode 70amplifier, however, has higher overshoot. This overshoot canbe compensated by increasing the load capacitor on the output 68 -3 -2 -1 0 1 2 3of the auxiliary amplifier M2–M4, but at the expense of 10 10 10 10 Frequency (MHz) 10 10 10bandwidth. The proposed amplifier also has a lower inputimpedance at high frequencies. The frequency response of the Figure 6. Frequency response for varying VCTRL.benchmark transimpedance amplifiers is more complicatedbecause it has one more pole, and one zero.
4.
impedance, whereas it has some limitations in terms of Transimpedance (dBOhm) 84 linearity with respect to a regulated cascode common-gate 82 amplifier. The transimpedance can be varied in a range of 80 about 10dB by setting a control voltage, and a wider range of 78 variability can be achieved at higher current densities. 76 The proposed amplifier doesn’t use any resistor and so it is 74 0.45 0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 suitable for very high-density integration: this may be important for applications such as ultrasound imaging, in 1000 which many ultrasound sensors are used to create sensor arrays with beam forming ability. Bandwidth (MHz) 800 600 70 400 200 65 0 60 0.45 0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 Vctrl (mV) 55 HD2 & HD3 (dB) Figure 7. Transimpedance and bandwidth vs VCTRL. 50 Fig. 8 shows the second and third harmonic distortion for 45several values of VCTRL. Although HD3 is heavily dependent on 40the control voltage, the improvement in terms of HD2 isreduced. The amplifier has a worse HD2 performance with 35respect to the benchmark case (50dB), whereas it achieves a 30comparable HD3 only in deep triode region, for low values ofVCTRL. The HD3 performance for the benchmark amplifier is 25 0.45 0.5 0.55 0.6 0.65 0.768dB. Tab. I summarizes the results. Vctrl (V) TABLE I. SIMULATED PERFORMANCE Figure 8. HD2 and HD3 vs VCTRL. Proposed TZA Benchmark TZA Unit Noise Power Density 4.7 4.3 pA/√Hz REFERENCES [1] Hasan, S.M.R., “Design of a low-power 3.5-GHz broad-band CMOS Bandwidth 850 830 MHz transimpedance amplifier for optical transceivers”, Circuits and Systems I: Regular Papers, IEEE Transactions on, Volume: 52 , Issue: 6, 2005, Transimpedance 68 69 dBΩ Page(s): 1061 – 1072. Overshoot 0.2 3.0 dB [2] Cenkeramaddi, L.R.; Ytterdal, T., "1V transimpedance amplifier in 90nm CMOS for medical ultrasound imaging", NORCHIP, 2009, Input impedance 45 45 dBΩ Page(s): 1 – 4. [3] Carvajal, R.G.; Ramirez-Angulo, J.; Lopez-Martin, A.J.; Torralba, A.;Second harmonic distortion 28 50 dB Galan, J.A.G.; Carlosena, A.; Chavero, F.M., "The flipped voltageThird hardmonic distortion 51 68 dB follower: a useful cell for low-voltage low-power circuit design", Circuits and Systems I: Regular Papers, IEEE Transactions on, 2005, Volume: 52 Issue: 7, page(s): 1276 - 1291. [4] Sackinger, E., “The Transimpedance Limit”, Circuits and Systems I: IV. CONCLUSIONS Regular Papers, IEEE Transactions on, Volume: 57 , Issue: 8, 2010, Page(s): 1848 – 1856. A novel transimpedance amplifier with variable gain,suitable for low-power applications, has been proposed. It hasgood performance in terms of noise, bandwidth and input
Be the first to comment