A Self-Oscillating LNA-Mixer                                         Tero Koivisto                                        ...
RF differential pair is now a LNA, a transconductor designed for low      B. Noise and Phase-Noiseinput noise and 50 ohm i...
TABLE I         P HASE NOISE CONTRIBUTION OF EACH NOISE SOURCE                               Noise source          Contrib...
Vdd                                                            Vdd                                                        ...
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  1. 1. A Self-Oscillating LNA-Mixer Tero Koivisto Esa Tiiliharju University of Turku University of Turku Microelectronics Laboratory Microelectronics Laboratory Turku Center for Computer Science, TUCS Turku, Finland Turku, Finland Email: tejuko@utu.fi Vdd Vdd Abstract—In this paper, a new circuit topology to realize a stacked self-oscillating LNA-Mixer is proposed. The basic idea has been to recognizethat in a high-performance down-conversion mixer its RF input-stagegain, linearity, and noise tradeoff is often improved by feeding it with lo1 lo2 lo1 lo1 lo2 lo1a bypass current source. This current source could be isolated with aninductor so as to allow free implementation of the oscillator block ontop of it. Using these guidelines, the presented circuit achieves high- rf1 rf2 rf1 rf2performance without sacrificing compatibility with modern low-voltage a) b)CMOS implementations. To further demonstrate usefulness of the circuit,an entire single-stage quadrature (IQ) RF front-end using this circuit as Vdd Vdda core has been developed. The IQ front-end, targeted for the Galileosatellite navigation system, has been designed using a 65-nm CMOS VCOtechnology, and it achieves NF=4.4 dB, IIP3=-15 dBm and Av =25 dB at1.575 GHz, while using only 1 mA from the low 1.2-V supply. lo1 lo2 lo1 lo1 lo2 lo1 I. I NTRODUCTION The growing market for satellite navigation systems, such as rf1 rf2 rf1 rf2Galileo and GPS, is considered the killer application for high- c) d)performance ultra low-power transceivers. For example, it is esti-mated that EU companies, users and society can make at least 55 Fig. 1. The mixer circuit configurations: a) Gilbert mixer with/without current bleeding, b) Gilbert mixer with current reused bleed technique, c) Gilbertbillion euros benefits from Galileo over the next 20 years. mixer with current bleeding through differential inductor center tap and d) a A popular technique to achieve a low-power consumption in RF current reused Gilbert mixer-oscillator.front-end circuits is reusing the current between different functionalblocks, i.e. the same DC bias current is shared between differentcircuit functionalities [1]. A well-known example of this trend is current source supplies part of the RF input stage current, the largercascoding the Gilbert mixer on top of the LNA circuit [2], while gm or RL can be utilized with RF and LO transistors operatingless popular is stacking voltage-controlled oscillator (VCO) [3] [4]. safely in saturation. The current source can be realized for exampleRecently, an entire RF front-ends merging the functionalities of the using PMOS current sources or passive resistor. Fig. 1b shows aLNA, mixer and oscillator in a single circuit has been proposed [1] current reused bleeding Gilbert mixer. In this topology, a PMOS[5]. A widely used technique to realize a high performance low- transistor operates simultaneously as a bleed current source and asvoltage RF CMOS circuits, such as mixer, is current boosting [6]. a part of the RF input amplification. In this case the higher overallThe intention of this idea is to allocate a smaller bias current to the transconduntance reduces the NF [9]. Fig. 1c shows a Gilbert mixer,switching transistors and to the load resistors than to the RF input wherein the boost current is provided via a differential inductor centerdevice. This relaxes the trade-offs between linearity, gain and noise tap. The inductor tunes out the parasitic capacitances present in switchfigure (NF) [7]. transistor sources at the LO frequency and lowers the flicker noise In this paper, a new RF front-end suitable for low-voltage operation of the mixer output. In addition, the boost current source noise isis presented. The front-end, called self-oscillating LNA-mixer, per- minimized [10] [11].forms amplification, down-conversion and local oscillator (LO) signal The intention is to use the boost current to realize other functionali-generation while sharing the same DC bias current between different ties demanded in the receiver. The most power-efficient approach is tocircuit blocks of the RF front-end resulting in a robust operation and use that current to provide LO signals. Fig. 1d shows a Gilbert mixer,low-power consumption (1.2 mW). wherein we have used the boost current to realize LO signals. The resulting circuit is a self-oscillating mixer (SOM), wherein the current II. E VOLUTION TOWARDS THE PROPOSED CIRCUIT allocation between RF, VCO and quad transistors can be selected The evolution towards the proposed circuit is seen from Fig. 1. A for best performance [4]. Furthermore, the RF signal present at theCMOS Gilbert mixer is shown in Fig. 1a. In a low-supply voltages, it outputs of the mixer RF transistors is zero at the inductor center tap,is typically composed of a grounded-source pair, which converts the since it is a virtual ground [10].RF input voltage to current-mode signal. The current-mode signalis then fed to the switching quad, which is driven by a large LO III. T HE S ELF -O SCILLATING LNA-M IXER AND CIRCUITsignal. The output current of the mixer in low-noise applications is ANALYSISusually driven to the passive resistor load (RL ). Fig. 1 (dashed lines The proposed self-oscillating LNA-Mixer circuit is shown in Fig. 2,and box) shows a current boosted Gilbert mixer [8]. Since boost where we have used the VCO topology presented in [12] [13]. The978-1-4244-8971-8/10$26.00 c 2010 IEEE
  2. 2. RF differential pair is now a LNA, a transconductor designed for low B. Noise and Phase-Noiseinput noise and 50 ohm input impedance. The chosen LNA topology In active current commutating CMOS mixers, switches contributeis a widely used inductively degenerated common-source LNA [8]. flicker noise to the mixer output in two different way. In direct The input impedance match to 50 ohm at the wanted frequency ω mechanism, flicker noise modulates the time instants of mixer switch-is achieved when the real part of the (1) is ing whereas in indirect mechanism, it induces current in the tail capacitance Cp , which is commutated to the output. The inductor Ls gm Rs = = ωt Ls (1) tunes out the Cp at the LO frequency and suppress the indirect Cgs mechanism causing flicker noise. The direct mechanism [8], and the imaginary part is set to zero [14], which leads to vn in = 4Isw (10) 1 ST ωo = (2) Cgs Lg + Ls where Isw is the bias current of each switch pair, vn is the equivalent flicker noise of the switching quad, S is the slope of the It is seen, that in practice the impedance to 50 ohm is achieved by LO signal at the switching event and T is LO period. It is seen thatcontrolling the value of the Ls , whereas the Lg sets up the wanted flicker noise caused by this mechanism can be reduced using largeoperating frequency by resonating out the parasitic gate capacitance. W/L ratio, reducing the switch current or increasing the slope of the LO signal [8]. The thermal noise contribution from the VCO isA. Gain minimized since its noise is common mode at the mixer output due The effective transconductance of the LNA part of the proposed to boosting through the differential inductor [10]. Therefore, the totalcircuit is [14] thermal noise voltage of the mixer output is approximately [16] 2 2RL Isw Gm = gm Qin (3) Von = 8kT RL (1 + γ + γgm RL ) (11) πA Overall, the gain of Gilbert cell mixer is [8] where γ is the FET noise factor and A is the LO signal amplitude. The three terms in (13) are output noise voltages due to the load 2 resistors, the mixer differential switches and the input transconductor Av = gm RL (4) π stage [16]. In the proposed circuit, the Isw is low and LO amplitude Taking into account the effective transconductance Gm of the LNA is rather high leading to small switch noise contribution. Furthermore,stage, the gain of the LNA-Mixer is assuming that the Gloss in (11) is ≈1, the total input-referred DSB white noise spectral density is [16] 2 2 ωo (Lg + Ls ) Av = Gm RL = gm RL (5) γ π π 2Rs DSBVin ≈ 2π 2 kT 2 (12) gm Q2 in However, due to current division between mixer quad and differen- wherein the assumption is that the input transconductor dominatestial inductor, part of the RF current flows to the differential inductor the noise.[15]. Fig. 3 shows a simplified small-signal model seen at the sources As far as the oscillator is concerned, the main drawback of theof mixer quad transistors M3 -M6 . The parasitic capacitance Cp seen proposed circuit is that the effective supply voltage used in the designat the source nodes A and B is resonated out with the differential of the VCO is reduced by the Vds of the input RF transistors. Theinductor at the desired operating frequency. The impedance seen at phase-noise of the oscillator according to heuristic Leeson formulathe source nodes of each quad transistor is 1/gm . For differential is [17]operation, the inductor center tap is a virtual ground and thereforethe effective impedance seen at the source nodes A and B is kT ω 2 SSSB = F (13) 1 1 2Psig Q2 Δω 2 Z= || ||Rp (6) gm gm where Q is the loaded quality factor of the tank, Δω=2πΔf is the angular frequency offset, and F is noise factor. It is seen that where in order to realize high spectral purity oscillator, it is important to maximize Q, Psig (or Vsig ) and minimize F. The F is a noise factor Rp = ωLQ (7) and analyzed in detail in [18]. To maximize Q in (16) the inductor must be chosen carefully, because the Q, especially at low microwave and Q is the quality factor of the inductor. Due to current division frequencies (≤ 5GHz) is mainly determined from the Q of inductor.the gain of the LNA-Mixer is reduced by to first order by a factor of Furthermore, in the proposed circuit the VCO is capacitively loaded gm + gm 2gm enabling a high loaded Q. Overall, the main drawbacks are the lower Gloss = 1 = 1 (8) effective supply voltage and the presence of the complex biasing g m + g m + Rp 2gm + Rp arrangement around the VCO in this circuit. To minimize the loss, the Rp should be as high as possible. This The achievable spectral purity characteristics of the proposedis the case when the L and Q are made as large as possible, as can be circuit techniques was determined using Spectre RF simulations. Theseen from (9) [15]. Finally, the gain of the proposed self-oscillating RF transistors M1 and M2 are biased using a current mirror to carryLNA-Mixer is 0.5 mA each, a total of 1 mA. This bias current is entirely by-passed to the VCO through differential inductor. The LC-VCO is voltage 2 2 ωo (Lg + Ls ) biased to operate in class-C mode for optimal performance. The mixer Av = Gm RL Gloss = gm RL Gloss (9) π π 2Rs quad is voltage biased to carry zero current. The differential 8 nH
  3. 3. TABLE I P HASE NOISE CONTRIBUTION OF EACH NOISE SOURCE Noise source Contribution Rvco Drain current 37 % Inductor 33 % Voltage bias [V] 20 % Rs Rs Current bias 5% Other 5% 1/gm L 1/gm 1/gm L 1/gm A B Vdd Cp Cp lo11 lo22 Fig. 3. The equivalent model of loading circuit seen at the source nodes of the switching quad M8 Vbias M9 by one phase of the LO, and the other pairs on the right by out1 out2 its quadrature phase. The quadrature LO signal using the circuit presented in [20]. Furthermore, the NMOS transistors are voltage biased to operate in class-C mode. The quadrature functionality lo1 lo2 lo1 is accomplished using linear-region transistors in series with the M3 M4 M5 M6 gates of the LC-VCO PMOS active transistors. The coupling devices modulate the negative gm of the switching pairs to accomplish anti- phase injection-locking. The coupling devices do not consume voltage Lg Lgrf1 rf2 headroom and dissipates no power. Furthermore, to first order, the M1 M2 devices do not contribute to phase-noise [20]. Moreover, only one differential inductor is needed in this topology, which saves silicon Ls Ls area. The voltage conversion gain of the circuit is LNA 12 Av = gm Qin RL (15) Fig. 2. The proposed Self-Oscillating LNA-Mixer 2π where the factor of 1 results from the fact that the LNA drives 2 two mixers in parallel. The noise performance of this topology asinductor has a high Q of 20. The capacitance consists of a 200 μm compared to that of the pair of Gilbert mixers depends on the relativeNMOS-transistor and a 1 pF MIM-capacitor. The achieved phase- sizes of the RF transistors to the mixer quad transistors. The circuitnoise is -129 dBc/Hz at the 1 MHz offset from 1.6 GHz oscillating has a RF transconductor advantage noise advantage but a mixer quadfrequency. Table I shows the contribution of each noise source to transistor noise disadvantage [19]. Therefore, it is seen that whenthe total phase-noise. It is seen that bias circuitry in this design quad transistors are made small, the noise performance of the circuitcontributes 25% of the total noise. The raw high spectral purity relative to a pair of Gilbert mixers improves.achieved in this self-oscillating LNA-Mixer is due to the class-Coperation of the VCO, high VCO amplitude swing and the high A. Case Study: A Galileo RF Front-Endloaded Q offered by the proposed circuit. The circuit of Fig. 4 has been designed for Galileo application using 65 nm CMOS technology with an RF option. The LNA isC. Linearity biased with an current mirror. The bias circuit is isolated from a The linearity of the circuit is limited in two different points. At signal path using 20 kΩ resistors. The RF transistors M1 and M2 arethe RF (input) port, it is limited by the bias Vef f = Vgs−vt of the biased at the drain current of 0.5 mA each. Then the required sourcetransistors M1−2 and the voltage gain in the matching circuit [8] inductance to realize 50 Ω input impedance is about 2.5 nH. The input network includes the total parasitic parallel capacitance (300 IIP 3M1−2 IIP 3 = (14) fF) including the ESD protection diodes and bonding pad structure. Q2in Finally, in order to series resonate the input impedance at 1.575 GHz, the external gate inductors of 30 nH are used. The Q of these IV. T HE S ELF -O SCILLATING IQ RF F RONT-E ND external inductors is 30. The circuit is biased such that the current The LNA in direct conversion receivers must drive two mixers through the Q-VCO is 1 mA and the current through switching quadto produce quadrature outputs. Essentially, there are two options to is almost zero. The quad is voltage biased in the vicinity of theaccomplish this, either the use of two separate circuits presented in threshold voltage. The load resistance RL is 1 kΩ. The differential 8Fig. 2 or using a modified version of the circuit in Fig. 2 wherein nH inductor, which isolates the Q-VCO from the other circuitry hasthe LNA part of the circuit drives two Gilbert cell switching quads a Q of 20 and resonates out the parasitic capacitance at the 1.575as shown in Fig. 4 [19]. The two pairs on the left are switched GHz. The Q-VCO consists of a 8 nH differential inductor with a Q
  4. 4. Vdd Vdd V. C ONCLUSION A new circuit to realize self-oscillating LNA-Mixer has been Q+ Q- I- I+ proposed. Furthermore, using the proposed circuit it is possible I+ I- Q+ Q- to realize an entire single-stage RF receiver. The presented circuit achieves a high-performance with a low-power consumption. The Vtune Vtune future work includes the complete analysis of the presented circuit out1 out2 out3 out4 and possibly fabrication of the circuit. mp mp lo1 lo2 lo1 lo3 lo4 lo3 ACKNOWLEDGMENT s1 s2 The first author would like to thank Jenny and Antti Wihuri foundation for financial support. mp R EFERENCESrf1 rf2 [1] Antonio Liscidini, Andrea Mazzanti et.al , ”Single-Stage Low-Power Quadrature RF Receiver Front-End: The LMV Cell”, IEEE Journal of Solid-State Circuits, vol 41, No.12, December 2006. [2] Henrik Sjoland, Ali Karimi-Sanjaani and Asad A. Abidi , ”A merged CMOS LNA and Mixer for a WCDMA receiver”, IEEE Journal of Solid- Fig. 4. The Self-Oscillating IQ LNA-Mixer State Circuits, vol 38, No.6, June 2003. [3] To-Po Wang, Chia-Chi Chang et.al , ”A Low-Power Oscillator Mixer in 0.18-um CMOS Technology”, IEEE Transaction on Microwave Theory TABLE II and Techniques, vol. 54, no. 1, January 2006. T HE RF FRONT- END PERFORMANCE AND COMPARISON WITH STATE OF [4] Tero Koivisto and Esa Tiiliharju, ”A current reused oscillator-mixer for THE ART DS-UWB”, Norchip, 2009. [5] M.Camponeschi, A.Bevilacqua, P.Andreani, ”Analysis and design of a This work [1] [5] [21] low-power single-stage CMOS wireless receiver”, Norchip, 2009. Results sim. meas. meas. meas. [6] W. H.Sansen, R. G.Meyer , ”Distortion in Bipolar Variable-Gain Ampli- fiers”, IEEE Journal of Solid-State Circuits, vol 8, No.4, August 1973. Av [dB] 25 36 27 26 [7] Behzad Razavi, ”Design Considerations for Future RF Circuits”, IEEE NF (SSB) [dB] 4.4 4.8 13 5.7 International Symposium on Circuits and Systems (ISCAS), 2007. [8] Pete Sivonen, ”Design, Analysis, and Optimization of Radio Frequency IIP3 [dBm] -15 -19 n.a. -14.5 front-ends for integrated wireless transceivers”, PhD Thesis, Helsinki IIP2 [dBm] 41 n.a. n.a. 26 University of Technology, 2006. S11 [dB] ≤ −15 n.a. ≤ −10 ≤ −25 [9] Sang-Gug Lee, J. K.Choi , ”Current-reuse bleeding mixer”, Electronics Letters, April 2000. PN(1MHz) [dBc/Hz] -125 -104 n.a. -126 [10] Jouni Kaukovuori, ”CMOS Radio Frequency Circuits for Short-Range IDD [mA] 1 4.5 1.3 15.3 Direct-Conversion Receivers”, PhD Thesis, Helsinki University of Tech- VDD [V ] 1.2 1.2 1 2.7 nology, 2008. [11] Jinsung Park, Chang-Ho Lee et.al , ”Design and analysis of low flicker- Technology 0.065µm CMOS 0.13µm CMOS 0.09µm CMOS 0.35µm BiCMOS noise CMOS mixers for direct-conversion receivers”, IEEE Transaction on Microwave Theory and Techniques, vol. 54, no. 12, December 2006. [12] Pietro Andreani, Ali Fard, ”A 2.3 GHz LC-tank CMOS VCO with optimal phase noise performance”, IEEE Int.Solid-State Circuits (ISSCC) Dig.Tech.Papers, 2006. [13] Andrea Mazzanti and Pietro Andreani , ”Class-C CMOS VCOs, with aof 20 and varactors from the RF library. The outputs of the Q-VCO general result on phase noise”, IEEE Journal of Solid-State Circuits, volare coupled to the switching quads through 4 pF coupling MIM- 43, No.12, December 2008. [14] Kuang-Wei Cheng, Karthik Natarajan and David J. Allstot, ”A currentcapacitors. The quadrature coupling transistors are voltage biased reuse quadrature GPS receiver in 0.13 µ CMOS”, IEEE Journal of Solid-through large resistors and the coupling between the Q-VCO and State Circuits, vol 45, No.3, March 2010.coupling devices has been accomplished using 4 pF capacitors. The [15] Eyad Abou-Allam, John J. Nisbet and Michael C. Maliepaard , ”Low-length of the all transistors is 0.13 μm. The width of the LNA Voltage 1.9-GHz front-end receiver in 0.5-µm CMOS technology”, IEEEtransistors is 50 μm. The widths of the Q-VCO active transistors Journal of Solid-State Circuits, vol 36, No.10, October 2001. [16] Hooman Darabi and Asad A. Abidi, ”Noise in RF-CMOS Mixers: Ais 20 μm and the width of the coupling transistors is 20 μm. The simple physical model”, IEEE Journal of Solid-State Circuits, vol 35,width of the quad transistors are 20 μm. The simulated performance No.1, January 2000.metrics for the proposed circuit are shown in Table 2. The voltage [17] Marc Tiebout , ”Low-Power Low-Phase-Noise Differentially Tunedgain of the circuit is 25 dB with an IIP3 value of -15 dBm. The Quadrature VCO Design in Standard CMOS ”, IEEE Journal of Solid- State Circuits, vol 36, No.7, July 2001.gain can be controlled by varying the size of the load resistance. The [18] J. Rael and A. Abidi , ”Physical processes of phase noise in differentialsingle-sideband noise figure (SSB NF) NF is 4.4 dB at the 4 MHz LC-oscillators ”, IEEE Custom Integrated Circuits Conference, May 2000.IF frequency. The flicker noise corner is 170 kHz. The IIP2 is +41 [19] J. Harvey and R. Harjani , ”An integrated quadrature mixer withdBm. The phase-noise of the Q-VCO is -125 dBc/Hz at the 1 MHz improved image rejection at low voltage ”, 14th International Conferenceoffset from the oscillation frequency. The linear tuning range of the on VLSI Design, January 2001. [20] Kuang-Wei Cheng, Karthik Natarajan, David Allstot, ”A 7.2mWQ-VCO is 10 %. The RF front-end scattering parameter S11 is -25 Quadrature GPS Receiver in 0.13 µm CMOS”, IEEE Int.Solid-StatedB at the exact operating frequency. The circuit draws 1 mA from a Circuits (ISSCC) Dig.Tech.Papers, 2009.1.2 V supply. The isolation properties of the circuit has been studied [21] P.Sivonen, S.Kangasmaa, A.Parssinen, ”A SiGe RF front-end with on-using Monte Carlo simulations with 200 iterations. The worst case chip VCO for a GPS receiver”, IEEE European Solid-State Circuits Conference (ESSCIRC), 2002.isolation from the RF input transistor to the inductor center tap is 40dB.

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