DRAKE et al.: RESONANT CLOCKING USING DISTRIBUTED PARASITIC CAPACITANCE 1521
Fig. 1. Buffer-driven clock network and resonant clock network diagrams.
(a) Buffer-driven clock distribution. (b) Reduced clock tree.
in that all circuit power is recycled. Adiabatic logic beneﬁts
from slow transition times making it impractical for high-per-
formance logic, although modiﬁed adiabatic circuits have been
developed that run above 100 MHz , . Another resonant-
clock generation technique establishes a standing or traveling
wave using the transmission-line characteristics of the clock
lines; this approach has yet to demonstrate a power advantage
over established clock-distribution techniques –.
The resonant-clock scheme presented here addresses the
power dissipation in the local clock directly by using the para-
sitic capacitance inherent in the local clock distribution as the
capacitor in an LC tank. All clock buffers and their associated
capacitance are removed and the clock energy is resonated
between integrated inductors and the local clock capacitance.
Unlike adiabatic circuits which power the logic from the clock
and rely on slow edges, this resonant-clock scheme has the
potential to run at frequencies used in modern microprocessors
since only the gate capacitance is driven by the clock.
By incorporating the capacitance as part of the oscillator,
clock generation, and distribution are designed concurrently and
Fig. 2. Ideal resonant clock-generation and distribution.
the oscillator naturally selects the most efﬁcient frequency; un-
like buffer driven resonant clock networks such as in  where
the natural frequency of the network has to be tuned to the clock
frequency. Unfortunately, clock-gating can only be achieved in
the proposed scheme by shutting down the oscillator, which is
possible if the startup time of the oscillator can be tolerated.
The resonant-clock generation scheme presented here can be
used to replace entire clock systems for small designs or the
quadrant clocks in larger designs. Thanks to improving inte-
grated inductors and copper metallization in advanced semi-
conductor technologies, the quality factor of the resonant cir-
cuit is sufﬁcient to effect clock power reduction over ungated
buffer-driven local clocking techniques. The next section will
review the theory behind distributed-capacitance resonant-clock
generation. Following that, a prototype resonant clock, built in
IBM’s 0.13- m partially depleted SOI (PD-SOI)  will be
II. RESONANT CLOCKING THEORY
The main advantage of resonant clocking is a reduction of
clock power, but the procedure introduces challenges for the
designer such as jitter and skew management and nonlinear load
capacitance. Each of these will now be examined.
The power reduction can only occur if less static power is dis-
sipated in the parasitic resistance of the resonant clock than is
dissipated switching the buffers and local clock capacitance of
a buffer-driven clock. To form the resonant clock, integrated in-
ductors are placed in parallel with the clock load, , creating
an RLC circuit as shown in Fig. 2. At resonance, the impedance
of the parallel elements is inﬁnite. Power is only dissipated
in the parasitic resistance, , which arises from the resistive
elements of the inductors and the distributed capacitance. The
clock generated by the resonant circuit is a sinusoid of the form
whose magnitude, , depends on the
magnitude of . The resonant frequency, , is determined by
. To allow comparisons to the buffer-driven clock,
is assumed to be , providing a clock that swings be-
tween 0 V and . The average power dissipation in the RLC
circuit at resonance is
Given that the quality factor, , of a parallel RLC circuit is
, that the clock load is , and that the clock fre-
1522 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 9, SEPTEMBER 2004
quency is , then the ratio of power dissipated in the proposed
resonant clock versus a buffer-driven clock, from (2) and (3), is
Thus, a resonant-clock distribution with a greater than
will use less power than a buffer-driven clock network with
a stage gain of 3. The quality of integrated inductors has im-
proved signiﬁcantly with each technology generation; inductor
values greater than 15 were reported for the target technology
. The achievable power reduction in the resonant clock de-
pends on how low the resistance of the clock distribution can be
Another advantage of the proposed resonant clock is the
ability to control the maximum voltage of the output clock
simply by varying . This feature can be used to overdrive
the clock signal for faster rise and fall times at the logic gates
at the expense of some extra power, but without having to
generate and propagate higher clock harmonics. To do this with
buffer-driven clocks, the output resistance of the drivers must
be made smaller by increasing the driver size which increases
the capacitance in the network. Unfortunately, the clock-distri-
bution network can nullify such efforts by ﬁltering the higher
clock harmonics, which is why clocks on high-performance
processors become more sinusoidal with each generation. Care
must be taken when overdriving the resonant clock to prevent
the clock waveform from clipping so that it is no longer sinu-
soidal. Clipping increases the power dissipation in the resonant
circuit and reduces its efﬁciency .
B. Skew Management
Since the clock network serves as the capacitance setting the
clock frequency, it must be made small enough to avoid trans-
mission-line effects and to keep skew manageable. Electromag-
netic signals propagate at a speed given by
or roughly 150 m/ps in wires in an SiO insulator. Thus, for
a 1-GHz clock, the clock sinks must extend less than 15 mm
from the clock source to meet a skew requirement of 10%. The
actual propagation time will be slower than predicted by (5)
due to the loading and branching in the clock network, so some
margin must be included in the design. A block of logic 7 cells
by 64 cells, where the cells are 16 16 wire tracks and contain
pass-gate clock loads, was simulated in IBM’s 0.13- m SOI
process with a 3-GHz clock. Using distributed RLC pi models,
the skew from top left to bottom right of the network was 10 ps,
which is longer than the 6.9 ps predicted by (5). Fortunately, the
skew requirements are stringent enough to ensure that the clock
network does not behave like a transmission line. Since the rise
time of a sinusoidal clock is 50% of its period and skew targets
are less than 10% of the clock period, a clock network that meets
skew requirements will never be long enough for reﬂections to
C. Quality Factor
There are a number of deﬁnitions for quality factor which are
Maximum Energy Stored
Energy Dissipated per Cycle
Maximum Energy Stored
Average Power Dissipation
where is the resonant frequency and is the bandwidth, the
difference between the half-power frequency above and below
. Quality factors of individual components are more
easily characterized than a resonant circuit, so it is useful to
be able to relate the quality factor of the components to that of
the overall circuit. Real inductors and capacitors contain lossy
elements and have somewhat complicated models when all
physical effects are taken into consideration. However, if the
RLC resonant frequency is well below the self resonance of the
circuit elements, then at resonance the inductor and capacitor
are inductive and capacitive with some real lossy component.
The quality factor of a nonideal inductor  in parallel form
is approximated from (6) by
where is the parasitic resistance of the inductor expressed
as a parallel resistance at resonance. The quality factor for a
nonideal capacitor  in parallel form is approximated by
where is the equivalent parallel resistance in the capac-
itor. Solving (7) and (8) for and , substituting into the
quality factor of a parallel RLC tank, ,
and performing some algebra provides the tank quality factor in
terms of its component’s quality factors:
From (9) it is apparent that a low-quality distributed capac-
itor will limit the quality of the resonant circuit. To improve the
quality factor, the clock resistance must be kept to a minimum by
utilizing techniques already needed for reducing skew in stan-
dard clock-distribution methods such as clock grids, fat wires,
and multiple vias. Unlike a standard clock distribution where
reduced capacitance is a must, the quality factor of the resonant
clock is improved by adding extra capacitance to the distribu-
tion network. In most integrated oscillators, the quality of the
inductor limits the quality factor, but the distributed nature of
the capacitor adds enough parasitic resistance to the capacitance
to limit the quality factor.
D. Nonlinear Capacitance
The most challenging part of the resonant clock is character-
izing the distributed, parasitic capacitor. If the clock network is
designed to meet skew requirements and avoid transmission-line
effects, its parasitic capacitance acts like a lumped capacitor, but
DRAKE et al.: RESONANT CLOCKING USING DISTRIBUTED PARASITIC CAPACITANCE 1523
Fig. 3. Equivalent negative-resistance oscillator circuit.
Fig. 4. Master–slave D-ﬂip-ﬂop.
with a time-varying characteristic. Fig. 3 shows an equivalent
model of a negative-resistance oscillator used for the resonant
clock. The time-varying capacitance is represented as a ﬁxed
capacitance, , in parallel with a periodically varying capac-
itance, , and time-dependent noise capacitance, . If
designed correctly, the negative resistance and parasitic resis-
tance cancel and the circuit behaves like an ideal LC tank which
has a transfer function of
and a natural frequency of . Unfortunately, the
capacitance in the distributed network is not constant, but a func-
tion of two independent voltages applied to the gate and drain
of the transistors. The gate voltage comes from the clock signal
and the drain voltage, which is pseudo-random, comes from the
data ﬂowing through the logic. Equation (10) is not an accurate
model of the transfer function. In Fig. 3, the time-varying ca-
pacitance associated with the gate voltage is the periodic capac-
itance, , and the time-varying capacitance associated with
the data signals is the noise capacitance, . The common
ﬂip-ﬂop design in Fig. 4 is used for the clock loads in this study.
Fig. 5 shows the ﬂip-ﬂop’s simulated input gate capacitance
variation for sinusoidal and square-wave gate voltages, ignoring
the effect of the data signals. The change in gate capacitance
is periodic with the input waveform and since the capacitance
changes are driven by the clock at steady state, a stable oscilla-
tion frequency will be reached, as will be explained later.
The noise capacitance, , is more difﬁcult to understand
because it results from logic signals travelling through the
clocked logic and will be pseudo-random in nature. The logic
is driven by sources independent of the clock and will cause
Fig. 5. Gate capacitance variation with input waveform.
some amount of mixing in the clock signal. An intuitive under-
standing of this effect is obtained from the KCL node equation
for the circuit in Fig. 3
There is no analytical solution to (11), but some insight into
the solution can be gleaned from its Fourier transform, which is
The last two convolution terms on the right-hand side of (12)
are not in (10) and result directly from the two time-varying
capacitances. In the steady-state solution for (12),
and are co-periodic so they will not cause phase noise.
The noise capacitance, , and data voltages, ,
are random and will modulate the gate voltage, causing jitter.
The difﬁculty is in analyzing the magnitude of this effect in a
The oscillator behaves like a frequency modulation circuit
where the data voltage acts as a modulating signal . The
instantaneous frequency of the oscillator is given by
1524 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 9, SEPTEMBER 2004
Thus, the frequency is composed of a fundamental frequency,
, divided by a normalized time-varying capacitance. If the
time-varying capacitance, , has a small maximum ampli-
tude variation, , then a reasonable approximation can be
made as follows
Equation (14) is valid when the change in capacitance is small
relative to the average capacitance value. Plugging (14) into (13)
gives an instantaneous frequency, , of
If the time-varying capacitance is then
the instantaneous frequency is
By deﬁning , then the phase shift
over time is given by
Equations (16) and (17) can be used to analyze the effects of
the time varying load capacitance on the oscillation frequency
and on the jitter of the oscillator . The most important ob-
servation is that the frequency deviation is small if the change
in capacitance is also small. If then the frequency
will only deviate by 5%. Simulations show that the clock load
capacitance of the D-ﬂip-ﬂop changes by 5% due to changes in
the data-ﬂow. If to of the clock network capacitance
is in the gates of the latches then the maximum change in the
clock network capacitance, which occurs when the data in all
the latches changes at the same time and in the same direction,
is between 1.7% and 2.5%. This results in a change in frequency
of 0.83% to 1.25%. Simulations of a simpliﬁed resonant clock
network showed less than 15-ps period jitter for a 2-GHz clock.
This analysis does not account for capacitive coupling between
the gate and the drain which will depend strongly on the edge
rate of the logic and which may be signiﬁcant if data movement
is the same through a majority of the logic gates.
E. Miscellaneous Concerns
Some other concerns for resonant clocking include the area
penalty associated with the integrated inductors, how to gate the
clock to reduce power when a functional unit is not needed, and
how to synchronize clock domains. Each of these will be brieﬂy
addressed. The area penalty can only be reduced by using res-
onant clocks that require the minimum number and size of in-
ductors. Fortunately, since the quality of the parasitic capacitor
limits the quality of the resonant clock, some tradeoffs can be
Fig. 6. Modiﬁed test-chip block diagram.
made between inductor area and quality. As for clock gating, it
is a challenge since the local clock buffers have been removed.
Turning off the clock is an option but wastes time and power
waiting for the clock to settle. Finally, to use the presented res-
onant clock in a large design that cannot be covered by a single
clock domain due to skew requirements, some tuning mecha-
nism would have to be incorporated to synchronize multiple
clock domains , or a hand-shaking system used between dif-
ferent clock networks.
III. RESONANT CLOCKING TEST CHIP
A test chip was fabricated in IBM’s 0.13- m RF PD-SOI
CMOS process  to analyze the frequency, power dissipa-
tion, and quality of the proposed resonant clock as compared to
a buffer-driven clock network. A block diagram of the test chip
is shown in Fig. 6 and a microphotograph of the test chip, minus
the external inductors, is shown in Fig. 7. The load of the local
distribution consists of three 8 64 scan-chains connected by
a clock grid. Each scan-chain has eight rows, for a total of 24
rows, where each row is composed of 64 D-ﬂip-ﬂops. The ex-
perimental clock load represents the clock load of a block of
static CMOS logic with 24 latch stages, as may be found in the
functional units of a 64-bit pipelined microprocessor. The clock
distribution is laid out differentially in metal-2 over each cell
with a parallel grid on metal-4. The logic gates are powered by
the supply voltage .
A negative-resistance oscillator was designed as the resonant
clock source with integrated inductors similar to those reported
in . The capacitance used to set the resonant frequency of
the oscillator is the parasitic capacitance of the clock network
and was estimated, based on wire models and manual extrac-
tion, since an automatic extraction deck was not available, to be
about 21 pF per clock phase. The parasitic resistance and induc-
tance in the clock wires joining the VCO with its clock load were
underestimated and so the NFETs in the VCO are undersized.
The wires have 3- resistance and about 0.8 nH which is enough
DRAKE et al.: RESONANT CLOCKING USING DISTRIBUTED PARASITIC CAPACITANCE 1525
Fig. 7. Microphotograph of resonant clock test macro.
to keep the VCO from starting. To get the chip to start up, the
integrated inductors were cut out and the internal clock-node
bonded to connect to off-chip inductors. The total inductance
consists of the bond-wire inductance, , and the external
inductor, . By using larger, off-chip inductors, the resonant
frequency of the oscillator was lowered to a value where the
transconductance of the crosscoupled NFETs could cancel the
parasitic resistance and maintain the oscillation. Using off-chip
inductance changes the resonant-clock experiment because the
inductance is no longer integrated and the clock frequency is
much lower than on a high-performance VLSI circuit. Never-
theless, the capacitance that sets the oscillation frequency is still
the local clock capacitance and its affect on clock stability due
to data signals can still be measured.
To simulate a conventional, buffer-driven clock distribution,
an 11-stage ring-oscillator and associated 7-stage buffer horn,
with a stage gain of 3, were added for power comparisons.
Both the resonant clock and the ring oscillator drive the same
local clock network but the ring oscillator is tri-stated, so
only one clock driver has access to the clock grid at a time.
The ring-oscillator characteristics were measured after cutting
out the inductors with a laser. Simulations were performed to
ensure that the ring-oscillator frequency was close to 2 GHz
with 10% edge rates at 1.2 V. Both clock phases are divided
down by 64 and output for frequency measurements. There
are three power-supply domains on the chip that separate the
oscillator, scan-chain, and ring-oscillator power.
IV. TEST RESULTS
The test chip operates in two modes for testing. In the reso-
nant mode, the ring oscillator is disabled and the resonant clock
controls the clock network. In the ring-oscillator mode, the res-
onant clock is disconnected from the clock grid using laser trim-
ming and the ring oscillator drives the clock network without the
inductors. Functionality of the latches, and by correlation the
Fig. 8. Operating frequency of the resonant clock and the ring-oscillator.
clock, was determined by monitoring the divided clock output.
The clock was also monitored at the junction of and .
Fig. 8 shows the measured clock frequency of the reso-
nant clock and the ring-oscillator as a function of the supply
voltage on the logic. The resonant-clock frequency varies from
147 MHz when the VCO is driven by 0.4 V to 112 MHz when
driven by 0.6 V, which is consistent with simulations. The
ring-oscillator frequency, on the other hand, increases rapidly
with power supply because of increasing current drive in the
individual delay elements. Measurements were taken with ex-
ternal inductors ranging from a simple wire to a 420-nH air-core
inductor. Measurements indicate that the clock load is between
38 and 45 pF in each phase of the clock, the bond-wire has an
inductance of 15.9 nH, and the external wire has an inductance
of 17.1 nH. The clock frequency measured is within 12.5%
of the predicted value. For all remaining measurements, the
external inductor consisted only of a simple wire between the
bond-pad and the power supply.
Fig. 9 shows the eye diagram of the dual-phase clock mea-
sured at the inductor bond-pin. The low voltage swing and dis-
tortion in the waveform occur, according to simulations, because
the eye diagram was measured between a voltage divider com-
posed of the external and the bond-wire inductance, not at the
clock gates. Simulations show a cleaner signal at the clock gates,
and the logic is functional in measurements, but the actual shape
of the clock signal cannot be veriﬁed in this test chip.
Fig. 10 shows the power dissipated in the ring-oscil-
lator-driven clock versus the resonant clock. The ring-oscillator
power has also been scaled by frequency and supply voltage
to compare the two techniques. Three things complicate this
comparison. First, the buffer-horn was not optimized and may
dissipate more power than a well designed clock distribution
network. Second, the tri-state inverters add an extra load to the
resonant clock that would not normally be present. Third, this
comparison is not exact because the actual amplitude of the
resonant clock voltage could not be measured. Knowing the
amplitude of the clock signal is necessary for an accurate com-
parison between resonant clocking and buffer-driven clocking
1526 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 9, SEPTEMBER 2004
Fig. 9. Dual-phase clock eye-diagram.
In the test chip, the clock voltage swing can be measured at
the point where the bond-wire inductor (16 nH) and the external
wire (17 nH), meet. Measurements taken at that tapped point
show the clock voltage swing at the logic gates is from
V to , depending on the bias on the resonant clock. The
clock voltage at the clock gates on chip is higher than that mea-
sured at the tapped point and its magnitude is determined by
the ratio of inductance in the bond-wire to the inductance in the
external wire. Simulations of the clock network, under the bias
conditions being tested, show that the on-chip clock voltage is
300 mV higher than the clock voltage at the pad where the two
inductors meet. The simulations predict a power dissipation of
2.8 mW at this bias point. Based on these simulations and the
measurements taken, the on-chip resonant clock swing at the
most efﬁcient point measured— of 0.67 V and of
0.43 V—is between 0.63 and 0.93 V. The resonant-clock power,
measured at V with a frequency of 147 MHz,
is 2.06 mW. The ring-oscillator clock power was measured as
7.72 mW with a clock frequency of 360 MHz at V.
Scaling the measured ring-oscillator clock power to 147 MHz at
0.63 V and 147 MHz at 0.93 V yields a scaled power of 3.15 mW
and 6.9 mW. The resonant clock thus dissipates between 35%
and 70% less power than the buffer-driven clock at the same fre-
quency and clock amplitude. The actual power savings is most
likely near the lower part of this range. From (4), the quality
factor of the resonant circuit is 2.4 if the power savings is 35%
and 5.3 if the power savings is 70%, which is quite low. For an
inductor quality of 15, this means the capacitance has a quality
between 2.8 and 8.2 from (9). Again, the lower number is prob-
ably most accurate based on simulations.
It is important to note that the resonant clock has a wide swing
in power dissipation without increasing clock frequency. This is
due to the resonant clock leaving its sinusoidal operation and
generating a waveform that looks like a half sine wave . For
the resonant clock to make sense from a power perspective, it
must be designed efﬁciently and kept in its sinusoidal operating
Fig. 11 shows the period jitter measurements of the output
of the divider as the scan-chain input frequency was increased
from DC to one-half of the resonant frequency. Period jitter is
Fig. 10. Power dissipation of the resonant clock and the ring oscillator.
Fig. 11. Period jitter of the resonant clock and the ring oscillator as the
frequency of data passing through the scan-chain is swept.
the square-root of the variance of the width of the clock period.
The measurements were made using an Agilent Inﬁniium Oscil-
loscope using the method described in . Since the clock load
is a scan-chain, the state of all of the ﬂip-ﬂops changes each time
the input changes and in the same direction, maximizing the ca-
pacitance change in the clock network. The resonant clock jitter
was measured with the oscillator running at 2.0 MHz and the
ring-oscillator jitter was measured with the ring oscillator run-
ning at 1.96 MHz. The output clock frequency is the internal
clock divided by 64. The on-chip period jitter can be approx-
imated by dividing the period jitter of the output clock by the
square root of the divider , or 8 in this case, although this
ignores the jitter contribution of the divider.
The ring-oscillator jitter is higher than the resonant clock
jitter due to well studied differences between and delay-
based oscillators. Jitter in the 2-MHz divided output clock is a
relatively ﬂat 400 ps until the data rate approaches one-half of
the clock frequency where jitter rapidly rises to 910 ps, or 0.18%
of the clock period. The maximum jitter of the internal clock,
measured at the inductor bond-pad, was 55 ps, or 0.68% of the
124-MHz clock. The closer the data frequency is to half the
resonant frequency, the worse the jitter becomes, which is ex-
pected because faster data rates mean more capacitive coupling
DRAKE et al.: RESONANT CLOCKING USING DISTRIBUTED PARASITIC CAPACITANCE 1527
and more data induced capacitance changes. Unfortunately, a
logic chip will have random data patterns, not deterministic pat-
terns as measured here. To approximate a more real scenario,
the data frequency was varied randomly between 1 and 60 MHz
for several minutes. The measured jitter was 555 ps, or 0.11%.
At higher frequencies, the edge rates are sharper, so capacitive
coupling should increase the jitter as the gigahertz range is ap-
proached, but since these changes are mostly local in the dis-
tributed clock network, they should not be signiﬁcant.
The test macro demonstrates that a stable resonant clock can
be implemented using the inherent parasitic capacitance of the
local clock network in an LC tank. Both a resonant clock using
the local gate capacitance and a ring-oscillator-driven buffer-
horn clock distribution were implemented. A stable sinusoidal
clock between 112 and 147 MHz, depending on biasing, was
measured using a straight wire for the external inductance. An
analysis of the voltage-varying gate capacitance shows that data
ﬂowing in the clock network should change the clock frequency
by less than 1.25%. A maximum period jitter of 0.68% was mea-
sured when the scan-chain data frequency approached one-half
of the clock frequency. Power comparisons indicate that the res-
onant clock dissipates around 35% less power than the buffer-
driven clock with an estimated quality factor between 2.4 and
5.3. Since the off-chip inductors used in the measurements have
quality factors between 15 and 30, the of the parasitic capac-
itance is nearly the quality of the tank. On-chip inductors in this
technology were measured with quality factors above 15 as well
, so moving the inductance on-chip should not adversely af-
fect the power savings. The main disadvantage of scaling the
clock into the multigigahertz range is the increase in wire re-
sistance due to skin effect which will decrease the already low
quality of the distributed capacitor. Some things can be done
to improve the quality of the parasitic capacitance. The clock
signal was partially routed in poly-silicon within the D-ﬂip-ﬂop,
so removing poly routing and using wider wires in lower metal
layers would improve the quality of the capacitor; clock wire
widths in general need to be wider to handle the current needed
at higher frequencies.
Since the capacitor quality is the limiting factor, some loss
of in the inductor can be tolerated to save area. Moving the
inductor close to the logic circuits and using a multiturn inductor
instead of a single-turn inductor would save area at the expense
of some inductor quality. A balanced VCO with crosscoupled
PFETs as well as NFETs uses only one inductor instead of two
for even more area savings. A second generation of the resonant
clock designed to operate in the multigigahertz range is being
developed that improves the clock load and area using these
The authors acknowledge the contributions made by R. Mon-
toye and U. Ghoshal of IBM’s Austin Research Laboratory and
the help with the technology given by N. Zamdmer, M. Sherony,
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Alan J. Drake (S’99) received the B.S. degree
in electrical engineering from the University of
Arizona, Tucson, in 1997 and the M.S. degree
in electrical engineering from the University of
Michigan, Ann Arbor, MI, in 2000. Currently, he is
working toward the Ph.D. degree at the University
His research interests include low-power VLSI,
resonant clock generation and distribution, and SOI
technology. In March, 2004, he joined the IBM
Austin Research Laboratory where he is conducting
research on clock distribution and high-performance processor circuit design.
Kevin J. Nowka (S’84–M’85) received the B.S. de-
gree in computer engineering from Iowa State Uni-
versity, Ames, in 1986 and the M.S. and Ph.D. de-
grees in electrical engineering from Stanford Univer-
sity, Stanford, CA, in 1988 and 1995, respectively.
He joined the IBM Austin Research Laboratory
in 1996 where he has conducted research on CMOS
VLSI circuits for two 1-GHz microprocessors and
for a low-power embedded PowerPC processor. He
currently manages the Exploratory VLSI Design
Department of the IBM Austin Research Laboratory.
He holds 35 patents related to microprocessor design.
1528 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 9, SEPTEMBER 2004
Tuyet Y. Nguyen was born in Vietnam.
She joined IBM in 1987. She has been involved in
process support, specializing in analyzing device fail-
ures resulting from manufacturing process and layout
design issues. Her current focus is VLSI mask design
for high speed analog and digital VLSI designs.
Jeffrey L. Burns received the B.S. degree in en-
gineering from the University of California, Los
Angeles, and the M.S. and Ph.D. degrees in electrical
engineering from the University of California at
In October 1988, he joined the IBM T. J. Watson
Research Center as a Research Staff Member, where
he worked in the areas of layout compaction, layout
synthesis for control logic, CAD system architecture,
and microprocessor design. In 1996, he joined the
IBM Austin Research Laboratory, Austin, TX, where
he worked initially on high-frequency microprocessor design and design-tools
strategy. From 1999 to 2003, he managed the Exploratory VLSI Design Depart-
ment of the Austin Research Laboratory, working in the areas of high-end mi-
croprocessors, ultra-low-power embedded processors, and high-bandwidth data
communications. Since mid 2003, he has been on the IBM Research Technical
Strategy staff, in Yorktown Heights, NY, where his main responsibility has been
to produce IBM Research’s long-term IT industry outlook.
Dr. Burns received an IBM Outstanding Technical Achievement Award in
1997 for his microprocessor tools and design work for IBM’s S/390 products,
and an IBM Research Division Award for his work on IBM’s 1.0-GHz PowerPC
prototype disclosed in 1998.
Richard B. Brown (S’74–M’76–SM’91) received
the B.S. and M.S. degrees in electrical engineering
from Brigham Young University, Provo, UT, in
1976, and the Ph.D. degree in electrical engineering
(solid-state) from the University of Utah, Salt Lake
City, in 1985.
From 1976 to 1981, he worked in computer design
as Vice-President of Engineering at Holman Indus-
tries, Oakdale, CA, and then as Manager of Com-
puter Development at Cardinal Industries, Webb City,
MO. He joined the faculty of the Department of Elec-
trical Engineering and Computer Science, University of Michigan, Ann Arbor,
in 1985. He has conducted major research projects in the areas of solid-state
sensors, mixed-signal circuits, GaAs and silicon-on-insulator circuits, and high
performance and low power microprocessors. He served as Associate Chair of
Electrical Engineering for four years and as Interim Chair of Electrical Engi-
neering and Computer Science for two years at the University of Michigan. He
became Dean of Engineering at the University of Utah in July 2004.
Prof. Brown serves as Chairman of the NSF MOSIS Advisory Council for Ed-
ucation. He was Chair of the 1997 Conference on Advanced Research in VLSI
and the 2001 Microelectronic System Education Conference. He has served as
Guest Editor of the IEEE JOURNAL OF SOLID-STATE CIRCUITS and Proceedings
of the IEEE, and as associate editor of IEEE TRANSACTIONS ON VLSI SYSTEMS.
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