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    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512High Performance Predictive Direct Torque Control Of Induction Motor Drives System A. Alwadie Electrical Engineering Department, Faculty of Engineering, Najran University, KSAABSTRACT This paper presents a practical and DSP board) are implemented and configured.implementation for direct torque control of In addition, Implementation of Digital Signalinduction motor drive. Control system Processor (DSP1102) on DTC has been presented.experiment is proposed using Digital Signal Through the implementation, it will be shown thatProcessor. This control scheme directly the high performance of the control system isdetermines the switching states of the inverter achieved. Computer simulation validates theand gives optimal characteristics for stator flux experimental results of this control method.and torque control. A simple switching patternfor direct torque control algorithm for induction 2. DIRECT TORQUE CONTROL MODELINGmotor drives has been presented. The main A detailed block diagram of the directresults with the simple approach are greatly torque controlled induction motor drive is shown inreduced torque pulsations, constant and Fig.(1). The figure shows a quick response of thecontrolled switching frequency. Results show that torque control system in which the instantaneousthe simulation meets the experimental results. speed (s) of the primary flux linkage vector (ys)of the motor is controlled in such a way that it allowsKeywords - Digital Signal Processor (DSP) - the developed torque (Td) to agree with the torqueInduction Motor (IM) - Direct Torque Control command (T*) within a certain error. ys and Td are(DTC) - Space Vector Modulation. calculated respectively by the following relations [4]:1. INTRODUCTION Nowadays, field oriented control (FOC)    (v s  R s i s ) dt (1) sand direct torque control (DTC) is amongst the most p i  i  3 T   ds qs qs ds important control methods and is extensively used in (2)induction motor drives. High performance induction d 2  motor drives without position sensors are desired forindustrial applications. From this point, DTC is a Where,suitable technique for squirrel-cage induction Vs the instantaneous space primary voltage vector.motors. In this technique, the stator flux is is the instantaneous space primary currentimplemented by measuring the stator voltage and vector.currents in a similar way that is done in direct FOC Rs the primary winding resistance.technique. On the other hand, the correct choice of yds, yqs d-axis and q-axis stator flux.the inverter voltage vector leads to direct and ids, iqs d-axis and q-axis stator current.decoupled control of motor flux and torque.According to this novel concept of decoupling DTC From equation (1), it is clear that ys can beis seen to be different from FOC and it has satisfied directly controlled by vs because of small statorsome advantages like simplicity and quick torque winding resistance drop. It is generally known thatresponse [1:3]. there are eight voltages made by a three-phase In this paper, a control method is proposed inverter. By proper selecting of these vectors boththat can provide transient performance similar to the amplitude and speed of ys are controlled toDTC with reduced torque pulsations at steady state. regulate the instantaneous value of Td as it will beThe inverter switching pattern is determined over shown later.constant switching period and hence constantswitching frequency is obtained. The torque and flux 2.1 Control system configurationcontroller design is investigated using a sliding For the purpose of control design, it ismode controller for direct torque control. Moreover, necessary to know a dynamic model of the inductionthe complete power circuits such as (Power supply, motor, which incorporates all dynamic effectsRectifier, pulse width modulation (PWM) inverter, occurring during steady state and transient operation.power interface, current sensors, voltage sensors, The state equations of the induction motor are given by [5]: 501 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512 * Speed  - + controller Injected signal Torque Comparator * + T Id, iq Calculation Td - - + Voltage vector of torque selection tables Calculation of s 6 1 2 , sd sq 3 5 4 Calculation of  - s stator flux Vd, vq * + s Sa Flux d-q transformation of Ia, Ib Comparator Sb currents and Sc voltages Va, Vb IM Ed 3phase-supply Incremental encoder Inverter Rectifier Fig.(1) Schematic diagram of direct torque control system dids   1 Vds  dt   Ls  di    Lm mutual inductance per phase referred to the 1 qs   Vqs  stator. dt    Ls  S total equivalent leakage factor of the motor. didr   Lm + dt   Vds  Ls Lr  L2  di      1  m   qr    Lm   Ls Lr   dt   L L qs  V   s r  Ls stator winding inductance per phase(Ls =Lls+Lm). Rs Lm p r 2 Rs Lm Lm p r   Lr rotor winding inductance per phase(Lr= 2 Ls  Ls Lr Ls Lr Lr   Llr+Lm). Lm p r Rs L p r Lm Rs  ids  L L   m   Ls Lr Ls Lr  iqs  The instantaneous space vectors of the s r   PWM inverter are regarded as discrete values, and Rs Lm Lm p r R  i   r  r   dr  the analysis using the instantaneous vectors is Ls Lr Lr Lr   iqr  L p   suitable for investigating the dynamic behavior of Lm Rs r Rr  the motor. To treat three phase quantities as a whole, m r   Lr Ls Lr  Lr   the three phase machine voltages and currents are represented by an instantaneous space voltage vector (3) (vs) and an instantaneous space current vector (i s), where, respectively, as [6]: Lls stator winding leakage inductance per phase. 2   j 2 j 4  (4) v  s v v e 3 v e 3  a c Llr rotor winding leakage inductance per phase. 3  b    502 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512  2  j 2 j 4  (5) is  ia  i e 3  ic e 3  3  b   where, va, vb and vc are the instantaneous values ofthe primary line-to-neutral voltages and ia, ib and icare the instantaneous values of the primary line-to-neutral currents. a Coupling Ed sa s sc n DC b Stat b c Gen. or Fig.(2 ) Schematic diagram of PWM inverter fed induction motor drive.Schematic diagram of the PWM fed induction motor d-axis V1(1,1,0) V2 (0,1,0)drive is shown in Fig.(2), where Sa, Sb, and Sc are theswitching functions with the value of 1 when theswitch is set to positive voltage or 0 when the switchis set to negative voltage. Considering the combinations of the status V6(1,0,0) V7(1,1,1) V8 (0,0,0) V3(0,1,1)of switches, the inverter has eight conduction q-axismodes[7,8]. By using these switching functions theprimary space voltage vector can be expressed as:  2 4 j    v s Sa , Sb , Sc  2 3 Ed  Sa  Sb e 3  Sc e 3  (6)  j  V5(1,0,1) V4(0,0,1) where, Ed is the dc link voltage of the Fig.(3) Representation of primary space voltage inverter. vectors. According to the combinations of switching modes, the primary space voltage vectors vs(Sa, Sb, Sc) is 2.2 DTC STATE VARIABLE SELECTION specified for eight kinds of vectors vs(1,1,1), (0,0,0) The direct torque control is achieved using and the others are the space nonzero active voltage three state variables [9]. The first variable is the vectors. i.e.(1,0,0),………..(0,1,1), as shown in difference between the command stator flux and the Fig.(3) and according to table(1). estimated stator flux magnitude, the second state Table (1) variable is the difference between the commandN(switching mode) 1 2 3 4 5 6 7 8 torque and the estimated electromagnetic torque, andphase-a 1 0 0 0 1 1 1 0 the third one is the angle of stator flux. The controlphase-b 1 1 1 0 0 0 1 0 variables of the direct torque control can be investigated using Fig.(4-a), Fig.(4-b) as follows.phase-c 0 0 1 1 1 0 1 0 Looking at the flux position in Fig (4-a) it is noticed that, states 3,4,5 will increase the flux while states 2,1,6 will decrease it. Similarly, states 4, 3, 2 will increase the current (i.e. the torque) while states 1,6,5 will decrease it. Gathering these states, it is found that for a large increase in flux and a small increase in torque, state 4 is selected, for a small increase in flux and a large increase in torque, state 3 is selected, for a small decrease in flux and a small increase in torque, state 2 is selected, for a large decrease in flux and a small decrease in torque, state 1 is selected, for a small decrease in flux and a large decrease in torque, state 6 is selected, for a small increase in flux and a large decrease in torque, state 503 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-5125 is selected, and for a small decrease in torque and a (2N-3)p /6 = <fn<= (2N-1)p /6 where;constant flux, state 0 is selected. N=1,2,……6.3. SLIDING MODE DIRECT FLUX AND TORQUE Since the flux is kept almost constant, theCONTROLLER DESIGN torque is then proportional to s (instantaneous-slip In order to achieve a high level of ac motor angular velocity). As mentioned above, the velocitycontrol, it is necessary to keep the flux as nearly as of the primary flux is proportional to the dc linkconstant except for the field weakening region [3,4]. voltage, and the direction of rotation of the primaryIn case of sinusoidal ac power supplies the trajectory flux is determined by the active voltage vectors andof primary flux is a smooth circle unlike, it happens the movement of the primary flux will be stoppedin electronic inverters, which is not purely when zero voltage vectors is applied. Therefore, bysinusoidal. Therefore, it is difficult to keep the air changing the ratio of the number of active and zerogap flux constant especially in the low speed region. voltage vectors, the slip angular velocity can beThe instantaneous space vector approach has made it regulated. Hence, the direct torque control ispossible to trace the primary flux and keep it almost obtained by adequate selection between active andconstant. The primary space flux vector of induction zero voltage vectors. Hence, when the torque Td ismotor can be calculated by equation (1). Assuming small compared with T* it is necessary to increase Tdthat the voltage drop of the primary winding as fast as possible by applying the fastest s. On theresistance is small, the trajectory of ys moves in the other hand, when Td reaches T* it is better todirection of vs (Sa, Sb, Sc), the velocity of space non decrease Td as slowly as possible. Thus the output ofzero voltage vector gives the direction and the the torque controller can be classified as follow:amplitude of ys , but by applying a space zero T=1 if Td < T*voltage vector to an induction motor, the movement T=0 if Td= T*of ys will be stopped. Therefore, by selecting these (8) Vs dI2 Is dIdI1 6 dI3 dI4 Is dI5 q q s d d d d d d d d (a) (b) Fig.(4) Vector diagram illustrating the state selection process of flux and torque during transient operation.vectors appropriately, the trajectory of ys will follow T=-1 if Td>T*the specified locus. i.e. by selecting adequate voltage The output of the torque controller is a three levelvectors, ys can be kept almost constant and the hysteresis comparator. Using the three variables, y,rotating velocity of ys can be controlled by changing T and the stator flux position f =(tan -1 y d / y q), thethe output ratio between zero vectors and the following optimum switching tables (2), and (3) areremaining non zero vectors. So, s and ys are greatly obtained. Accordingly, accessing the tables, inconcerned with the direct torque control. In DTC which the inverter output voltages are given by thesystem under study, constant flux can be obtained by output of the two comparators and the angle f(n), thecomparing the primary flux with the command flux. optimum switching model (base drive signals)The output of the flux comparator will be as follows: desirable for driving the base of the inverter can bey = 1 when y s < y s*., and obtained.y = 0 when y s > y s*. The inverter output voltage is then defined as inwhere, e denote the error signal in both y s and Td. equation (6). The entire system has been verified using Matlab, Simulink and the following simulationIn the six step inverter, the d-q plane can be divided results has been obtained.into six regions resulting from the general case: 504 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512Table (2)  Stator flux estimation using motor speed and  T 1 2 3 4 5 6 two line currents (current model) 0 1 V2 V3 V4 V5 V6 V1 One way to eliminate the drift is to use the rotor model in stator coordinate equation (9) in which the 0 0 V8 V7 V8 V7 V8 V7 required signals to be measured are stator currents 0 -1 V4 V5 V6 V1 V2 V3 and rotor speed.Table (3) d r Rr * Lm * is R  T 1 2 3 4 5 6   ( r  j ) (9) dt Lr r L r 1 1 V1 V2 V3 V4 V5 V6 r 1 0 V7 V8 V7 V8 V7 V8 Lm  2 L   i (L  )  m  (10) 1 -1 V5 V6 V1 V2 V3 V4 s s s Lr Lr r3.1 STATOR FLUX ESTIMATION ANALYSIS The accuracy of this estimator depends on In order to have a fast acting and accurate the correct setting of the motor parameterscontrol of induction motor, the flux linkage of the particularly on the rotor time constant which is amotor must be measured. However, it is difficult and critical and the mutual inductance which is lessexpensive. Therefore, it is preferred to estimate the critical. This estimator has good properties at lowmotor flux linkage based on measurement of motor speeds; however, it is sensitive to parameter errors atvoltages, currents and speed if it is required. There high frequency.are several flux estimation techniques, some of themwork with stator flux while others prefer rotor flux.  Stator flux estimation using flux observerAs stator flux control is used in the experiment, the As the voltage model estimator has goodestimators discussed here are mainly stator flux properties at high frequency and the current modelestimators. Two cases are considered throughout this estimator has good properties at low frequency,study: first, neglecting the rotor current while the various ways have been tried to combine them. Amotor is running at no load, the second is to consider straightforward way of combination is adopted [4]the rotor current during load condition. using a low pass filter where the current model estimator and the voltage model estimator are Stator flux estimation using dc link voltage selected at low and high frequency, respectively. and two line currents (voltage model) The same idea of combining the voltage and current One desirable feature of direct torque models to obtain a flux estimator at all speeds iscontrol, or any control method based on stator flux observed in [8]; however, instead of using a low passcontrol is to consider usually the stator flux as an filter to combine the models an observer topology isestimated quantity using terminal measurements [6]. created which uses a closed loop linear controllerAt first, simple flux estimator is obtained from with two deterministically set gains to govern theequation (1) in integral form which is known as transition between the low and high speedvoltage model [7]. The input signals to this estimator estimators. However, the drawback of this type ofare measured stator currents and voltages and if observer is that they require a speed or positionthese signals are without errors like offset, noise and information using a shaft sensor and if the speedif the stator resistance is well tuned, then this information can be obtained without sensor then aestimator will be a perfect solution even if the rotor wide speed range flux estimation could be achieved.current is considered not to be zero. In fact thisestimator is difficult to be used in practice, because The attractive feature of stator fluxof the acquired error signals, offset and drift in the regulated, rotor flux oriented control technique isintegrating hardware. As a result, accumulation of that it is capable of achieving correct torque and fluxthese errors will exist if there is no feedback from control even under detuned conditions [9]. It is clearthe integrator output to the input. The resulting run that the performance of flux estimation depends onaway is a fundamental problem of pure integration. voltages, currents and the model parameterHowever, the challenges in stator flux estimation matching.appear at low frequencies especially when the statorresistance drop is in the same order of the stator 3.2 IMPROVED STATOR FLUX ESTIMATIONvoltage. The problems related to drifting integrators Induction motor stator flux estimation hasare often pointed out as the main difficulties with taken a lot of interest as it is used with severalflux estimation at low speeds and a solution to stator control techniques, such as, classical DTC, statorflux estimation will be presented later. flux field oriented control. Even though Stator flux can be measured [10], this is really difficult and expensive; consequently, the flux is preferred to be estimated based on measurement of voltages, 505 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512currents and velocity if it is required. Several synchronous frame flux regulator requires d-q to a-problems are still under investigation like zero speed b-c additional transformation and the angle ofoperation without speed sensor. The main reasons transformation can be calculated as follows, As arelated to the voltage model are offset and drift simple solution to delete d-q to a-b-c additionalcomponent in the acquired signals and the increased transformation and then the control algorithm will besensitivity to stator resistance variation. These nearly similar to standard DTC which has thereasons decrease the accuracy of stator flux advantage of no reference frame transformation. Theestimation around zero speed. Other difficulties motor slip angle will be calculated through torquearises if the dc link voltage and the switching signals comparator and PI controller as shown in Fig. (1).of the inverter are used to calculate the stator flux, The angle of the stator flux can be obtained easily inone is the voltage drop across the power devices stator reference frame using the rotor positionthemselves, the other is the voltage losses due to through a position sensor or a sensor less solution indead time between two inverter switches in the same addition to the slip angle which has been calculated.branch. The exact voltage calculation is difficultwith respect to the dead time, but these two errors 4. SIMULATION RESULTSmust be compensated. A simple solution is to use The developed control system has beenlow pass filter to avoid integration drift [12], implemented in a computer simulation program,however, this results in magnitude and phase error of modeling the two subsystem induction motor andthe estimated flux. Implementing a pure integrator control algorithm. The controller simulation uses theavoids the bandwidth limitation associated with low parameters of an experimental laboratory prototype.pass filter, especially at low frequencies [13]. These parameters are listed in the Appendix. TheThe modified block diagram of stator flux vector graphs shown in Fig.(5) depict the response of thecontrol scheme is represented in Fig. (1) and the system with a torque command of 4 N.m. Themotor torque is expressed as a cross product of the transient and steady state flux vector in Fig.(5-a)stator and rotor flux as in equation(11). shows nearly a circular path indicating a good flux 3 Lm   regulation. The electromagnetic torque shown inT  P d 2 Ls Lr  r . s   r . s     Fig.(5-b) is close to the commanded value, while the   time taken for the torque reach its commanded value(11) is about 0.02 Sec., ensuring good dynamic torqueand the motion equation is as follow: response. Fig.(5-c) shows the stator currents in phaseT  TL dr a, b and c which are nearly sinusoidal. Fig.(5-d) e  P  B * r (12) shows the waveforms of stator currents in d-q axis, it J dt is noticed that the phase shift between them equalWhere: P is the number of pair poles, J is the 900.combined rotor and load inertia coefficient,  is the Figure(6) shows the response of the systemcombined rotor and load friction coefficient. In for a step change in torque from 3 N.m. to 7 N.m.similar to rotor field orientation the stator flux keeping the flux command constant. Fig.(6-a) showscomponents can be written in synchronous reference the steady state stator flux, which demonstrates anframe as follow: excellent dynamic flux control under changed Ls command torque. Fig.(6-b) shows the developed *e  * (1   Tr * P) λ* (13) torque which is enclosed to a commanded value of 3 ds Lm r N.m. region or 7 N.m. region. The three phase stator currents are shown in Fig.(6-c). It is clear that, these 2LsLr Te * *e qs  * (14) stator currents are changed at the instant of changing 3PLm r * the command torque and swiftly retrieve their steady state values. In equation (4) it is clear that a change inthe control variables yds in synchronous referenceframe is followed with a time constant Tr which ismuch lower than the rotor time constant Tr involvedin traditional stator current control and this gives thefacility to faster controller. From equation (5) it isnoticed that with constant rotor flux command achange in yqs is followed by a corresponding changein torque and this will give decoupling. In principle,the control method proposed here is based on drivingthe stator flux vector toward its reference vector byapplying the appropriate voltage vector and a normalPWM method to obtain constant switchingfrequency and reduced torque pulsations. The 506 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512qs qs ds (a) Stator flux. ds (a) stator flux. (b) Electromagnetic torque. (b) Electromagnetic torque. (c) Stator currents. (c) Stator currents. Fig. (6) System response for step change in load torque. Figure(7) shows simulation results in which a step response of the system is studied at a step change of command torque from 3 N.m. to its 7 N.m. at 0.4 Sec., then, at 0.8 Sec. the command torque is decreased to its initial value (3 N.m.). Behavior of developed torque shows clearly that the response of the torque is as quickly as possible as shown in Fig.(7-b). As can be seen from Fig.(7-c), it is clear that, the stator currents in phases a, b, and c are changing according to the variation of the command torque. The values of currents in the region, where (d) d-q axis stator current. the torque is increased to 7 N.m, is increased to 3.5 Fig. (5) System response at load torque 4 N.m. and Amp.(r.m.s.). motor speed 1500 rpm. 507 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512 5. EXPERIMENTAL IMPLEMENTATION Several dynamic tests were performed to investigate the validity of the proposed control method. The experimental set-up used for the practical investigation is shown in Fig.(8). The induction motor under test is coupled to a self- excited DC generator. The motor is energized from an inverter bridge, which consists of six IGBTs in conjunction with stepper circuits. The DC-link is provided from three-phase bridge rectifier. An incremental encoder is coupled to the motor shaft to measure the motor speed. Actual stator currents are measured using current sensors. The stator voltages are sensed using voltage transducers. This system is (b) Electromagnetic torque. fully controlled by a DSP controller board which is installed on a PC computer. Fig. (7) System response for step change in load torque. A series of experimental results was carried out to verify the validity of the proposed control scheme for the drive system. The results were obtained at different operating points. The results were obtained at different operating points. First, the drive system response was obtained at heavy load and a speed below the base speed. Figure (9) showsqs the motor speed response at 1200 rpm. The motor started with almost no overshoot and follows its command with nearly zero steady state error. The motor phase currents are also shown in Fig. (9). The response of the control system due to a step change in speed command is shown in Fig. (10). The speed command is changed from 120 rad/s to ds 100 rad/s. It can be seen from Fig. (10) that the (a) stator flux. motor speed is decelerated smoothly to follow its reference value. Figure (11), show the motor response for step up in speed command, which change from 100 rad/s to 150 rad/s. 508 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512 3-phaseAC supply Edc Sa Sb Sc Load Sa Sb Sc Incremental encoder Gating drives IM DC Gen. Oscilloscope Vector Control Gate Pulse Generator D/A A/D TMS320C31 Converte Converte r r Incremental Data-bus encoder RS-232C interface Host Computer DSP board (ds1102) Fig. (8) Experimental set-up for DSP-based control of IM drive Fig. (9) Drive response for reference speed of 1200 rpm (x-axis Time (ms)) 509 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512 Fig. (10) System performance for step down in speed command (x-axis Time (ms)) Fig. (11) System performance for step up in speed command (x-axis Time (ms)) 510 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-5126. CONCLUSIONS Components and Systems Journal, Vol. In this paper, the basic concept of DTC is 34, No. 3, pp. 303-319, March 2006.implemented using a DSP control system by [7] A. Linder: “ A Rapid-Prototyping Systemmodifying the classical method. The advantage of Based on Standard PCs with RTAI asthe proposed method lies in achieving constant Real-Time Operating System, ” Thirdswitching frequency and solving the starting Real-Time Linux Workshop, Milan, Italy,problem without a dither signal. A further November 26 -29, 2001improvement of this work can be achieved by using [8] R. Kennel, A. El-refaei, F. Elkady, E.stator flux estimation and reduced torque pulsation Elkholy and S. A. Mahmoud “Improvedtechniques. The existence of non linear torque and direct torque control for induction motorflux hysteresis controllers result in noticeable drives with rapid prototyping system”torque pulsations and variable switching frequency. IEEE, IECON, 2003 U.S.A.Thus, the proposed control method can give [9] R. Kennel, A. El-refaei, F. Elkady, E.transient performance like DTC and reduced torque Elkholy and S. A. Mahmoud “ Torquepulsations at steady state. An inverter switching Ripple Minimization for Induction Motorpattern that can achieve DTC method for induction Drives with Direct Torque Control (DTC),motors in which the analytical derivation is based ”The Fifth International Conference onon the prediction of the voltage vector direction Power Electronics and Drive Systemswhich is required to compensate electromagnetic PEDS 2003, Singapore, November 17 –torque and stator flux errors. In general, an 20, 2003.optimistic experimental results and an improved [10] J. Chen. Yongdong “ Virtual Vectorsperformance over other existing schemes is Based Predictive Control of Torque andobtained. Flux of Induction Motor and Speed Sensorless Drives, ” IEEE, IECON 1999.REFERENCES [11] Y. Xue, X. Xu, T. G. Habetler and D. M. [1] J. Holtz and J. Quan “ Sensorless Vector Divan “ A Stator Flux-Oriented Voltage Control of Induction Motor at Very Low Source Variable-Speed Drive Based on dc Speed Using a Non Linear Inverter Model Link Measurement,” IEEE Transaction and Parameter Identification,” IEEE Industry applications, Vol. 27, No. 5, pp. Transaction Industry Applications Society 962-969, September/October 1991. Annual Meeting, Chicago, Sept. 30- Oct. [12] T. G. Habetler, F. Profumo, G. Griva “ 4, 2001. Stator Resistance Tuning In A stator Flux [2] M. Depenbrock, “ Direct Self Control Field Oriented Drive Using An (DSC) of Inverter-fed Induction Motors, ” Instantaneous Hybrid Flux Estimator, ” IEEE Transaction Power Electronics, Vol. The European Power Electronics 3, No. 4, pp. 420-429, October1988. Association, Brighton, No. 13, pp. 292- [3] I. Takahashi and T. Noguchi “ A new 299, September, 1993. Quick Response and High Efficiency [13] N. R. Idris, A. Halim, N. A. Azly, “ Direct Control Strategy of An Induction Motor, ” Torque Control of Induction Machines IEEE Transaction Industry Applications, With Constant Switching Frequency and Vol. IA-22, No.5, PP. 820-827, .Sept./Oct. Improved Stator Flux Estimation,” 27th 1986. Annual Conference of the IEEE Ind. Elect. [4] E. Flach, R. Hoffmann, P. Mutschler “ Society, pp. 1285-1291. Direct Mean Torque Control of Induction [14] L. Jansen, D. Lorenz “Observer Based Motor, ” in Conf. Rec. of the European Direct Field Orientation Analysis and Conference on Power Electronics and Comparison of Alternative Methods,” Applications, pp. 3672-3676, 1997. IEEE Transaction Industry applications, [5] E. E. El-Kholy , S. A. Mahmoud, R. Vol. 30, No. 4, pp. 945-953, July /August Kennel, A. El-Refaei, and F. El-Kady, 1994. “Torque Ripple Minimization for [15] B. Kenny, D. Lorenz “ Deadbeat Direct Induction Motor Drives With Direct Torque Control Of Induction Machines Torque Control” Electrical Power Using Self Sensing at low and zero speed, Components and Systems Journal, Vol. ” Ph. D thesis Wosconsin, Madison, U. S. 33, No. 8, pp. 845-859, August 2005. A., 2001. [6] E. E. El-Kholy , S. A. Mahmoud, R. [16] J. Holtz, J. Quan “ Sensorless Vector Kennel, A. El-Refaei, and F. El-Kady, Control of Induction Motors at Very Low “Analysis and Implementation of a New Speed, ” Ph. D thesis, Wuppertal Space-Vector Current Regulation for university, 2002. Induction Motor Drives” Electrical Power [17] A. B. Plunkett, J. D. D’atre, and T. A. Lipo,“ Synchronous Control of Static AC 511 | P a g e
    • A. Alwadie / International Journal of Engineering Research and Applications (IJERA) ISSN: 2248-9622 www.ijera.com Vol. 2, Issue 6, November- December 2012, pp.501-512 Induction Motor Drive, ” IEEE Trans. On Industry Applications, Vol. IA-15, No.4, pp. 430-437, July/August 1979. [18] D. Casadei, G. Serra, A. Tani, L. Zarri “ Performance Analysis of a Speed Sensorless Induction Motor drive Based on a Constant Switching Frequency DTC Scheme, ” IEEE, IECON, 2000. [19] Habetler “Direct Torque control of an Induction Machines Using Space Vector Modulation,” IEEE Transaction Industry Applications, Vol. 28, No. 5, pp. 11045- 1053, September/October 1992. [20] J. Monteiro, G. Marques, and J. Palma “A Simplified Stator Flux Vector Control Method in d-q Co-ordinates For Induction Machines,” ISIE 2000.I. AppendixThe induction motor is a squirrel cage and has thefollowing data:Rated power : 1.1 KWRated line voltage : 380 volt.No. of pole pair : 2.0Stator resistance : 4. 4826 ohm.Rotor resistance : 3. 6840 ohm.Stator leakage inductance : 0.0221 Henry.Rotor leakage inductance : 0.0221 Henry.Mutual inductance : 0.4114 Henry.Supply frequency : 50.0Rated load torque : 11.0 N. m. 512 | P a g e