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Prediction of viable performance of wireless sensor network by using finite

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  • 1. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN INTERNATIONAL JOURNAL OF ELECTRONICS AND 0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEMECOMMUNICATION ENGINEERING & TECHNOLOGY (IJECET)ISSN 0976 – 6464(Print)ISSN 0976 – 6472(Online)Volume 4, Issue 2, March – April, 2013, pp. 144-150 IJECET© IAEME: www.iaeme.com/ijecet.aspJournal Impact Factor (2013): 5.8896 (Calculated by GISI) ©IAEMEwww.jifactor.com PREDICTION OF VIABLE PERFORMANCE OF WIRELESS SENSOR NETWORK BY USING FINITE-DIFFERENCE TIME-DOMAIN S.R.Shankar a, Dr.G.Kalivarathanb a Research Scholar, CMJ University, Meghalaya, Shillong. b Principal/ PSN Institute of Technology and Science, Tirunelveli, Tamilnadu, Supervisor, CMJ University, Shillong. Email:sakthi_eswar@yahoo.com ABSTRACT Wireless Sensor Networks (WSNs) offer a promising solution to monitor the physical world around us. A WSN is comprised of a large number of sensing devices, often referred to as motes or sensor nodes, which are deployed at the region of interest. The wireless sensor nodes have processing and communication capabilities, which enable them to autonomously gather information from the environment and then to generate and deliver “report-messages” to the remote base stations (remote users). The economic benefits of WSNs are mainly due to the exclusion of expensive infrastructure required by the wired sensor networks. The Finite- Difference Time-Domain (FDTD) method introduced powerful tool for solving various electromagnetic (EM) problems. The development of a 3-D FDTD method for planar devices is presented in this work. This method offers an accurate design technique for new type of microstrip filters. A signal estimation technique was developed in order to reduce the FDTD computation time. By using this signal estimation technique, the number of FDTD iterations was reduced up to five times. A design algorithm uses FDTD and Neural Networks. This is much faster than the FDTD method alone. Mobile communications systems require preselect filters with enhanced properties. This work presents the research on novel microstrip filters. The technology required by the newly developed filters is economical, no short-circuit elements and no lumped components are needed. The designs can be easily extended for planar HTS technology. An emphasis is put on the development of dual mode filters and filters with cross-coupled novel resonators. This introductive section presents the work on low-pass and band-pass conventional filters. Keywords: Finite-Difference Time-Domain, Mobile communications, Resonator, Microstrip, Dual mode resonators 144
  • 2. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME1.0 INTRODUCTION The FDTD signal excitation contains a voltage source below the microstrip line. Thesignal needs to propagate a certain distance along the line to let the transient modes to vanishand reach their true modal nature. In order to minimize the computational domain, the pulsepropagation along a simple input microstrip line is first simulated. After the signal acquiresthe correct transversal profile, the pulse is copied at the input of the microstrip device in orderto be analyzed. The same input signal can be used in several simulations of various planarstructures with the substrate and the FDTD grid not changed. FDTD method allows theanalysis of the electromagnetic field in the planar structure at different time instances. Asexpected, the FDTD simulations expose the concentration of the electromagnetic energy justunderneath the microstrip lines and patches. The analysis in time domain, easily illustratesthe incident, reflected and transmitted signals. For example, the plot of the electric fieldcomponent Ez the propagation of a pulse along a bent microstrip line. After 550 time steps,the incident pulse is still on the input line. After additional 400 time steps (∆t =0.27 ps), thepulse turned left, and propagated along a wider microstrip.2.0 DISPERSION EFFECTS The fundamental propagation mode for the microstrip line is considered asapproximating the Transversal Electric and Magnetic (TEM) mode, when the fields areoscillating only in a perpendicular plane on the direction of propagation. For an ideal TEMmode, not including the material effects, the pulse should not encounter any dispersion. Inpractice, the microstrip properties are considered frequency independent for a frequencybandwidth up to 2 GHz, when designing on low (εr=2.55) dielectric constant substrate.However, the dispersion effect increases with the increase of the dielectric constant and theworking frequency. The FDTD simulations clearly illustrate the dispersion effects in timedomain. The Ez field just underneath a straight microstrip line versus time at differentlocations. The distances y from the source plane to the measuring points are given in ∆y units.In this case, the substrate was alumina (Al2O3) having a dielectric constant εr=8.88 andthickness h=0.535 mm. The 50 line width was w50=0.546 mm. The FDTD grid had∆x=∆y=0.5636 mm and ∆z=0.127 mm. The time step ∆t=0.27 ps and the incident Gaussianpulse had T=28 ∆t in width and T0=4 T as initial delay. While at the source position, thepulse is an undistorted Gaussian, after propagating a certain distance, the amplitude decreasesand the pulse broadens with a strong negative tail. Since the dielectric layers were considerednon-dispersive and the numerical dispersion of FDTD is negligible, the pulse distortionobserved along a microstrip line is intrinsic to the fundamental “Quasi-TEM” propagatingmode.3.0 FDTD ANALYSIS OF DIFFERENT TYPES OF MICROSTRIP DEVICES Several microstrip devices have been analyzed to verify accuracy of the developedFDTD method. The method was applied to numerous microstrip devices manufactured onsubstrates having different values for dielectric constant and thickness. In some simple cases,the simulated scattering S parameters were compared with the S parameters provided bycommercial software. In other cases, the simulated response was compared withmeasurements or data taken from literature. A linear low impedance resonator on a substratea dielectric constant of εr=2.38 and a thickness of h=0.71 mm was simulated using the 145
  • 3. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEMEdeveloped FDTD method. The 50 line had the width of w50=1.87 mm. The resonator linewidth was wreson=3.07 and the resonator length was lreson=15.54 mm. For such a simpledevice, a commercial software, such as Touchstone, modeled the step discontinuities andcould provide accurate S parameters.4.0 GAP END COUPLED LINEAR RESONATOR An end coupled linear resonator presents a higher overall quality factor (Q-factor)than the linear low impedance section discussed above. Consequently, the output signaldecays slowly and the FDTD simulation requires a large number of time steps to completethe simulation. The gap end coupled linear resonator was designed as having the line widthwreson=w50= 0.61 mm, the length lreson=10 mm and the coupling gap s=0.3 mm. The usedsubstrate had dielectric constant of εr=9.98 and thickness of h=0.635 mm. The used FDTDgrid had ∆x=∆y=0.1525 mm and ∆z=0.127 mm and the time step was ∆t=0.27 ps.5.0 DUAL MODE RESONATORS AND FILTERS Dual mode resonators (DMR) are resonators perturbed in such a way that tworesonating modes, initially degenerate, can couple each other. A DMR offers a dual modefilter (DMF) behavior, when certain conditions on the input and output couplings aresatisfied. A planar design of dual mode filters was presented using λ meander resonators. Aresonator forms a 2-pole filter and consists of a meander loop with the input and outputstructures, two optional stubs for independent tuning of the resonant frequencies of theorthogonal modes, and a stub providing a coupling between modes. Two types of filter arepossible depending on the stubs location. For the symmetric filter, the stub is located on theAA plane in Fig. 4.5 and the frequency response has two transmission zeros located on bothsides of the passband. The asymmetric filter has the stub on BB plane and does not presentany transmission zero. Four pole elliptic filters, also exhibit transmission zeros, but, unlike inthe 2-pole symmetric filter, the position of the zeros can be fully controlled. Each of thesetwo rings has three lines attached: an input (or output) line, a line providing major couplingbetween rings, and a line providing minor coupling. Moreover, each ring has a stub for tuningof the center frequency of one of the modes and obviously a stub providing coupling betweenorthogonal modes. The input/output structure can be realized in many ways, but foroptimized sensitivity they have been carried out as sections of coupled transmission lines.Coupling between rings have been realized by using capacitive gaps. The lines between ringsand coupling elements provide appropriate transformations from the gaps or in/out structuresas well as a spatial separation between rings.6.0 QUASI FRACTAL DUAL MODE RESONATORS The DMF patch filters have good power handling properties but they occupy a largersurface area. In order to reduce the patch size, a technique from microstrip antenna designwas borrowed. When slots are cut in the square patch, The perturbation required for the dualmode effect is provided by the slots’ asymmetry. The coupling between the modes can becontrolled by the difference in the length of the diagonal slots. For the tuning configuration inFig. 4.26a, the insertion loss decreased to approximately 12 dB. The filter presents twotransmission zeros on each side of the pass-band 146
  • 4. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME7.0 RESULTS The dispersion effect increases with the increase of the dielectric constant and theworking frequency. The FDTD simulations clearly illustrate the dispersion effects in timedomain. The dispersive effects increase with frequency, therefore these absorbing boundaryconditions gave spurious numerical reflections at higher frequency. Figure1. The pulse is guided by the microstrip line along Oy axis Figure2. The pulse changes direction after a mittered corner and reaches Wider microstrip 147
  • 5. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEMEThe accuracy and the consistency of the present developed FDTD method were proved bycomparing the results of a simulation on a meander line to the measurements. The performednumerical experiments showed that a straight transition from the lumped source to themicrostrip line provides a better matching than using a tapered line. Figure3. Calculated S21 (dotted line) and S11 (continuous line), and measured, S11 (*) for a meander line on Al2O3 with εr=9.98, h=0.635 mm, w = 0.61 mmThe decimation of y2 with the desampling rate desra=200 corresponded to an increase in thetime step. The very small FDTD time step was required by the stability Courant criterion, butit could cause a coarse frequency step, when translated in frequency domain. The signal y3resulted from decimation was provided to the ARMA algorithm. The obtained ARMAcoefficients could be considered as the coefficients of an Infinite Impulse Response (IIR)filter, which could identify the estimated signal. In this case, the order of the IIR filter was K= M = 16. Figure4. S parameters (in dB) versus frequency (GHz) of the meander loop Dual mode filter 148
  • 6. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME8.0 CONCLUSION Dual mode filters (DMF) can offer a solution when a narrow band is desired. Openloop and patch DMF are investigated and DMF for GSM / GPRS bands are designed. Anovel quasi-fractal resonator is developed in order to reduce the square DMF patch size.Filters with cross-coupled resonators are investigated due to their ability to show generalizedChebyshev (or quasi-elliptic) response. Filters with cross-coupled open half-wavelengthloops are designed. For mobile communications bands, the simply half-wavelength resonatorsare inconveniently long, therefore novel type of resonators are developed. Original theoreticalprinciples of size reducing for these resonators are presented. The newly developed filterstake up to 32% of the surface of a simple square half-wavelength resonator, both beingdesigned for 900 MHz. The coupling coefficients of the novel resonators, function of theirrelative positions are obtained using the 3D-FDTD method. The external quality factorfunctions of the input / output line positions are obtained in the same way. An iterativeARMA signal estimation technique was developed in order to reduce the FDTD computationtime. This is of importance especially in the case of the analysis of narrowband resonatingstructures. Therefore, with the present technique, the required number of iterations can bereduced up to five times, keeping the same accuracy of the results. Finally, a new designtechnique using FDTD method and Neural Networks was developed and applied to amicrostrip filter. The total design time was reduced twofold. The ARMA signal estimationtechnique was first utilized to reduce the computation time for each FDTD run. Secondly, thenumber of FDTD simulations was decreased using the device model provided by a neuralnetwork with the ARMA coefficients at the output. The trained network was thenincorporated in an optimization procedure for a microstrip filter design.REFERENCES[1] M. G. Banciu, E. Ambikairajah, R. Ramer, “Microstrip Filter Design Using FDTD andNeural Networks”, Microwave and Optical Technology Letters, vol. 34, No. 3, August 5,2002, pp. 219-224.[2] M. G. Banciu, R. Ramer, “Analysis of Microstrip Circuits Using a Finite DifferenceTime-Domain”, Proceedings of the 4th World Multi conference on Circuits, Systems,Communications and Computers, Proceedings CSCC 2000,Vouliagmeni, Greece, July 2000, ISBN 960-8052-19-X, pp. 4611-4615[3] M. G. Banciu, R. Ramer, “Analysis of Microstrip Circuits Using a Finite DifferenceTime-Domain”, in Advances in Physics, Electronics and Signal Processing Applications,edited by N. E. Mastorakis, World Scientific and Engineering Soc.Press, Danvers, MA, 2000,ISBN: 960-8052-17-3, pp. 156-160[4] E. H. Fooks, R. A. Zakarevicius, “Microwave Engineering Using Microstrip Circuits”,Prentice Hall, 1989[5] R. L. Veghte, C.A. Balanis, Dispersion of Transient Signals in Microstrip TransmissionLines, IEEE Trans. Microwave Theory Tech., Vol. MTT-34, No. 12, 1986, pp. 1427-1436[6] HP-Eesof Microwave & RF Circuit Design – Circuit Element Catalog, HewlettR-Packard,March 1994[7] M. G. Banciu, R. Ramer, “Design of Microstrip Dual Mode Filters Using FiniteDifference Time-Domain Method”, Proceedings of the Asia-Pacific Microwave Conference –APMC 2000, December 2000, Sydney, vol. 1, pp. 975-978 149
  • 7. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME[8] A. C. Kundu, I. Awai, T. Kajitani, "Attenuation pole frequency control of a dual modecircular microstrip ring resonator", 29th European Microwave Conference 99. IncorporatingMIOP 99. Conference Proceedings. Microwave Eng. Eur. Part vol.2, 1999, pp.329-32 vol.2.London, UK.[9] I. Awai, “General Theory of a Circular Dual-Mode Resonator and filter”, IEICE Trans.Electron., vol. E81-C, November 1998, pp.1757-1763[10] A. C. Kundu, I. Awai, “Control of Attenuation Pole Frequency of a Dual-ModeMicrostrip Ring Resonator Bandpass Filter“, IEEE Transactions on Microwave Theory andTechniques, vol. 49, 2001, pp. 1113-1117[11] T.Regu and Dr.G.Kalivarathan, “Prediction of Wireless Communication Systems in theContext of Modeling” International journal of Electronics and Communication Engineering&Technology (IJECET), Volume 4, Issue 1, 2013, pp. 11 - 17, ISSN Print: 0976- 6464,ISSN Online: 0976 –6472[12] Neeraj Tiwari, Rahul Anshumali and Prabal Pratap Singh, “Wireless Sensor Networks:Limitation, Layerwise Security Threats, Intruder Detection”, International journal ofElectronics and Communication Engineering & Technology (IJECET), Volume 3, Issue 2,2012, pp. 22 - 31, ISSN Print: 0976- 6464, ISSN Online: 0976 –6472[13] T.Regu and Dr.G.Kalivarathan, “Prediction of a Reliable Code for WirelessCommunication Systems” International Journal of Electrical Engineering & Technology(IJEET), Volume 4, Issue 1, 2013, pp. 19 - 26, ISSN Print : 0976-6545, ISSN Online:0976-6553[14] Varun Shukla, Arti Saxena and Swati Jain, “A New Rectangular Dielectric ResonatorAntenna Compatible for Mobile Communication or Broadband Applications”, Internationaljournal of Electronics and Communication Engineering &Technology (IJECET), Volume 3,Issue 2, 2012, pp. 360 - 368, ISSN Print: 0976- 6464, ISSN Online: 0976 –6472 150

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