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  • 1. EMI Protection forCommunication Systems
  • 2. DISCLAIMER OF WARRANTYThe technical descriptions, procedures, and computer programs in this book havebeen developed with the greatest of care and they have been useful to the author in abroad range of applications; however, they are provided as is, without warranty ofany kind. Artech House, Inc. and the author and editors of the book titled EMI Pro-tection for Communication Systems make no warranties, expressed or implied, thatthe equations, programs, and procedures in this book or its associated software arefree of error, or are consistent with any particular standard of merchantability, orwill meet your requirements for any particular application. They should not berelied upon for solving a problem whose incorrect solution could result in injury to aperson or loss of property. Any use of the programs or procedures in such a manneris at the user’s own risk. The editors, author, and publisher disclaim all liability fordirect, incidental, or consequent damages resulting from use of the programs or pro-cedures in this book or the associated software. For a listing of recent related Artech House titles turn to the back of this book.
  • 3. EMI Protection forCommunication Systems Kresimir Malaric
  • 4. Library of Congress Cataloging-in-Publication DataA catalog record for this book is available from the U.S. Library of Congress.British Library Cataloguing in Publication DataA catalogue record for this book is available from the British Library.ISBN-13: 978-1-59693-313-2Cover design by Greg Lamb© 2010 ARTECH HOUSE685 Canton StreetNorwood, MA 02062All rights reserved. Printed and bound in the United States of America. No part of this bookmay be reproduced or utilized in any form or by any means, electronic or mechanical, includ-ing photocopying, recording, or by any information storage and retrieval system, withoutpermission in writing from the publisher. All terms mentioned in this book that are known to be trademarks or service marks havebeen appropriately capitalized. Artech House cannot attest to the accuracy of this informa-tion. Use of a term in this book should not be regarded as affecting the validity of any trade-mark or service mark.10 9 8 7 6 5 4 3 2 1 Disclaimer: This eBook does not include the ancillary media that was packaged with the original printed version of the book.
  • 5. Contents Preface xiii CHAPTER 1 Communications Systems 1 1.1 Components of Communications Systems 1 1.2 Transmitter Systems 2 1.2.1 Transmitter 3 1.2.2 Randomization 4 1.2.3 Encryption 5 1.2.4 Encoder 5 1.2.5 Interleaving 9 1.2.6 Modulation 10 1.2.7 Mixer (Upconverter) 10 1.2.8 Filter 11 1.3 Receiver Systems 11 1.3.1 Filter 11 1.3.2 Mixer (Downconverter) 12 1.3.3 Demodulator 12 1.3.4 Deinterleaver 12 1.3.5 Decoder 13 1.3.6 Decryptor 15 1.3.7 Derandomizer 15 1.3.8 Demultiplexer 16 1.3.9 Received Power 16 1.4 User Interface 18 1.4.1 Graphical User Interface (GUI) 18 1.4.2 Voice User Interface (VOI) 19 1.5 Antenna Systems 19 1.5.1 Duplexer 19 1.5.2 Antenna 20 1.6 Power Supplies 22 1.6.1 Power Supply Types 23 1.6.2 Power Amplifier 23 v
  • 6. vi Contents 1.7 Considerations for Voice Versus Data 23 1.7.1 Text 23 1.7.2 Images 24 1.7.3 Voice 24 1.7.4 Video 24 Selected Bibliography 24 CHAPTER 2 Electromagnetic Spectrum Used for Communications 27 2.1 Electromagnetic Spectrum 27 2.1.1 Extra Low Frequency (ELF) 28 2.1.2 Super Low Frequency (SLF) 28 2.1.3 Ultra Low Frequencies (ULF) 29 2.1.4 Very Low Frequency (VLF) 29 2.1.5 Low Frequency (LF) 29 2.1.6 Medium Frequency (MF) 29 2.1.7 High Frequency (HF) 29 2.1.8 Very High Frequency (VHF) 29 2.1.9 Ultra High Frequency (UHF) 29 2.1.10 Super High Frequency (SHF) 30 2.1.11 Extra High Frequency (EHF) 30 2.1.12 Infrared (IR) 30 2.1.13 Visible 30 2.2 Spectrum Division 30 Selected Bibliography 33 CHAPTER 3 Electromagnetic Properties of Communications Systems 35 3.1 Fundamental Communications System Electromagnetics 35 3.1.1 Smith Chart 39 3.1.2 Snell’s Law of Reflection and Refraction 42 3.2 Wave Generation and Propagation in Free Space 44 3.2.1 Maxwell’s Equations 44 3.2.2 Wave Propagation 46 3.2.3 Wave Polarization 47 3.2.4 Fresnel Knife-Edge Diffraction 48 3.2.5 Path Loss Prediction 51 3.3 Wave Generation and Propagation in the Terrestrial Atmosphere 53 3.3.1 Absorption and Scattering 53 3.3.2 Wave Propagation in the Atmosphere 54 Selected Bibliography 55 CHAPTER 4 Electromagnetic Interference 57 4.1 Electromagnetic Interference with Wave Propagation and Reception 57 4.1.1 Additive White Gaussian Noise (AWGN) 57
  • 7. Contents vii 4.1.2 Thermal Noise 58 4.1.3 Shot Noise 58 4.1.4 Flicker (1/f ) Noise 58 4.1.5 Burst Noise 59 4.1.6 Noise Spectral Density 59 4.1.7 Effective Input Noise Temperature 59 4.2 Natural Sources of Electromagnetic Interference 59 4.2.1 Lightning and Electrostatic Discharge 59 4.2.2 Multipath Effects Caused by Surface Feature Diffraction and Attenuation 64 4.2.3 Attenuation by Atmospheric Water 65 4.2.4 Attenuation by Atmospheric Pollutants 67 4.2.5 Sunspot Activity 68 4.3 Manmade Sources of Electromagnetic Interference 69 4.3.1 Commercial Radio and Telephone Communications 69 4.3.2 Military Radio and Telephone Communications 74 4.3.3 Commercial Radar Systems 74 4.3.4 Industrial Sources 75 4.3.5 Intentional Interference 76 Selected Bibliography 77 CHAPTER 5 Filter Interference Control 79 5.1 Filters 79 5.1.1 Lowpass Filter 80 5.1.2 Highpass Filter 80 5.1.3 Bandpass Filter 81 5.1.4 Bandstop Filter 83 5.1.5 Resonator 83 5.2 Analog Filters 85 5.2.1 Butterworth Filter 85 5.2.2 Chebyshev Filters 86 5.2.3 Bessel Filters 87 5.2.4 Elliptic Filters 88 5.2.5 Passive Filters 88 5.2.6 Active Filters 91 5.3 Digital Filters 91 5.3.1 FIR Filters 93 5.3.2 IIR Filters 94 5.4 Microwave Filters 97 5.4.1 Lumped-Element Filters 97 5.4.2 Waveguide Cavity Filters 98 5.4.3 Dielectric Resonator 100 Selected Bibliography 101
  • 8. viii Contents CHAPTER 6 Modulation Techniques 103 6.1 Signal Processing and Detection 103 6.2 Modulation and Demodulation 105 6.2.1 Analog Modulations 105 6.2.2 Digital Modulation 112 6.3 Control of System Drift 120 Selected Bibliography 120 CHAPTER 7 Electromagnetic Field Coupling to Wire 123 7.1 Field-to-Wire Coupling 123 7.1.1 Skin Effect 123 7.1.2 Unshielded Twisted Pair (UTP) 125 7.1.3 Ferrite Filter 126 7.2 Electric Field Coupling to Wires 128 7.3 Magnetic Field Coupling to Wires 131 7.4 Cable Shielding 132 7.4.1 Tri-Axial Cable 133 7.4.2 Cable Termination 133 7.4.3 Shielded Twisted Pair Cables 134 Selected Bibliography 136 CHAPTER 8 Electromagnetic Field-to-Aperture Coupling 137 8.1 Field-to-Aperture Coupling 137 8.1.1 Shielding Effectiveness (SE) 138 8.1.2 Multiple Apertures 138 8.1.3 Waveguides Below Cutoff 140 8.2 Reflection and Transmission 141 8.2.1 Electric Field 145 8.2.2 Magnetic Field 146 8.3 Equipment Shielding 147 8.3.1 Gasketing 147 8.3.2 PCB Protection 148 8.3.3 Magnetic Shield 149 Selected Bibliography 151 CHAPTER 9 Electrical Grounding and Bonding 153 9.1 Grounding for Safety 154 9.1.1 Shock Control 154 9.1.2 Fault Protection 155 9.2 Grounding for Voltage Reference Control 156 9.2.1 Floating Ground 156 9.2.2 Single Point Ground 157
  • 9. Contents ix 9.2.3 Multipoint Ground 158 9.2.4 Equipotential Plane 158 9.3 Bonding for Current Control 159 9.3.1 Bonding Classes 160 9.3.2 Strap Bond for Class R 160 9.3.3 Resistance Requirements 162 9.4 Types of Electrical Bonds 162 9.4.1 Welding and Brazing 163 9.4.2 Bolting 163 9.4.3 Conductive Adhesive 164 9.5 Galvanic (Dissimilar Metal) Corrosion Control 164 Selected Bibliography 166 CHAPTER 10 Emissions and Susceptibility—Radiated and Conducted 167 10.1 Control of Emissions and Susceptibility—Radiated and Conducted 167 10.1.1 Sources of Electromagnetic Interference 167 10.1.2 Test Requirements for Emission and Susceptibility 171 10.1.3 Standard Organizations 173 10.2 Commercial Requirements 177 10.3 Military Requirements 178 10.3.1 Specific Conducted Emissions Requirements Mil-Std 461E 178 10.3.2 Specific Conducted Susceptibility Requirements Mil-Std 461E 179 10.3.3 Radiated Emissions Requirements Mil-Std 461E 181 10.3.4 Radiated Susceptibility Requirements Mil-Std 461E 182 Selected Bibliography 182 CHAPTER 11 Measurement Facilities 185 11.1 Full Anechoic and Semianechoic Chambers 185 11.1.1 Absorbers 187 11.1.2 Ferrite Tiles 189 11.2 Open Area Test Site (OATS) 191 11.3 Reverberation Chamber 193 11.4 TEM Cell 195 11.4.1 Characteristic Impedance 196 11.4.2 Higher-Order Modes 197 11.4.3 TEM Cell Construction 198 11.4.4 Parameter Measurements 200 11.5 GTEM Cell 201 11.5.1 GTEM Cell Characteristics 203 11.5.2 GTEM Cell Construction 203 11.5.3 GTEM Cell Parameter Measurement 204 11.5.4 Current Distribution at Septum 211 Selected Bibliography 212
  • 10. x Contents CHAPTER 12 Typical Test Equipment 215 12.1 LISN—Line Impedance Stabilization Network 215 12.2 Coupling Capacitor 216 12.3 Coupling Transformer 217 12.4 Parallel Plate for Susceptibility Test 217 12.5 Coupling Clamps and Probes 218 12.5.1 Capacitive Coupling Clamp 219 12.5.2 Current Probe 220 12.6 Injection Clamps and Probes 221 12.6.1 Current Injection Probe 221 12.6.2 EM Clamp 221 12.6.3 Electrostatic Discharge (ESD) Generator 223 12.7 EMI Receiver 224 12.8 Spectrum Analyzer 225 12.9 Oscilloscopes 225 Selected Bibliography 225 CHAPTER 13 Control of Measurement Uncertainty 227 13.1 Evaluation of Standard Uncertainty 227 13.1.1 Type A Evaluation of Standard Uncertainty 227 13.1.2 Type B Evaluation of Standard Uncertainty 228 13.2 Distributions 228 13.2.1 Normal (Gaussian) Distribution 229 13.2.2 Rectangular Distribution 229 13.2.3 U-Shaped Distribution 230 13.2.4 Combined Standard Uncertainty 230 13.2.5 Expanded Uncertainty 231 13.3 Sources of Error 231 13.3.1 Stability 231 13.3.2 Environment 231 13.3.3 Calibration Data 231 13.3.4 Resolution 232 13.3.5 Device Positioning 232 13.3.6 RF Mismatch Error 232 13.4 Definitions 232 Selected Bibliography 232 Appendix A Communication Frequency Allocations 235 A.1 Frequency Allocation in the United States 235 A.2 International Frequency Allocation 245
  • 11. Contents xi Appendix B List of EMC Standards Regarding Emission and Susceptibility 255 B.1 Cenelec 255 B.2 Australian Standards 256 B.3 Canadian Standards 256 B.4 European Standards 258 B.5 Other Standards 259 Acronyms and Abbreviations 261 Glossary 265 About the Author 267 Index 269
  • 12. Preface Communication today is not as easy as it was in the past. Protecting numerous com- munication services, which are operating in the same or adjacent communication channels, has become increasingly challenging. Communication systems have to be protected from both natural and manmade interference. Electromagnetic interfer- ence can be radiated or conducted, intentional or unintentional. Understanding physical characteristics of wave propagation is necessary to comprehend the mecha- nisms of electric and magnetic coupling in the communication signal paths. Differ- ent modulating techniques, as well as encoding and encrypting, can improve bit error rates (BER) and signal quality. Communication systems must be designed properly, so that the performance of their system capabilities is not subject to degra- dation or complete loss due to electromagnetic interference. Although there are numerous books available on electromagnetic compatibil- ity, signal processing, and electromagnetic theory, there is no book offering a com- prehensive description of technologies for the protection of communication systems, which includes discussions on the improvement of existing communication systems and the creation of new systems. The book provides laymen with basic information and definitions of problems regarding electromagnetic interference in communication systems. In addition, it gives an experienced practitioner knowl- edge of how to solve possible problems in both digital and analog communication systems. The examples given in the book are intended for an easier comprehension of otherwise demanding electromagnetic problems. The book’s primary audience includes designers, researchers, and graduate students in the area of communications. The book is organized into 13 chapters dealing with fundamental concerns of developers and users of communication systems. • Chapter 1 gives an overview of communication system components. • Chapter 2 deals with the use of the electromagnetic spectrum for communica- tions. • Chapter 3 describes wave propagation in free space and the terrestrial atmo- sphere. • Chapter 4 discusses natural sources of electromagnetic interference, such as attenuation of atmospheric water or lightning, as well as numerous manmade sources of electromagnetic interference. • Chapter 5 covers analog, digital, and microwave filters. • Chapter 6 deals with signal processing and modulation/demodulation issues in communication systems. xiii
  • 13. xiv Preface • Chapters 7 through 9 are devoted to electromagnetic field-to-wire and aper- ture coupling, as well as to electrical grounding and bonding. • Chapter 10 gives the commercial and military requirements for radiated and conducted emission and susceptibility. Facilities for EMI measurement such as TEM and GTEM cells, open area test sites, and reverberation chambers are covered in Chapter 11. • Chapter 12 includes the description of coupling capacitors and transformers, coupling and injection clamps and probes, and other test equipment. • Chapter 13 deals with the control of measurement uncertainties. • Appendix A gives a list of communication frequencies, and Appendix B gives a list of EMC standards. The program TEM-GTEM on the CD-ROM accompanying this book works in the LabVIEW environment on a PC Windows operating system. The TEM-GTEM program requires prior installation of LabVIEW Run-Time Engine 8.6 – Windows 2000/Vista x64/Vista x86/XP on any computer which does not already have LabVIEW 8.6 installed. The Run-Time engine can be found on the CD-ROM in the folder: runtimeengine. The second folder, tem-gtem, contains the tem-gtem.exe as well as Installation.doc and Instructions.doc file. The application TEM-GTEM has two programs: TEM and GTEM. The first program, TEM, calculates the characteristic impedance Z0 (in ohms), and cutoff fre- quencies (fc) for modes TE01, TE10, TE11, TM11, TE02, TE12, TM12, and TE20 depend- ing on the TEM-cell dimensions. The second program, GTEM, calculates the cutoff (fc) and the associated stimulated resonant frequencies (fr1, fr2, and fr3) in mega- hertz for higher-order modes H10, H01, H11, H20, E11, and E21. The resonances are also shown graphically. A detailed explanation on the theory for the TEM cell and the GTEM cell can be found in Sections 11.4 and 11.5, respectively. I wish to thank Artech House editors Lindsey Gendall, Barbara Lovenvirth, and Mark Walsh for their encouragement in writing this book. Finally, I thank my wife Blazenka and my parents Marija and Vladimir for their love and support through- out this project.
  • 14. CHAPTER 1Communications Systems1.1 Components of Communications Systems A communication system usually consists of the information source, transmitter, channel, receiver, antenna systems, amplifier, and end user (Figure 1.1). It converts information into a format appropriate for the transmission medium. A transmitting antenna’s purpose is to effectively transform the electrical signal into radiation energy, whereas the receiving antenna’s purpose is to effectively receive the radiated energy and its electric signal transformation for further process- ing at the receiver. Communication systems can be either analog or digital. Historically, analog systems (Figure 1.2) are simpler but less resilient to interference. Analog communi- cation systems convert (modulate) analog signals into modulated signals. Signals that are analog are converted into digital bits by sampling and quantization, also called digitization and coding. Digital systems can reconstruct original information, are better protected from interference, and have the potential to code signals, thus enabling larger amount of data transportation. It is important that the information sent from the source and the information sent to the end user are similar as possible (i.e., identical in the case of digital infor- mation). Even though digital systems are used more for communications, analog systems will not be neglected in this and other chapters. With digital systems, there is a source and channel coder instead of signal pro- cessing in the transmitter, as is the case when dealing with analog systems. In the receiver, there are also channel and source decoders instead of the signal processing of analog systems. The modulator and demodulator should be designed to lessen the distortion and noise from the channel. The channel transports the signals using electromagnetic waves. There is always noise in the channel along with the useful signal. The source coder (Figure 1.3) converts the analog information to digital bits using analog to digital conversion (A/D). The transmitter converts the signal (ana- log) or bits (digital) into a format that is appropriate for channel transmission. Dis- tortion, noise, and interference are brought into the channel. The receiver decodes the received signal back into the information signal and then the source decoder decodes the signal back to the original information (analog or digital). 1
  • 15. 2 Communications Systems Source of Transmitter Receiver User information Channel Figure 1.1 Communication system. Transmitter in baseband Source Processing Modulator RF stage Channel User Processing Demodulator RF stage Receiver in baseband Figure 1.2 Analog communication system. Transmitter in baseband Source Channel Digital RF stage Source coder coder modulator Channel Digital RF stage User Decoder Decoder demodulator Receiver in baseband Figure 1.3 Digital communication system.1.2 Transmitter Systems A transmitter system is used to send information/data from one user to another. The source information can be analog or digital. Digital systems are usually more com- plex than the analog ones and require more modules. Both the transmitter and receiver systems should have the same complexity. For example, if we have a modu- lator on the transmitter side, there should be a demodulator on the receiver side. The same applies for the coder, multiplexer, and so forth. Usually, transmitter sys- tems have a multiplexer, randomization, encryption, an encoder, interleaving, a modulator, a mixer (upconverter), a power amplifier, a filter, a duplexer, and a
  • 16. 1.2 Transmitter Systems 3 transmitting antenna. Not all systems require all of the mentioned modules; this depends on the type of communication used and/or on the required security level. 1.2.1 Transmitter The multiplexer provides multiple dedicated channels to users and combines the data (bits) from every channel into one combined stream of data bits. These bits are organized into frames; a frame has a fixed length of bits. Every user is allocated a specific position in the frame. There must be a synchronization code or sequence of bits inside the frame in order to provide the ability to identify each frame and the rel- ative position of the bit inside the frame. There should also be a clock for the correct transmission of the data. In this way several signals can share one communication line, instead of having one line for every signal (Figure 1.4). Multiplexers (MUX) can range from two input signals up to sixteen or more. For more input signals, a cascade (consisting of simpler multiplexers) is used. Typi- cally they are 2/1, 4/1, 8/1, or 16/1, indicating the number of input signals and only one output signal. Figure 1.5 shows a 4/1 multiplexer. This means that four input signals share one communication line. The input signals are I0, I1, I2, and I3. The output is Z. Which signal will pass from input to output is decided on the basis of control signals a1 and a0 as shown in Table 1.1. Input E enables (E = 1) or disables (E = 0) the multiplexer. MUX DEMUX Conversation 1 Conversation 1 Conversation 2 Conversation 2 Conversation 3 Conversation 3 Conversation 4 Conversation 4 Conversation 5 Conversation 5 Figure 1.4 Use of one line for several communications. MUX 4/1 I0 I1 I2 Z I3 E a1 a0 Figure 1.5 Multiplexer 4/1.
  • 17. 4 Communications Systems Table 1.1 Combination Table for Multiplexer 4/1 E a1 a0 Z 0 x x 0 1 0 0 I0 1 0 1 I1 1 1 0 I2 1 1 1 I3 1.2.2 Randomization Randomization is used to ensure the even number of 0s and 1s in the data informa- tion, which should be randomly distributed. The process of randomization is car- ried out by the exclusive or adding (XOR), which adds a bit from a selected bit sequence to each bit within the multiplexer frame, except for the synchronization bits. The bit sequence that is used to randomize is called pseudorandom or pseudonoise (PN) sequence. The random distribution of the bit sequence matches the Gaussian distribution. This function happens simultaneously with each multiplexing frame. PN codes can be generated using a series of shift registers and logic gates in feedback as shown in Figure 1.6. There is also a modulo-2 adder (adding without carry). The shift registers receive a clock signal every Tc seconds. The feedback lines can be used to obtain dif- r ferent output codes. For r shift registers, a maximum of 2 − 1 bit sequence can be produced. This means that with four shift registers, a maximum of fifteen bit sequences can be achieved. After that, the combinations will start repeating them- selves. It is possible to connect the feedback gates to produce a shorter sequence, but shorter sequences are less random and will repeat more often. A circuit configured to produce the maximum sequence of nonrepeating bits for a given number of shift registers is called a maximal length PN code generator. The main characteristic of this maximal length code is that the Modulo 2 sum of any sequence with a shifted version of itself will produce another shifted version of the same sequence. All com- binations will appear only once except all the 0 combinations, as this state will cause no changes to occur in the shift register values or in the output. The number of 1s will always be 1 larger than the number of 0s, independent of the length of the code. PN signal out Modulo 2 adder Shift register 1 2 3 4 r Clock signal Figure 1.6 Pseudonoise generator.
  • 18. 1.2 Transmitter Systems 5 This type of PN generator is used in the transmitter to modulate a continuous wave signal, as well as in the receiver, where an identical PN generator is used to demodulate the received signal. 1.2.3 Encryption Encryption is used to protect the data should it be intercepted. Usually the bitstream is changed (encrypted) in such a way that it would be difficult to reconstruct the original bitstream without a decryption device. A problem develops if there is an error in the received bitstream, which results in an additional error in the decryption process. This is called error extension. Encryption has been used in wars and for information protection for a long time now. It can be used in computer systems and communication systems for authorization, copyright protection, and other applica- tions. For the encryption process, an encryption key is required. It is usually 40 to 256 bits long. The longer the key (cipher strength), the harder it is to break the code. There are two methods available: the secret and the public key. With the secret key, both sender and receiver use the same key to encrypt and decrypt the bitstream. This is the fastest method, but there is the problem of getting the secret key to the receiving side. With the public key, each recipient has a private key that is kept secret and a public key known to everyone. The sender uses the public key to encrypt the data, whereas the recipient uses the private key to decrypt the data. In this manner the private key is never transmitted, and thus is not vulnerable to inter- ception. The most spread encryption standards are the Data Encryption Standard (DES) and the Advanced Encryption Standard (AES). DES (Figure 1.7) is the most widely used encryption standard, dating from the 1970s. It has blocks of 64 bits at a time, and the key length is 56 bits. The 64 bits of the input block to be enciphered are first subjected to initial per- mutation. The permutated input block becomes the input to a complex key-depend- ent computation. The output of that computation, called the preoutput, is then subjected to permutation, which is the inverse of the initial permutation. The com- putation which uses the permuted input block as its input to produce the preoutput block consists of 16 iterations of a calculation depending on the cipher function. Today DES is considered insecure because a key of 56 bits is not long enough. That is why in 2002, AES was adopted, which is capable of processing data blocks of 128 bits using cipher keys with lengths of 128, 192, and 256 bits. More on AES can be found in “Announcing the Advanced Encryption Standard (AES),” which is free to download from the Internet [National Institute of Standards and Technol- ogy (NIST),]. 1.2.4 Encoder Encoders are used in transmitter systems for detection and correction of errors that may occur during transmission due to noise or interference. Coding can also be used for compressing the information. Most encoders add the redundant (known) bits expanding the data (information bits). This slows the traffic. How many redundant bits will be added depends on the surrounding of our communication service (inter- ference) and on the importance of the information being transmitted in real time.
  • 19. 6 Communications Systems Input (64 bits) Key (64 bits) Initialization Initial permutation Key permutation Left half (32 bits) Right half (32 bits) Left half (28 bits) Right half (28 bits) Cipher Binary rotation Binary rotation Round 1 function Subkey #1 (48 bits) Permutation Cipher Round 2 function Binary rotation Binary rotation Subkey #2 (48 bits) Permutation Round 16 Cipher function Binary rotation Binary rotation Subkey #16 (48 bits) Final permutation Finalization Permutation Output (64 bits) Figure 1.7 Data Encryption Standard algorithm. The differential encoder, convolutional encoder, Reed Solomon coding, and Golay encoder are most often used in communication systems. Differential Encoder Differential encoding of data is required for modulations such as duobinary and dif- ferential phase shift keying. These modulation types are used for optical links and high data rates of 10 to 40 Gbps. The principle of differential encoding is shown in Figure 1.8. dk c k = ck−1 d XOR 1 bit period c k−1 delay Figure 1.8 Differential encoder.
  • 20. 1.2 Transmitter Systems 7 Let dk be a sequence of binary bits that are the input to a differential encoder and let ck be the output of the differential encoder. Then we have c k = c k −1 ⊕ d k (1.1) where ⊕ is the modulo 2 addition. The direct implementation of the above equation is the use of an exclusive –OR (XOR) gate with a delay in the feedback path of 1 bit period delay. At 40 Gbps, 1 bit period is equal to 25 ps. Convolutional Encoder Information data is susceptible to errors. For useable data, there are methods of encoding information. This means organizing the 0s and 1s so that errors can be corrected. Convolutional encoding is applied to the data link signal in order to cor- rect bit errors that might occur during transmission, which results in coding gain for the system. Through the convolutional encoding/decoding process, the majority of transmission errors will be corrected before they are passed onto the decryption process. Codes have three primary characteristics: length, dimension, and minimum dis- tance of a code. The code’s length is the amount of bits per code word. The code dimension is the amount of actual information bits contained within each code word and the minimum distance is the minimum number of information differences between each code word. Convolutional codes are commonly specified by three parameters: (n, k, m) where n is the number of output bits, k is the number of input bits and m is the number of memory registers. The quantity k/n is called the code rate R and is a measurement of coding efficiency: k R= (1.2) n Commonly k and n parameters range from 1 to 8 and the code rate accordingly from 1/8 to 7/8. Memory registers, m, can range from 2 to 10. Another parameter, the constraint length K, is defined by: K = k ⋅ ( m − 1) (1.3) which represents the number of bits in the encoder memory that affects the genera- tion of n output bits. A convolutional encoder can be made with a K-stage shift register and n modulo-2 adders, where K is called the constraint length of the code. An example of such an encoder with K = 3 and n = 2 is shown in Figure 1.9. For each bit entering into the register, the output switch samples n = 2 code bits out (u1 and u2); hence the rate of the code k/n is 1/2. Each output code bit will be a function of the input bit (located in the leftmost stage of the register) plus two of the earlier bits (in the rightmost stages). The larger the constraint length K, the greater the number of past bits that have an effect on each output code word.
  • 21. 8 Communications Systems u1 First code bit Input bit Output m branch word Second u2 code bit Figure 1.9 Convolutional encoder: K = 3, rate = 1/2. Reed-Solomon Coding As with convolutional encoding, RS coding adds redundant bits and creates code words that enable the decoding process to correct errors. RS differs from convolutional encoding by performing block encoding (using bytes) rather than bitwise encoding. Because of the block encoding, RS is eight times faster than convolutional encoding. The incoming data stream is first packaged into small blocks, which are treated as a new set of k symbols to be packaged into a super-coded block of n symbols, by appending the calculated redundancy. Such symbols can either be comprised of one bit or of several bits (symbol code). There- fore, the information transfer rate is reduced by a factor called code rate (R), and the modulator is expanded by the ratio: 1 n = (1.4) R k A Reed-Solomon decoder can correct up to t symbols that contain errors in a code word, where 2t = n − k (1.5) A Reed-Solomon code word is generated using a special polynomial. All valid code words are exactly divisible by the generator polynomial. The general form of the generator polynomial is ( g( x ) = x − a1 )( x − a )K( x − a ) i+1 i+ 2t (1.6) The code word is constructed using: c( x ) = g( x )i( x ) (1.7) where g(x) is the generator polynomial, i(x) is the information block, c(x) is a valid code word, and a is referred to as the primitive element of the field.
  • 22. 1.2 Transmitter Systems 9 Golay Code Marcel J. E. Golay discovered the possible existence of a perfect binary (23, 12, 7) code, with error-correcting capability t = 3, that is, capable of correcting all possible patterns of three errors in 23 bit positions, at the most. So the Golay (23, 12, 7) code 12 is a perfect linear error-correcting code consisting of 2 = 4,096 code words of length 23 and a minimum distance of 7. Golay also defined the parity check matrix for this code as: H = ( MI11 ) (1.8) where I11 is the 11 × 11 identity matrix and M is a 11 × 12 defined matrix. Since the code’s length is relatively small (length = 23), the number of redundant bits is 11, and the dimension is 12, the Golay (23, 23, 7) code can be encoded by simply using look up tables (LUTs). A look up table is an array that holds a set of precomputed results for a given operation. This array provides access to results faster than com- puting the result of the given operation each time. Beside the perfect binary Golay code, there is the extended binary Golay code that encodes 12 bits of data in a word with a length of 24 bits, so that a triple-bit error can be corrected and a quadru- ple-bit error detected. 1.2.5 Interleaving Interleaving is used to intermix the bits of the code words generated through convolutional encoding. The motivation for interleaving is to compensate for burst or sequential errors, which can otherwise exceed the capability of the decoder to correct errors. Each code word generated through convolutional encoding can only correct a limited number of errors that occur in that code word. Sequential errors can cause multiple errors in a single code word, which can exceed the error-correct- ing capability of the decoding process. Interleaving distributes bits in such a way that, if sequential errors do occur, they will be distributed over multiple code words. For example, seven errors in a single code word will be distributed during interleav- ing into seven code words each having a single error. While the decoder may not be able to recover data in a code word with seven errors, it can easily recover a single error in seven code words. The disadvantage of interleaving is the delay created by writing a block of bits into memory, intermixing the bits, and then pulling the bits from memory. This delay is dependent on the number of bits that are interleaved at a time and the data rate of the aggregate bitstream. Interleaving is performed only on a finite block of bits at a time. Similar to multiplexing, interleaving requires framing the aggregate bitstream and adding synchronization bits. Interleavers are divided into periodic and pseudorandom. In periodic interleavers, symbols of the transmitted sequence are scrambled as a periodic func- tion of time. Periodic interleavers can be either block or convolutional.
  • 23. 10 Communications Systems 1.2.6 Modulation Modulation is the process of changing one or more parameters of an auxiliary sig- nal, depending on the signal that carries the information. This auxiliary signal is called the transmission signal. The signal that carries the information (and controls the parameter changes of the transmission signal) is called the modulation signal. The result of the modulation is the modulated signal. The process is performed in a device called the modulator, which converts the total digital bitstream into the radio frequency (RF) analog signal. The digital bitstream is usually modulated into the intermediate frequency (IF), which after amplification is upconverted to the transmit frequency. There are many analog and digital modulations that are used in communication systems. Analog modulations include: amplitude modulation (AM), frequency modulation (FM), phase modulation (PM), and several others. Digital modulations include frequency shift keying (FSK), phase shift keying (PSK), amplitude shift key- ing (ASK), quadrature amplitude modulation (QAM), pulse code modulation (PCM), and others. Chapter 6 will discuss more on modulation and demodulation. 1.2.7 Mixer (Upconverter) Mixers are used in transmitter systems for easier processing of the signal. It is much cheaper and easier to amplify the signal at a lower intermediate frequency (IF) than at a higher radio frequency (RF). The mixer inputs two different frequencies (one of them is a local oscillator fre- quency) and mixes them. The result is the sum and difference of the input signals. The frequency that is not needed must be filtered out. Figure 1.10 shows the mixer, which has a local oscillator frequency added to or subtracted from the input frequency. For upconversion of the frequency, the local oscillator frequency fLO is added to the input signal frequency fin: f out = f in + f LO (1.9) True systems mixers will produce more than just the sum and difference of the input signals. There will be intermodulation products from the input signals. If a second signal fin2 arrives at the input with the fin, the mixer will generate intermodulation products at its output due to inherent nonlinearity, in the form Mixer Input Output signal signal Local oscillator Figure 1.10 The mixer.
  • 24. 1.3 Receiver Systems 11 ± m ⋅ f in ± n ⋅ fi in2 (1.10) where m and n are positive integers, which can assume any value from 1 to infinity. The order of the intermodulation is defined as m + n. Accordingly, 2 fin − fin2, 2 fin2 − fin, 3 · fin and 3 · fin2 are third order products by definition. The first two products are called two-tone third-order products as they are generated when two tones are applied simultaneously at the input. Two-tone third-order products are very close to the desired signals and are very difficult to filter out. 1.2.8 Filter Filtering of the frequency range is an important part of every communication sub- system—hence the transmitter. Filtering is the ability to select the frequency range we wish to process and to block all other frequencies. Filters can be analog or digi- tal. Figure 1.11 shows the symbols used for lowpass, bandpass, and highpass filters, depending on which frequency range needs to be processed further. On lower frequencies, LC filters are used, and on higher frequencies (such as microwave) the microstrip is used. Filters will be discussed in more detail in Chapter 5.1.3 Receiver Systems The receiver system largely depends on the transmitter system. If in a communica- tion system a multiplexer is used on the transmitter’s side, there must be a demultiplexer on the receiving side. The same applies for other blocks mentioned in the previous section, which on the receiving side are placed in reverse order. Some receivers must deal with very small signals. Better and more expensive receivers will introduce very little noise themselves. Again, depending on the complexity of the communication system, the following blocks are optional: filter, downconverter, demodulator, deinterleaver, decoder, derandomizer, and demultiplexer. 1.3.1 Filter The filter is part of every receiving system. It selects the frequency band of use to be processed further and it stops signals on all other frequencies. The selectivity of the filter is shown by Q factor which can be calculated as fc Q= (1.11) f 2 − f1 Figure 1.11 Lowpass, bandpass, and highpass filters.
  • 25. 12 Communications Systems where f2 and f1 are frequencies where the power drops by 50% (3 dB), and fc is the central or resonant frequency as shown in Figure 1.12. 1.3.2 Mixer (Downconverter) On the receiving end of the communication system there is also a mixer, which in this case serves as a downconverter. It is necessary to downconvert the received RF frequency because it is much easier to amplify the signal at intermediate frequencies (IF) than at RF frequencies. Here, again, the local oscillator is necessary, and the output frequency is obtained as f out = f in − f LO (1.12) where the input frequency fin must be greater than fLO; otherwise an error will occur. The other result, that is, the adding of the two frequencies, will be filtered out. Again, intermodulation products may occur here. That is why it is necessary to take into consideration all possible transmitters in the vicinity (depending on the applica- tion, this can be up to 50 km) and calculate the intermodulation in order to deter- mine whether additional filtering is required. 1.3.3 Demodulator The demodulator converts an analog RF signal into a digital bitstream. It extracts the original information from the modulated carrier wave. There are different types of demodulation, such as envelope detection, differential, coherent, and synchro- nous demodulation. Demodulation and demodulators will be discussed in greater detail in Chapter 6. 1.3.4 Deinterleaver If a bitstream was interleaved in the transmission process, deinterleaving is required in the receiving process to reassemble the code words created by the encoder. Should any errors have occurred before deinterleaving, they will be distributed depending on the selected algorithm. Figure 1.13 shows the deinterleaving of an array of three element structures. Synchronization bits must be present to recognize when one frame is finished and the other is starting. Bandwidth −3dB f1 fc f2 Figure 1.12 Selectivity of the filter.
  • 26. 1.3 Receiver Systems 13 X[0] A X[0] B X[0] C X[1] A X[1] B X[1] C X[2] A X[2] B X[2] C X[3] A X[3] B A 3 A 2 A 1 A 0 Y0 X[3] C B 3 B 2 B 1 B 0 Y1 C 3 C 2 C 1 C 0 Y2 Figure 1.13 Deinterleaving an array of three element structures. 1.3.5 Decoder The decoder in the receiver system must match the encoder that was used in the transmitter system. Differential Decoder If differential encoding was used in the transmitter system, differential decoding must be performed in the receiving system. The differential encoding process does not introduce redundant bits, but transforms the waveform by converting the space signal (zeros) into transitions. Accordingly, the decoding process converts transi- tions back to spaces. Since a single bit error affects two transitions, the differential decoding process doubles any bit error, corresponding to a 3-dB loss to the system. The decoder decodes the binary input signal. The output is the logical difference between present and previous input. The input and the output are related with m(t 0 ) = d (t 0 ) XOR initial condition parameter value (1.13) m(t k ) = d (t k ) XORd (t k −1 ) where d is the differentially encoded input, m is the output message, tk is the kth time step, and XOR is the logical exclusive-or operator.
  • 27. 14 Communications Systems Viterbi Decoder The Viterbi decoder is used together with convolutional coding, and is applied to the data link signal to correct errors that may have occurred during the RF transmis- sion. The encoding/decoding process adds what is referred to as coding gain, which may be necessary for the successful data link transmission. The decoding corrects errors before the decryption process of total bitstream. Correction of the errors occurs because the convolutional encoder (or transmit side of the data link) creates code words, which contain data bits with added redundant bits. The redundant bits allow the decoder to detect and correct errors that may exist in each code word. The process of decoding is much more complicated than the encoding process, and limits the speed of the bitstream that needs to be decoded. If the convolutional code uses 2n possible symbols, the input vector length is K · n for positive integer K. If decoded data uses 2k possible output symbols, the output length will be K · n. The integer number K is the number of frames processed in each step. The entry into the decoder input can be a real number (positive real is logical zero, while negative real is logical one), 0 and 1 (0 is logical zero, 1 is logical 1). The latter is called a hard decision. The third possible input is a soft decision. It can be any integer between 0 and 2b − 1, where b is the number of the soft decision bit b parameter. Here 0 is the most confident decision for logical zero, 2 − 1 is the most confident decision for logical one, and other values are less confident decisions. Table 1.2 shows the decisions for three bits. Reed-Solomon Decoding The Reed-Solomon encoder and decoder are commonly used in data transmission and storage applications, such as: broadcast equipment, wireless LANs, cable modems, xDSL, satellite communications, microwave networks, and digital TV. The block diagram of the RS decoder is shown in Figure 1.14. The received code word r(x) is the original (transmitted) code word c(x) plus additional errors: r( x ) = c( x ) + e( x ) (1.14) Table 1.2 3-Bit Soft Decision Input Value Decision 0 Most confident zero 1 Second most confident zero 2 Third most confident zero 3 Least confident zero 4 Least confident one 5 Third most confident one 6 Second most confident one 7 Most confident one
  • 28. 1.3 Receiver Systems 15 r(x) Si Xi Syndrome Error L(x) Error Error Error c(x) Input calculator polynomial locations magnitudes corrector Yi Output v Figure 1.14 Reed-Solomon decoder. The decoder will try to identify the position and the magnitude of maximum t errors (or 2t erasures) and correct the errors and erasures. A Reed-Solomon code word has 2t syndromes (Si) that depend only on errors and not on the transmitted code word. The syndrome is calculated by substituting the 2t roots of the generator polynomial g(x) into r(x). To find the symbol error location, solving simultaneous equations with t unknowns is necessary. First, the error locator polynomial (L(x)) is found using the Berklekamp-Massey or Euclid’s algorithms, with v being the num- ber of errors. The roots of the polynomial [i.e., the error locations (Xi)], are found with the Chien search algorithm. Next, the symbol error values (Yi) are found using the Fornay algorithm. Golay Decoder The Golay coding can detect up to four bit errors in 24 bits (12 information bits) and correct up to three bit errors in 24 bits. If in 24 received bits there are three or less errors, the Golay decoding algorithm will detect the errors and correct them. If four errors appear, they will be detected but the exact pattern will not be deter- mined. An error message will be displayed. If there are more than four errors, the Golay decoding will not provide the actual error pattern, and the information in the 12 bits will be lost. 1.3.6 Decryptor Decryption is the process that reconstructs the original signal, which was altered through the encrypter in the transmitter. Decryption is required in the receiver sys- tem only if the encrypter was used in the transmitting system. Encryption is used as a protection means from signal interception. Error extension is possible during decryption (where multiple errors will be added for every error bit received). For decryption of the Data Encryption Standard (DES), the same encryp- tion algorithm (Figure 1.7) is used, with the same key, but reversed key schedule (16, ..., 1). 1.3.7 Derandomizer When randomization is used in the transmitting system, a derandomization of the data bitstream must be done in the receiving system. If synchronization bits are not randomized, they do not need to be derandomized. Synchronization bits identify the multiplexing frame, which is derandomized by Modulo 2 adding the same PN sequence that was used to randomize the frame to the bits within the frame.
  • 29. 16 Communications Systems 1.3.8 Demultiplexer The demultiplexer (Figure 1.15) is a device that receives data from one input and distributes it on 2n possible outputs, where n is the number of control bits. Table 1.3 shows the combination table for the demultiplexer 4/1. The data c coming to input “Z” will be distributed to four outputs I0, I1, I2, and I3 according to the controlling combinations of a1 and a0. All other outputs will have 0 as the output value. The demultiplexer recreates the user channels from the total bitstream. The bitstream is organized into multiplexing frames with a fixed bit length. Every user channel is allocated a specific bit position inside the frame. Inside the frame there are synchronization bits, which are used in the demultiplexing process, in which the bits are distributed to the appropriate user channel. This is not all that has to be thought of when considering the receiver system. Received power, sensitivity, required ratio of signal to noise, and noise factor are just some of the important parameters that have to be taken into consideration when planning a communication link. 1.3.9 Received Power Received power (Pr) at the receiving point is calculated using the effective area of an antenna (λ2/4π) and power density (Pt /4 · π · d2) as 2 λ2 Pt ⎛ λ ⎞ Pr = ⋅ = Pt ⎜ ⎟ (1.15) 4⋅ π 4⋅ π ⋅ d 2 ⎝4⋅ π ⋅ d⎠ where Pr is the received power, d is the distance from the transmitter to the receiver, Pt is the transmitted power, and λ is the wavelength of the signal. There are transmit- ting and receiving antenna gains Gt and Gr, for the antenna so the previous expres- sion can be written as 2 ⎛ λ ⎞ Pr = Pt ⋅ Gr ⋅ Gt ⎜ ⎟ (1.16) ⎝4⋅ π ⋅ d⎠ DEMUX 1/4 I0 I1 Z I2 I3 a1 a0 Figure 1.15 Demultiplexer 1/4.
  • 30. 1.3 Receiver Systems 17 Table 1.3 Combination Table for Demultiplexer 1/4 a1 a0 Z I0 I1 I2 I3 0 0 c 0 0 0 c 0 1 c 0 0 c 0 1 0 c 0 c 0 0 1 1 c c 0 0 0 Receiver Sensitivity Receiver sensitivity is the minimum level of signal at the input of the receiver, which is required to achieve a sufficient level of signal-to-noise ratio for the demodulation. The sensitivity is determined with thermal noise Pterm, required ratio of signal to noise (S/N)req for demodulation, and noise factor (NF) as ⎛S⎞ Pr min = Pterm ⋅ ⎜ ⎟ ⋅ NF (1.17) ⎝ N ⎠ req Receivers have the lowest level of signal strength required to process the infor- mation without loss of data. With digital systems, a lower received signal strength will result in a lower rate of received information. Typically receivers have a sensi- −9 −13 tivity ranging from −60 to −94 dBm (10 to 4 × 10 W). Thermal Noise The thermal noise of the receiver is defined as Pterm = k ⋅ T ⋅ B (1.18) −23 where k is the Boltzmann constant 1,38⋅10 J/K, T is the temperature in Kelvin (290–300K), and B is the frequency range width in hertz. The density of the thermal noise at room temperature (290K) is 204 dBw/Hz. The width of the frequency chan- nel B is determined by the receiving filter width. Required ratio signal to noise in the receiver is the ratio of signal to noise required for a certain quality of the link (i.e., relative number of bits or frames with errors). The ratio of signal to noise is the difference between received signal and noise: S N[dB] = 10 log( S ) − 10 log( N ) (1.19) For analog systems, the S/N ratio must always be above zero. In digital systems (spread spectrum), the signal can be buried in the noise. The higher the bit rate, the larger the signal to noise ratio must be. Noise Factor The noise factor of the receiver (NF) is the ratio of signal to noise at the input and output of the receiver:
  • 31. 18 Communications Systems S N in NF = (1.20) S N out This ratio can be from a fraction of a decibel for low noise microwave convert- ers [0.3 dB for low noise block downconverters (LNB) for satellite applications] to 30 or 40 dB for spectral analyzers; typically the ratio ranges from 2 to 10 dB. This is actually the noise the receiver itself introduces into the system. The noise threshold is the sum of the thermal noise and the noise factor.1.4 User Interface The user interface is the means for people to interact with the communication sys- tem. It consists of input of some sort and output. The input can be a command using a keyboard, voice, or text. We have heard of the phrase “user-friendly,” which means that it is simple to operate a certain device. When designing an application, a lot of care is taken to make a suitable user interface. Nowadays, there are hearing aids and other tools available for individuals with a handicap. Normally, the user interface is graphical (GUI), but it can also be operating via voice or touch. 1.4.1 Graphical User Interface (GUI) The graphical user interface interacts with electronic devices through icons or visual indicators. Touch screens are one of the GUI types. There are also the command line and text user interfaces, which use a keyboard to type the commands. Main touchscreen technologies are resistive and capacitive. Resistive LCD touchscreen monitors rely on a touch overlay, which is composed of a flexible top layer and a rigid bottom layer separated by insulating dots attached to a touchscreen controller. The inside surface of each of the two layers is coated with a transparent metal oxide coating that facilitates a gradient across each layer when voltage is applied. Pressing the flexible top sheet creates electrical contact between the resistive layers, and closes a switch in the circuit. The control electronics alternate voltage between the layers and pass the resulting X and Y touch coordinates to the touchscreen control- ler. The touchscreen controller data is then passed on to the computer operating sys- tem for processing. Capacitive touch screens work by placing a very small charge at each of the four corners of the screen. When a finger touches the screen, the touch controller determines the change of capacitance of the screen from each of the four points and provides a touch value at the correct location. Surface acoustic wave touch screen technology is based on sending acoustic waves across a clear glass panel with a series of transducers and reflectors. When a finger touches the screen, the waves are absorbed, causing a touch event to be detected at that point. Because the panel is all glass, there are no layers that can be worn, which results in durability. Infrared technology is based on the interruption of an infrared light grid in front of the display screen. The touch frame contains a row of infrared LEDs and photo transistors, each mounted on two opposite sides to create a grid of invisi- ble infrared light.
  • 32. 1.5 Antenna Systems 19 1.4.2 Voice User Interface (VOI) The voice user interface can be activated through speech. Today hands-free com- mands are possible. The possibility of error when inputing a command or data by voice is higher than when entering it through a keyboard. Figure 1.16 shows a possi- ble user interface for a communication device.1.5 Antenna Systems Antenna systems consist of a duplexer and an antenna used to transmit and receive information from one user to another. There are many types of antennas that can be used for a communication system, depending on the frequency of use, power, appli- cation, and even international standard regulations. Most communication systems use the same antenna for transmitting and receiving a signal. This normally requires two antennas, which have to be physically separated. This is impractical, except in some cases of high interference when antenna diversity could be an option. That is why in most cases a single antenna is used for both transmitting and receiving the signal. This is possible with the use of a duplexer. 1.5.1 Duplexer The duplexer makes it possible for receiver and transmitter systems to use the same antenna; otherwise it would be necessary to use two antennas. The duplexer has fil- ters, which isolate the transmitting frequency from the receiving frequency. Since the transmitting and receiving frequency are usually not the same (because of inter- ference), there must be a separation between them. The duplexer must be designed to operate in the frequency band used by both the receiver and the transmitter. It also must be able to operate on the power from the power amplifier. When working at the transmitting frequency, it must reject the noise from the receiver and vice versa. The duplexer can be made with a hybrid ring, cavity notch, and a band-pass/ band-reject design. Figure 1.17 shows the design of a reject duplexer using notch cavities. Audio Signal Transmitter processor processor Screen User Communication interface interface Keyboard Figure 1.16 User interface.
  • 33. 20 Communications Systems To antenna Optimal cable length Cavity tuned to Cavity tuned to transmitter frequency receiver frequency To transmitter To receiver Figure 1.17 Reject duplexer with notch cavities. Using only two notch cavities would probably not provide sufficient isolation for most situations. The cavity in the transmitter is tuned to the receiver frequency, and the cavity in the receiver is tuned to the transmitter frequency. That means that the cavity in the transmitter area will pass the transmitting frequency and notch (reject) the receiving frequency. The same applies for the cavity in the receiver area, which will pass the receiver frequency and notch the transmitter frequency. If there are other strong signals present, this design will not be enough. A more advanced design must include four to six cavities. Two or three cavities in each leg are more effective than just one. Usually the cavities require tuning with a spectrum analyzer or wattmeter. 1.5.2 Antenna The antenna is a device that transforms a guided electromagnetic wave from the transmission line (waveguide or cable) into a space wave in free space. The antenna actually makes a transition between the guided wave in the transmission line and the space wave in free space. The most important characteristics of an antenna are: radiation pattern, directivity, impedance, gain, and affective area. Radiation Pattern Electric field intensity falls with 1/d, where d is the distance from the antenna. To measure the electric or magnetic field from the antenna, we have to be far enough from the antenna (only the radiating field exists). This happens at the distance d, 2D 2 d = (1.21) λ where D is the largest dimension of the antenna and λ is the wavelength of the sig- nal. Then, knowing the electric field E, the magnetic field H can be calculated from E H= (1.22) η
  • 34. 1.5 Antenna Systems 21 where η is the impedance of free space, that is 120π or 377Ω. Usually the radiation pattern is given in two perpendicular planes: horizontal and vertical; it usually has one main lobe and several sidelobes. Directivity Often the goal of an antenna is to have most of the radiation in just one direction with much less radiation in other directions. The directivity angle is calculated as the angle where the power density is one half of the maximum and the field density drops for a factor of 1/ 2. Directivity is defined as the ratio of the power density radiated by the antenna in the direction of maximum intensity and the power den- sity radiated by the isotropic radiator. The isotropic radiator radiates equally in all directions. Antennas with higher directivity are used to radiate as much energy to the receiver as possible. At the same time, dispersion of the signal in unwanted directions is diminished, thus making the interference to other systems smaller. Antenna Impedance Antenna impedance is the ratio of the voltage and current at the antenna. The most power from the generator will be given to the antenna if the antenna and generator impedances are complex conjugates (i.e., Z a = ZG ). That means that their * resistances must be equal, whereas their reluctances must be equal in magnitude but of opposite signs. Usually generators have an output impedance of 50Ω or 75Ω, so the antenna will have to be of the same impedance if possible. Gain Gain is related to the power received from the generator and represents the number showing how much larger the power from the isotropic radiator must be compared to the received power of the antenna, in order for the radiation from the isotropic radiator to be the same as the radiation from the observed antenna in the direction of maximum radiation. For an ideal antenna without losses, the gain would be equal to directivity. Gain of the antenna is usually given in decibels. Effective Area The effective area of a receiving antenna, Aeff, is defined as the ratio of received power, Pr, absorbed on a matched load connected to the antenna, and power den- sity of the incident electromagnetic wave, Sr: Pr A eff = (1.23) Sr The power density of the transmitting antenna in the maximum direction is equal to
  • 35. 22 Communications Systems Gt Pt Sr = (1.24) 4⋅ π ⋅ d 2 where Gt and Pt are gain of the transmitting antenna and power of the transmitting antenna. The relation that connects the effective area and gain for all antennas is λ2 A eff = ⋅G (1.25) 4⋅ π Antenna Types There are many types of antennas, depending on the type of the application needed. They can be divided into four groups: electrically small antennas, wideband anten- nas, resonant antennas, and aperture antennas. Electrically small antennas are much smaller in dimension than the wavelength associated with the frequency on which they operate. They have small directivity and radiation effectiveness. They include Hertz’s dipole and monopole. To increase directivity, antenna arrays can be built. By changing the phase of the supplying cur- rents, different radiation patterns can be obtained. Wideband antennas have a stable radiation pattern, gain, and impedance in the wide frequency range. The gain is small to medium. The biconical antenna and log-periodic antenna are examples of this type of antenna. Resonant antennas oper- ate in one or more selective frequency ranges. They have a small to medium gain. The microwave microstrip antenna is a resonant antenna. Aperture antennas receive and radiate electromagnetic waves through an aperture. They have large gain, which increases with the frequency. The horn antenna and parabolic dish are examples of this type of antenna. Smart Antenna Systems A smart antenna system uses multiple antenna elements including signal processing to optimize its radiation pattern depending on the signal environment. The smart antenna interference is smaller, which enables reuse of the frequency more often. This can also improve the capacity of the link. Greater signal gain will result in lower power requirements at the receiving system with a smaller size and battery. The power amplifier used can be cheaper, with less total power consumption.1.6 Power Supplies Power in communication systems is necessary for the operation of electronic com- ponents. For simple systems, a DC supply is sufficient. For the high power of a transmitter, a power amplifier is necessary. The transmitter system usually requires more power than the receiving system. The majority of power is used to amplify the signal before reaching the antenna. In calculating the communication link, a free space loss must also be taken into consideration. In addition, cable loss and match-
  • 36. 1.7 Considerations for Voice Versus Data 23 ing losses require the transmitted power to be raised. For small transmitting power, a simple 12-V DC supply is sufficient. Larger power requires power amplifiers. 1.6.1 Power Supply Types Linear power uses a transformer to convert the voltage from the mains to a lower voltage. Converters, which transform 120-V or 220-V AC into a lower DC voltage (typically 12V or 24V), are often used for electronic circuits. There are many types available. An uninterruptible power supply (UPS) must be used in applications for which a constant power supply is necessary. UPS usually takes the power from the AC mains and charges its own battery at the same time. If there is a loss of power, the battery will provide the necessary power for some time. There are solutions where the UPS charges a battery with energy generated from internal combustion engines or turbines. Batteries are also often used for mobile communications. In some situations solar power might be used, especially in areas with a lot of sun. 1.6.2 Power Amplifier Power amplifiers are used to increase the level of the signal, both in transmitter and receiver systems. They are used to amplify the low-level signal to a higher value. Power gain is described as ⎛P ⎞ G(dB) = 10 log10 ⎜ out ⎟ (1.26) ⎝ Pin ⎠ where Pin is input power and Pout output power. Power amplifiers can be divided in classes A, B, AB, C, D, and E. Class A uses 100% of the input signal. This amplifier is inefficient and is used for small signals or low power amplification. Class B uses 50% of the input signal. It is more efficient than class A, but subject to signal distortions. Class AB is a combination of class A and class B. It uses more than 50% of the signal. Class C uses less than 50% of the signal. Distortions are high, but so is the efficiency. Class D uses switching (on/off) for high efficiency. It can be used in digital circuits. There are also some other special classes.1.7 Considerations for Voice Versus Data Input information into the communication system can be voice or data (text, pic- tures, video, and so forth). In this section just some of the codecs are mentioned. There are many more; some of them are obsolete, while others are being developed. 1.7.1 Text ASCII uses 7 bits per character. Extended ASCII uses 8 bits per character.
  • 37. 24 Communications Systems Table 1.4 Data Rates for Audio Codecs ADPCM G.711 G.729a Sample Rate 8 KHz 8 KHz 8 KHz Effective Sample Size 8 bits 4 bits 1 bit Data Rate 64 Kbps 32 Kbps 8 Kbps 1.7.2 Images The Graphics Interchange Format (GIF) (lossless compression) uses 8 bits per pixel and a 256 color palette. The Joint Photographic Exchange Group (JPEG) (lossy compression) format most often uses a 10:1 compression. 1.7.3 Voice Pulse code modulation (PCM) has 8,000 samples per second—with 8 bits per sec- ond it results in 64 Kbits per second. Compression techniques are adaptive differen- tial pulse code modulation (ADPCM) (32 Kbps) and residual excited linear predictive coding (8–16 Kbps). Audio music requires 32–384 Kb/s. The audio signal is sensitive to delay and jitter. Latency is the end-to-end delay from mouth to ear. It must not exceed 100 ms for excellent quality. For acceptable quality it should not exceed 150 ms. For higher delays an echo canceller is required. There is a propagation delay in free space, which depends on the frequency used and the distance between the transmitter and receiver. Packetization delay is the time required to create an audio packet and send it on a network—it depends on the codec. Table 1.4 gives the data rates for some audio codecs. The G.711 codec used in telephony works at 64 Kbps. 1.7.4 Video H.261 coding uses 176 by 144 or 352 by 258 frames at 10–30 frame/sec. MPEG-2 and HDTV use 1,920 by 1,080 frames at 30 frames/sec.Selected Bibliography Balanis, C. A., Antenna Theory—Analysis and Design, New York: John Wiley & Sons, 2005. Brown, S., and Z. Vranesic, Fundamentals of Digital Logic with VHDL Design, New York: McGraw-Hill, 2001. Couch, L. W., Digital and Analog Communication Systems, 6th ed., Upper Saddle River, NJ: Prentice-Hall, 2001. Dunlop, J., and D. G. Smith, Telecommunication Engineering, London, U.K.: Chapman and Hill, 1994. Diffie, W., and M. E. Hellman, “Privacy and Authentification: An Introduction to Cryptogra- phy,” Proceedings of the IEEE, Vol. 67, No. 3, March 1979, pp. 397–428. Hanna, S. A., “Convolutional Interleaving for Digital Radio Communications,” Proc. 2nd Inter- national Conference on Personal Communications: Gateway to the 21st Century, 1993, Vol. 1, pp. 443–447.
  • 38. 1.7 Considerations for Voice Versus Data 25 Federal Information Processing Standards Publications 197: “Announcing the Advanced Encryp- tion Standard (AES),” Gardiol, F. E., Introduction to Microwaves, Dedham, MA: Artech House, 1984. Morelos-Zaragoza, R. H., The Art of Error Correcting Coding, New York: John Wiley & Sons, 2006. Sklar, B., Digital Communication: Fundamentals and Applications, Upper Saddle River, NJ: Prentice-Hall, 2001. Xiong, F., Digital Modulation Techniques, Norwood, MA: Artech House, 2000.
  • 39. CHAPTER 2Electromagnetic Spectrum Used forCommunications2.1 Electromagnetic Spectrum The electromagnetic (EM) spectrum of an object is the distribution of electromag- netic radiation from that object. The EM spectrum (Figure 2.1) covers frequencies from 3 Hz (ELF) to gamma rays (30 ZHz) and beyond (cosmic rays). The corresponding wavelengths λ can range from thousands of kilometers to a fraction of an atom size (Table 2.1). The frequency and the wavelength are related by the following expression: c λ= (2.1) f where c is the speed of light—approximately 30,000,000 m/s. The energy of the particular range is defined as E = h⋅ f (2.2) where f is the frequency in hertz and h is the Planck’s constant, 6.62606896e−34 Js. Energy can be expressed in eV, where 1 eV is approximately 1.60217653e−19 J. One eV is equal to the amount of energy gained by a single unbound electron when it accelerates through an electrostatic potential difference of 1 volt. It is also the energy needed to break the chemical bond in the cell. The higher the frequency, the higher the energy in each photon (Table 2.1). Table 2.2 gives the prefix converters used in Table 2.1. The spectrum is divided in decades. The radio spectrum (including microwaves) is considered to cover frequencies from 9 kHz to 300 GHz, that is, from VLF to SHF. Most communications take place in the radio spectrum, but the infrared and the visible spectrum can be used as well. The use of frequency bands for communi- cation is discussed latter in Sections 2.1.1 to 2.1.13. 27
  • 40. 28 Electromagnetic Spectrum Used for Communications SOFT HARD SOFT HARD Visible ELF SLF ULF VLF LF MF HF VHF UHF SHF EHF IR UV X-ray Gamma-rayFigure 2.1 Electromagnetic spectrum.Table 2.1 Electromagnetic SpectrumRange Frequency Wavelength Energy (eV) 8 7 −14 −13Extremely low frequency (ELF) 3 Hz–30 Hz 10 m–10 m 1.24 · 10 –1.24 · 10 7 6 −13 −12Super low frequency (SLF) 30 Hz–300 Hz 10 m–10 m 1.24 · 10 –1.24 · 10 6 5 −12 −11Ultra low frequency (ULF) 300 Hz–3 kHz 10 m–10 m 1.24 · 10 –1.24 · 10 5 4 −11 −10Very low frequency (VLF) 3 kHz–30 kHz 10 m–10 m 1.24 · 10 –1.24 · 10 4 3 −10 −9Low frequency (LF) 30 kHz–300 kHz 10 m–10 m 1.24 · 10 –1.24 · 10 3 2 −9 −8Medium frequency (MF) 300 kHz–3 MHz 10 m–10 m 1.24 · 10 –1.24 · 10 2 1 −8 −7High frequency (HF) 3 MHz–30 MHz 10 m–10 m 1.24 · 10 –1.24 · 10 1 −7 −6Very high frequency (VHF) 30 MHz–300 MHz 10 m–1m 1.24 · 10 –1.24 · 10 −1 −6 −5Ultra high frequency (UHF) 300 MHz–3 GHz 1m–10 m 1.24 · 10 –1.24 · 10 −1 −2 −5 −4Super high frequency (SHF) 3 GHz–30 GHz 10 m–10 m 1.24 · 10 –1.24 · 10 −2 −3 −4 −3Extremely high frequency (EHF) 30 GHz–300 GHz 10 m–10 m 1.24 · 10 –1.24 · 10 −3 −9 −3Infrared (IR) 0.3 THz–400 THz 10 m–750 · 10 m 1.24 · 10 –1.65 −9 −9Visible 400–790 THz 750 · 10 m–380 · 10 m 1.65–3.27 −9 −9Ultraviolet (UV) 750 THz–30 PHz 400 · 10 m–10 · 10 m 3.10–124 −9 −9X-ray 30 PHz–30 EHz 10 · 10 m–0.01 · 10 m 124–124,000 −9 −15 3Gamma ray 30 EHz–30 ZHz 0.01 · 10 m–10 · 10 m 0.124–124 · 10 Table 2.2 Prefix Converters Symbol Z E P T G M k m µ n p f Prefix zetta exa peta tera giga mega kilo milli micro nano pico femto 21 18 15 12 9 6 3 −3 −6 −9 −12 −15 Factor 10 10 10 10 10 10 10 10 10 10 10 10 2.1.1 Extra Low Frequency (ELF) Most sources in the ELF band are natural or accidental. However, ELF can be used for submarine communications, since signals with a transmitter power of 100 MW can penetrate up to several hundred meters deep. However, the messages are very short. 2.1.2 Super Low Frequency (SLF) The SLF band, like ELF, can be used for submarine communications. Unwanted sources can occur from power lines (50 or 60 Hz), which run for kilometers. This signal is called hum. Another natural source example is the interaction of solar wind with the ionosphere.
  • 41. 2.1 Electromagnetic Spectrum 29 2.1.3 Ultra Low Frequencies (ULF) Along with ELF and SLF, the ULF band can also be used for submarine communica- tions. ULF is used in mines as well. Only slow modulation can be applied (Morse code), limiting the amount of information. If only the phase is required, as is the case with navigation systems, a limited amount of information is not a disadvantage. 2.1.4 Very Low Frequency (VLF) Similar to ULF, VLF is used for navigation systems and communication over large distances. The information capacity is small with VLF. Communication with sub- marines is possible only near the surface. Lightning also happens in this band. Fre- quencies below 9 kHz are not allocated by the International Telecommunication Union and can be used freely for communications in some countries. 2.1.5 Low Frequency (LF) Communication in this band is possible around the Earth by refraction from the ionosphere and reflection from the Earth’s surface. It can be used for navigation, AM radio, and radio frequency identification (RFID). 2.1.6 Medium Frequency (MF) Like the LF band, MF can use refraction from the ionosphere—but only at night. It is used for AM radio, amateur radio, and navigation. 2.1.7 High Frequency (HF) The HF band is also known as the short wave band. It is used for medium- and long-range communications, such as marine and aviation communications, ama- teur radio, and RFID. More information can be sent in channels in the HF bands than in the previously described bands. 2.1.8 Very High Frequency (VHF) The VHF band is used for radio (FM) and television at short distances (little more than line of sight (LOS). VHF antennas are usually one quarter or one half wave- length long. VHF can also be used for land mobile communications, radio astron- omy, cordless telephones, amateur radio, navigation, satellite communications, and railways. 2.1.9 Ultra High Frequency (UHF) UHF is used for television, mobile phones, satellites, radar, RFID, the global posi- tioning system (GPS), Bluetooth, WLAN, and so forth. The communication is point-to-point over line-of-sight (LOS). For larger distances, a repeater is necessary.
  • 42. 30 Electromagnetic Spectrum Used for Communications UHF is strongly affected by rain. The antenna size in this frequency range is about a wavelength. 2.1.10 Super High Frequency (SHF) SHF is used for satellite communications, microwave links, and radar. It is used for line-of-sight communications. 2.1.11 Extra High Frequency (EHF) The EHF band is mostly used for satellite communications, but not yet for other types of communications as it is hard to modulate and demodulate high frequencies on the band. 2.1.12 Infrared (IR) IR is used for short-range wireless communications and in astronomy. Computers, PDAs, and remote controls use Infrared Data Association (IrDA) technology. Devices must be in line-of-sight (LOS) and the data transmitted must be short. 2.1.13 Visible Optical fiber is suitable for large distances because light propagates with little atten- uation. It is used for a large amount of data traffic. Visible light communications (VLC) is a new technology that uses light that is visible to human eyes. It must be line-of-sight and suffers from interference from other light sources.2.2 Spectrum Division The International Telecommunication Union (ITU) is the leading United Nations’ agency for information and communication technologies. It has three sectors: radio communications, standardization, and development. ITU manages international radio frequencies, allocating the spectrum and frequencies in order to avoid inter- ference between radio stations of different countries. In recent years, radio commu- nication systems have expanded largely. The radio frequency spectrum is a natural resource, and its allocation has to be planned well ahead. Apart from the traditional division shown in the previous section, there are a number of other divisions, such as radar, satellite, and military frequency band des- ignations, given in Tables 2.3–2.8. The bands for TV receive only (TVRO) are given in Table 2.4. TVRO is a satel- lite technology for receiving satellite TV programs from fixed service satellites. Military secret radar bands originate from World War II and were used for radars. After the war, the secrecy was lifted. IEEE adopted the codes, and today they are in use in radar, satellite, countermeasures, and terrestrial communications. ITU bands are subbands of military designations.
  • 43. 2.2 Spectrum Division 31 Table 2.3 IEEE Radar Band Designations Bands (According to IEEE Standard 521-2002) Frequency Wavelength Band 3–30 MHz 100–10m HF 30–300 MHz 10–1m VHF 300–1000 MHz 100–30 cm UHF 1–2 GHz 30–15 cm L 2–4 GHz 15–7.5 cm S 4–8 GHz 7.5–3.75 cm C 8–12 GHz 3.75–2.50 cm X 12–18 GHz 2.5–1.67 cm Ku 18–27 GHz 1.67–1.11 cm K 27–40 GHz 11.1–7.5 mm Ka 40–75 GHz 7.5 mm–4 mm V 75–110 GHz 4 mm–2.73 mm W 110–300 GHz 2.73 mm–1 mm mm 300–3,000 GHz 1 mm–100 µm µm Table 2.4 Satellite TVRO Band Designations Frequency Band 1.7–3 GHz S 3.7–4.2 GHz C 10.9–11.75 GHz Ku1 11.75–12.5 GHz Ku2(DBS) 12.5–12.75 GHz Ku3 18.0–20.0 GHz Ka Table 2.5 Military Electronic Countermeasures Band Designations (NATO) Frequency Band 30–250 MHz A 250–500 MHz B 500–1,000 MHz C 1–2 GHz D 2–3 GHz E 3–4 GHz F 4–6 GHz G 6–8 GHz H 8–10 GHz I 10–20 GHz J 20–40 GHz K 40–60 GHz L 60–100 GHz M
  • 44. 32 Electromagnetic Spectrum Used for Communications Table 2.6 Traffic Radar Designations (Police) Frequency Band 2.455 GHz S 10.525 GHz ± 25 MHz X 13.450 GHz Ku 24.125 GHz ± 100 MHz K 24.150 GHz ± 100 MHz K 33.4–36.0 GHz Ka 332 THz IR (Infrared) Table 2.7 Military Radar Frequency Band 3–30 MHz HF 30–300 MHz VHF 300–1,000 MHz UHF 1–2 GHz L 2–4 GHz S 4–8 GHz C 8–12 GHz X 12–18 GHz Ku 18–27 GHz K 27–40 GHz Ka 40–300 GHz Mm Table 2.8 ITU Radar Bands Frequency Band 138–144 MHz VHF 216–225 MHz 420–450 MHz UHF 890 - 942 MHz 1.215–1.400 GHz L 2.3–2.5 GHz S 2.7–3.7 GHz 5.250–5.925 GHz C 8.500–10.680 GHz X 13.4–14.0 GHz Ku 15.7–17.7 GHz 24.05–24.25 GHz K 33.4–36.0 GHz Ka 59.0–64.0 GHz V 76.0–81.0 GHz W 92.0–100.0 GHZ 126.0–142.0 GHz mm 144.0–149.0 GHz 231.0–235.0 GHz 238.0–248.0 GHz
  • 45. 2.2 Spectrum Division 33 The selected United States radio frequency allocation from 30 MHz to 300 GHz according to the FCC’s “Online Table of Frequency Allocations,” 47 C.F.R., 2. 106; and the selected European radio frequency allocation from 30 MHz to 300 GHz according to the ERC Report 25, “The European Table of Frequency Alloca- tions and Utilizations in the Frequency Range 9 kHz to 1000 GHz”; are given at the end of this book in the Appendix A.Selected Bibliography The European Table of Frequency Allocations and Utilizations Covering the Frequency Range 9 kHz to 275 GHz, ERC Report 25, Copenhagen 2004, cial/pdf/ErcRep025.pdf. “FCC Online Table of Frequency Allocations 47 C.F.R. § 2.106,” Revised on September 23, 2008, Manual of Regulations and Procedures for Federal Radio Frequency Management, National Tele- communications and Information Administration,
  • 46. CHAPTER 3Electromagnetic Properties ofCommunications Systems3.1 Fundamental Communications System Electromagnetics This chapter will focus on the free space relations, leading into basic propagation theory. Electromagnetic waves are mostly characterized by their wavelength, as we have seen in Chapter 2. They are also characterized by their frequency and energy. Every electromagnetic source whose characteristics change (oscillate) with time will produce waves with certain properties. An electromagnetic wave is a propagating electromagnetic field through a medium. The speed of the wave depends on the medium through which it propagates. The wave is polarized depending on the ori- entation of its oscillation. The waves can carry energy from the source into the medium through which they propagate. Radiation is an example of this energy transfer. Electromagnetic waves propagate via reflection, refraction, diffraction, and dispersion. The electromagnetic wave can propagate through different types of mediums: partial conductor, perfect dielectric (insulator), free space, and good conductor. The wave consists of both electric and magnetic fields. The ratio of these two fields (impedance) depends on the losses in the medium. If we are dealing with a partial conductor (i.e., seawater), the wave impedance will be jωµ η= (3.1) σ + jωε −7 where σ is the conductivity in S/m, µ is the permeability (4π · 10 H/m), and ε is the permittivity (8.852 · 10−12 F/m) of the medium. The phase velocity, ω equals 2πf, where f is the frequency of the wave. The angle between the electric and magnetic field, θ, is 0º < θ < 45º. The velocity of the wave is obtained from: ω 1 v= = (3.2) β µε ⎛ ⎞ 2 ⎜ 1 + ⎛ σ ⎞ + 1⎟ ⎜ ⎟ 2 ⎜⎝ ⎝ ωε⎠ ⎟ ⎠ where β = 2π/λ is the phase constant. Wavelength, λ, is calculated from: 35
  • 47. 36 Electromagnetic Properties of Communications Systems 2π 1 λ= = (3.3) β µε ⎛ ⎞ 2 ω ⎜ 1 + ⎜ σ ⎞ + 1⎟ ⎛ ⎟ 2 ⎜⎝ ⎝ ωε⎠ ⎟ ⎠ When dealing with a perfect dielectric, where the conductivity σ = 0, (3.1) is simplified, the wave impedance becomes µ η= ∠0º (3.4) ε In this case, there is no attenuation of the electric or magnetic component of the electromagnetic wave, and they are in phase all the time, that is, θ = 0°. Phase velocity is equal to ω 1 v= = (3.5) β µε and the wavelength 2π 2π λ= = (3.6) β ω µε If the electromagnetic wave propagates through free space, then permeability and permittivity are: µ = µ 0 = 4π ⋅ 10 −7 H m (3.7) ε = ε 0 = 885 ⋅ 10 −12 F m . The wave impedance in this case is: η = 120π ≈ 377Ω (3.8) and the velocity is equal to the speed of light, that is, v = c ≈ 3 ⋅ 10 8 m s (3.9) It is valid for a good conductor: σ >>ωε. Then, the spreading constant γ can be written as: γ = α + jβ (3.10) where α is the attenuation constant and β is the phase constant as given before. Both of them are equal to ωµσ α= β= = πfµσ (3.11) 2
  • 48. 3.1 Fundamental Communications System Electromagnetics 37 The wave impedance can be written as: ωµ η= ∠45º (3.12) σ The wave entering the conductor is attenuated very fast. The intensity is attenu- ated with the factor e−αy. Both the electric and magnetic fields are attenuated to the value of 1/e, or 36.8% of the surface value at the depth for which αy = 1. This spe- cial value is defined as the skin depth δ and is calculated as 1 δ= (3.13) πfµσ In summary, the medium is a good conductor if the loss tangent is large (σ >> ωε). If the loss tangent is very small (σ >> ωε), the medium is a good dielectric. Equa- tions for calculating the attenuation constant, phase constant, and impedance for various medium types are given in Table 3.1. When a wave comes to the border of two mediums, one part will be reflected, and the other part will go through into the second medium. For the electric field in any point, Eg ⎡ η 1 ⎤ + jγ l E R ⎡ η 1 ⎤ − jγ l E = E ′ e + jγ l + E ′ e − jγ l = ⎢1 + η ⎥ e + ⎢1 − η ⎥ e (3.14) 2 ⎣ 2 ⎦ 2 ⎣ 2 ⎦ is valid, where vectors E´ and E´´ represent the components of incident and reflected wave on the border of two mediums, and l is the distance from the border. There- fore, the component of the incident and reflected wave can be written as: ER ⎡ η1 ⎤ ER ⎡ η1 ⎤ ER = ′ ⎢1 + η ⎥, ER = ′′ ⎢1 − η ⎥ (3.15) 2 ⎣ 2 ⎦ 2 ⎣ 2 ⎦ where η1 and η2 are the impedances of the medium 1 and 2. The reflection coefficient is obtained from Table 3.1 Attenuation Constant, Phase Constant, and Impedance for Various Medium Types Medium with Losses Good Good Free Conductor Dielectric Space Attenuation ωµσ ≈0 0 µε ⎡ ⎤ 2 ⎛ σ ⎞ Constant α ω ⎢ 1 + ⎜ ⎟ − 1⎥ 2 2 ⎢ ⎝ ωe ⎠ ⎥ ⎣ ⎦ Phase ωµσ ω µε ω µ0 ε0 ωε ⎡ ⎤ 2 ⎛ σ ⎞ Constant β ω ⎢ 1 + ⎜ ⎟ + 1⎥ 2 2 ⎢ ⎝ ωε ⎠ ⎥ ⎣ ⎦ Impedance η jωµ ωµσ µ 377 (1 + j ) σ + jωε 2 ε
  • 49. 38 Electromagnetic Properties of Communications Systems → ′′ ER η 2 − η1 rR = = (3.16) → ′ ER η 2 + η1 and the transmission coefficient from → E ′′′ R 2η 2 rT = = (3.17) → ′ ER η1 + η 2 The vector of the transmitted electric field is equal to the sum of the vector of incident and the reflected wave: → → → ER = ER + ER ′′′ ′ ′′ (3.18) The power density of the electromagnetic wave is equal to → 2 → 2 → E H P = cos ϕ = η cos ϕ (3.19) 2η 2 where ϕ is the angle between the electric and magnetic field. The free space imped- ance is equal to Ex η= (3.20) Hz The wave propagates in the z direction, and electric (x) and magnetic (y) com- ponents are perpendicular to each other and to the direction of the wave propaga- tion. Power densities are correlated as follows: → → → PR = PR PR ′ ′′+ ′′′ (3.21) which means that the incident power is divided into reflected and transmitted power. If there is a need to calculate the input impedance on some distance l from the border of two mediums (Figure 3.1), the following expression will be used η 2 + jη1 tan β1 l Z in = Z B = η1 (3.22) η1 + jη 2 tan β1 l The input impedance can be solved with the Smith chart as the propagation through the free space can be substituted with the transmission line, or more pre- cisely, on the circle of constant attenuation.
  • 50. 3.1 Fundamental Communications System Electromagnetics 39 ε2 ε1 A B η2 η1 ZB Figure 3.1 Electromagnetic wave at the border of two mediums. 3.1.1 Smith Chart Let a load ZR be connected to some voltage source Ug, with inner impedance Zg (Fig- ure 3.2). The line has impedance Z0. From the theory of networks, it is well known that most of the energy from the source will be given to the load if inner source impedance is equal to the load impedance, and both of them are equal to the imped- ance of the line (Zg = Z0 = ZR). If a load is different on the transmission line (wave- guide or cable in real situations) from the line characteristic impedance (usually 50Ω or 75Ω) connected, then not all of the energy from the source will be transmit- ted to the load. There will be a reflection, which might even damage the source (generator). The aim is to adapt the load to the generator with compensation elements that are either inductive or capacitive in character. In some special cases, only an open or shorted transmission line of a certain length will be sufficient. This will perform the adapta- tion and then there will be no reflection on the load (usually an antenna) and no return of the power back to the transmitter. The input impedance in the line can be found from the following expression: Z0 ZR Z0 Ug Zin Figure 3.2 Voltage source Ug, with inner impedance Zg.
  • 51. 40 Electromagnetic Properties of Communications Systems Z R + jZ 0 tan βl Z in = Z 0 (3.23) Z 0 + jZ R tan βl where βl is the electric length of the line. The reflection coefficient, rR, determines how much of the power is reflected on the load: ZR − Z0 rR = (3.24) ZR + Z0 The incident and the reflected wave result in the standing wave. The standing wave ratio (SWR or ρ) defines how much a load is adapted to the source: Emax 1 + rR SWR = = (3.25) Emin 1 − rR SWR can also be defined as the ratio of the maximum electric field and mini- mum electric field. When SWR is equal to 1, it means that the reflection coefficient is equal to 0—which means that we have a perfect match. SWR cannot be smaller than 1. This is the center of the Smith chart (Figure 3.3). In the Smith chart, the upper part represents the inductive character (+j), and the bottom part the capacitive character (−j). The impedance of the short circuit is equal to 0 and can be found at the left side of the chart next to the wavelength of 0λ. The admittance of the short circuit is equal to ∞ and is found at the far right side next to the wavelength of 0.25λ. The open line, in turn, has an impedance of ∞, next to the wavelength of 0.25λ, and the admittance is 0 with the wavelength of 0λ. +j1 0.125λ Curve of constant real part 0 1.0 3.0 00 0λ 0.25λ SWR=3 −j1 0.375λ Figure 3.3 The Smith chart.
  • 52. 3.1 Fundamental Communications System Electromagnetics 41 If the load is reactance, it will be placed on the circle of reactive reactance (or capacitive susceptance) on the top or bottom of the circle, depending on the value. The goal of matching (or adapting) the load is to get it as close as possible to the cen- ter of the chart. Usually, the values that are inserted into the chart are divided with characteristic impedance of the line, that is, Y=1/Z0. Moving along the transmission line corresponds to moving along the circle with a center in the center of the Smith chart. This circle is called the circle of constant attenuation and is not drawn in the Smith chart. An individual solving the problem must draw it on his or her own. The circles of the constant real part (0.1, 0.3, . . . 1.0, 3.0, . . . 10.0, . . .) are drawn in the chart; they represent the point in the transmission line and differ only in reactance. The parts of the circles in the upper and lower part, which have a common reactance part (± j) and different real part, are also shown. The matching of the load to the generator impedance can be performed by add- ing one or more compensation elements, and in some cases with λ/4 transformers. These elements can be open line, short circuited, or have some reactance (C or L). Their purpose is to create a standing wave on the compensation element, so that the reflected wave from the compensation element and the load nullify each other (Figure 3.4). Here, the circulation of the energy is at hand. Theoretically, there are no losses on the transmission line, since we assume that the line itself has no losses. If the load has only resistors (no reactance) and is used at a fixed frequency, λ/4, transformers (Figure 3.5) can be used for impedance matching. The input power (voltage and current are in phase) is equal to the one dissipated on the load: 2 U1 U2 P = U1 I1 = U 2 I 2 = I1 Z ul = I 2 Z R = 2 2 = 2 (3.26) Z ul ZR The characteristic impedance of the line is calculated from Z0 = Z in ⋅ Z R (3.27) Z0 Z0 ZR Z0 Energy circulation Figure 3.4 Circulation of the energy on the compensation element.
  • 53. 42 Electromagnetic Properties of Communications Systems I2 I1 U1 U2 ZR λ/4 Zin Figure 3.5 A λ/4 transformer. 3.1.2 Snell’s Law of Reflection and Refraction If the wave is entering from one medium into the other at an angle rather than per- pendicular, there will be a transmitted component as well as a reflected component of the incident wave, and their intensities will depend on the angle of incidence. Snell’s law of reflection says that the angle of incidence θi will be equal to the angle of reflection θr, that is, θi = θr (3.28) while Snell’s law of refraction says that the angle of refraction (transmitted wave), θt, and the angle of incidence, θi, will be sin θ i µ2 ε2 = (3.29) sin θ t µ 1 ε1 Total reflection appears when θt = 90º , which can happen only when a wave travels from an electrically denser medium into a less dense medium. For instance, when the wave from the Teflon ( r = 2,1, µr = 1) enters free space, the critical angle when the total reflection will appear is ε2 1 θ c = sin −1 = sin −1 = 4364º . (3.30) ε1 2,1 The electric component of the wave can be vertically or horizontally polarized in regard to the incident plane. If it is vertically polarized (Figure 3.6), it is parallel to the incident plane and the wave will be totally or partially reflected. With horizontal polarization (Figure 3.7), the electric component lies in the incident plane, and with µ1 = µ2, there can be an incident angle where there will be no reflected wave (total transmission). Total transmission exists only when a wave travels from an electrically less dense to an electrically denser medium. This angle is called the Brewster’s angle and is defined as:
  • 54. 3.1 Fundamental Communications System Electromagnetics 43 x H” 1 2 E” E’” θr θt H’” θl y E’ H’ Figure 3.6 Vertical wave polarization. x 1 2 H” E’” E” H’” θr θt θl y E’ H’ Figure 3.7 Horizontal wave polarization. ε2 θ B = tan −1 (3.31) ε1 For instance, when a wave travels from air to glass (εr = 5, µr = 1), the incident wave at which there will be no reflection (total transmission) will be ε2 5 θ B = tan −1 = tan −1 = 65.91º (3.32) ε1 1 If the angle of incidence is at an angle other than perpendicular, there are two components of the wave: one which is transmitted, and the other vibrating at the border, so that the following expression cannot be used for the reflection coefficient: η 2 − η1 rR = (3.33) η 2 + η1
  • 55. 44 Electromagnetic Properties of Communications Systems Instead, the following can be used: Zη2 − Zη1 rR = (3.34) Zη2 + Zη1 The values Z η are different for vertical and parallel polarization. For vertical polarization, it will be: η1 Zη1 = cos θ 1 (3.35) η2 Zη2 = cos θ 2 and thus, the reflection coefficient, Z n2 − Z η 1 sin( θ 2 − θ 1 ) rR = = (3.36) Z n2 + Z η 1 sin( θ 2 + θ 1 ) For parallel polarization it will be: Z η 1 = η1 cos θ 1 (3.37) Z η 2 = η 2 cos θ 2 and the reflection coefficient is Zη2 − Zη1 tan( θ 2 − θ 1 ) rR = =− (3.38) Zη2 + Zη1 tan( θ 2 − θ 1 ) Other types of electromagnetic wave propagation are diffraction and disper- sion. Diffraction is the bending of waves around small obstacles and the spreading out of waves past small openings. With dispersion, an electromagnetic wave is sepa- rated into components with different wavelengths due to refraction, interference, or diffraction. Diffraction will be covered in more detail in Section Wave Generation and Propagation in Free Space The propagation of electromagnetic waves deals with the way the wave travels from the transmitting antenna to the receiving antenna. The electromagnetic waves can travel through guided structures like transmission lines, waveguides, and free space. This will be the subject of interest in this section. 3.2.1 Maxwell’s Equations Maxwell’s equations can be written in integral or differential form. Written in inte- gral form they look as follows:
  • 56. 3.2 Wave Generation and Propagation in Free Space 45 q ∫ E ⋅ dA = ε 0 (3.39) ∫ B ⋅ dA = 0 (3.40) dΦ B ∫ E ⋅ dl = − dt (3.41) 1 ∂ ∫ B ⋅ dl = µ 0 i+ c 2 ∂t ∫ E ⋅ dA (3.42) Written in differential form the equations look as follows: ∇ ⋅D = ρ (3.43) ∇⋅B = 0 (3.44) ∂B ∇×E= − (3.45) ∂t ∂D ∇×H = J+ (3.46) ∂t where E is the electric field strength in V/m; H is the magnetic field strength in A/m; D is the electric flux density in C/m2 (Coulombs); B is the magnetic flux density in Wb/m2 (Webers); J is the conduction current in A/m2; and v is the electric charge density in C/m3. Additional equations describing the relation for the medium are: D = εE (3.47) B = µH (3.48) J = σE (3.49) where ε = ε0εr is the permittivity, µ = µ0µr is the permeability, and σ is the conductiv- ity of the medium. The first Maxwell equation (3.39) is also called Gauss’ law for electricity. It says that the electric flux out of any closed surface is proportional to the total charge enclosed within the surface. In the second (3.40), Gauss’ law for magnetism says that the net magnetic flux out of any closed surface is zero. Equation (3.41), or Fara- day’s law of induction, says that the line integral of the electric field around a closed loop is equal to the negative of the rate of magnetic flux change through the area enclosed by the loop. Equation (3.42), or Ampere’s law, says that in the case of a
  • 57. 46 Electromagnetic Properties of Communications Systems static field, the line integral of the magnetic field around a closed loop is propor- tional to the electric current through the loop. 3.2.2 Wave Propagation If it is assumed that the wave propagates in the z direction and the wave is polarized in the x direction, the values of the electric and magnetic field will depend on the dis- tance and time according to: E( z, t ) = E 0 e − αz cos( ωt − βz )a x (3.50) E 0 − αz H( z, t ) = e cos(ωt − βz − θ η )a y (3.51) η where η is the intrinsic impedance of the medium calculated from µ ε σ η = 1 , tan 2 θ = , 0 ≤ θ ≤ 45 º (3.52) 4 ωε ⎛ σ⎞ 4 1+ ⎜ ⎟ ⎝ ωε⎠ Equations (3.50) and (3.51) show the attenuation of the electromagnetic wave while it propagates through a medium with the factor e az (Figure 3.8). Power density of the electromagnetic wave is 2 E 0 −2 αx P= e cos θ η a z (3.53) 2η In free space the E and H fields are perpendicular to each other and to the direc- tion of wave propagation. The loss of the electromagnetic wave in the z direction happens due to the rela- tive permittivity of the medium. x e−αz E z H y Figure 3.8 Components of the electric and magnetic fields in a lossy medium.
  • 58. 3.2 Wave Generation and Propagation in Free Space 47 3.2.3 Wave Polarization The electric field can have various orientations depending on the transmitting antenna. The polarization is the orientation of the tip of the electric field in a plane perpendicular to the direction of the propagation at some point in space as a func- tion of time. The types of polarization are linear (vertical or horizontal), circular, and elliptic. Under extreme conditions in the atmosphere (e.g., rain), electromag- netic wave depolarization is possible. Depolarization can also happen from reflections. With linear polarization, the orientation of the field is constant in space and time. For a wave traveling in the z direction, the electric field can be written as E = Ex a x + Ey a y (3.54) where E x = a cos( ωt − kz + φ a ) (3.55) E y = b cos( ωt − kz + φ b ) (3.56) The trajectory of the electric field vector E will be drawn on the x, y plane. The tip of the electric field vector moves as time goes by. The polarization will be linear when phase angles a and b are equal and the trajectory is a line. The phase difference between the angles must be ∆φ = φ b − φ a = nπ, n = 0, 1, 2, K (3.57) Linearly polarized waves can be generated using simple antennas such as dipoles. Figure 3.9 shows the linear polarization. Circular polarization will have different phase angles φa and φb, but the ampli- tudes a and b will be the same. This results in a circle trajectory. The phase differ- ence between the angles is ⎛1 ⎞ ∆φ = φ b − φ a = ± ⎜ + 2n⎟ π, n = 0, 1, 2, K (3.58) ⎝2 ⎠ y E x Figure 3.9 Linear polarization.
  • 59. 48 Electromagnetic Properties of Communications Systems y E x Figure 3.10 Circular polarization. Circularly polarized waves can be generated by a helically wound wire antenna or with two linear sources perpendicular to each other. Figure 3.10 shows the circu- lar polarization. Both the linear and circular polarizations are special cases of elliptical polariza- tion. Elliptical polarization will occur when the phase angles φa and φb, as well as the amplitudes a and b, are different. The trajectory in this case is elliptical. The phase difference between the angles is the same as in the circular polarization. Fig- ure 3.11 shows the elliptical polarization. 3.2.4 Fresnel Knife-Edge Diffraction Diffraction occurs when an electromagnetic wave encounters an obstacle. The wave will bend around the obstacle and continue to spread. If there is no obstacle, the electromagnetic wave will travel in a straight line from the transmitter to the receiver. However, if there are obstacles near the path, they will influence the wave by possible power reduction or phase distortion. Fresnel’s zones are ellipsoids (Fig- ure 3.12) where obstacles can create signals that will be out of phase. The first Fresnel zone creates signals that are 0° to 90° out of phase; the second zone creates signals that are 90° to 270° out of phase; the third zone creates signals that are 270° to 450° out of phase, and so forth. Odd number zones are construc- y E x Figure 3.11 Elliptic polarization.
  • 60. 3.2 Wave Generation and Propagation in Free Space 49 Fourth Fresnel zone Third Fresnel zone Second Fresnel zone r1 First Fresnel zone T R d1 d2 Figure 3.12 Fresnel zones. tive, since they reinforce the signal, and even numbered zones are destructive, since they destroy the signal. The obstacles in the first zone are potentially the most dangerous ones. The radius r1 of the first Fresnel zone is calculated from λd 1 d 2 r1 = (3.59) d1 + d 2 where d1 and d2 are the distance between the obstacle and the transmitter and receiver. The above expression is valid when the distances d1 and d2 are much larger than r1. To achieve communication, it is desirable to have the first Fresnel zone clear of any obstacles—to be more precise 60% of the first Fresnel zone should be clear of obstacles, meaning a radius of 0.6 r1. α<0 d1 d2 h<0 T R T R h<0 d1 d2 α<0 Figure 3.13 Knife-edge diffraction.
  • 61. 50 Electromagnetic Properties of Communications Systems Figure 3.13 shows the knife-edge diffraction from an obstacle in the line of sight. This obstacle can be above or below the line of sight. The Fresnel-Kirchoff diffraction parameter v is the dimensionless quantity, which can be calculated from 2( d 1 + d 2 ) v=h (3.60) λd 1 d 2 which depends on the distances from the transmitter and receiver to the tip of the obstacle, height of the obstacle, and wavelength. The value of h can be positive or negative. The value of v can also be calculated from the angle of diffraction α, which can be either positive or negative, like the height h. 2d 1 d 2 v=α (3.61) λ( d 1 + d 2 ) The diffraction loss for knife-edge obstacle can be calculated from ⎧ 0 v ≤ −1 ⎫ ⎪ 20 log(05 − 062 v) . . −1 ≤ v ≤ 0⎪ ⎪ ⎪ ⎪ ⎪ 20 log 05 ⋅ e . ( −0. 95 v ) 0≤ v≤ 1⎪ ⎪ A( v) = ⎨ ⎬ dB (3.62) 20 log⎛0.4 − 01184 − (038 − 01v) ⎞ 1 ≤ v ≤ 2.4⎪ 2 ⎪ ⎜ . . . ⎟ ⎝ ⎠ ⎪ ⎪ ⎪ ⎛ 0225 ⎞ . 20 log⎜ ⎟ v ≤ 2.4 ⎪ ⎪ ⎩ ⎝ v ⎠ ⎪ ⎭ The results of losses (dB) are shown in Figure 3.14. 0 −5 −10 A(v), dB −15 −20 −25 −2,0 −1,0 0,0 1,0 2,0 3,0 v Figure 3.14 Knife-edge diffraction loss.
  • 62. 3.2 Wave Generation and Propagation in Free Space 51 Most obstacles in real situations are large in comparison to the signal wave- length and are not knife-edge. In such situations different models for path loss pre- dictions are used. With higher frequencies, the first Fresnel zone gets smaller and smaller. However, when the Fresnel zones are smaller, the diffraction loss will become greater if the receiver antenna is lowered. 3.2.5 Path Loss Prediction Path loss or attenuation is the reduction of the power density of an electromagnetic wave as it travels through space. When calculating the link budget, the path loss is of great importance. It is calculated for free space, but different factors such as reflection, absorption, and refraction influence its value. Terrain is also of impor- tance, so different models are used for urban, semiurban, or rural terrain. Path loss is calculated in free space from ⎛ 4πd ⎞ L f = 20 log10 ⎜ ⎟ (3.63) ⎝ λ ⎠ where d is the distance between the transmitter and the receiver and λ is the wave- length. It is usually given in dB/km or dB/m. In closed areas (buildings) the addi- tional loss is 1 dB/m (i.e., an office). The exact path loss will depend on the actual situation, width of the walls, and so forth. The path loss for 2.4 GHz is shown in Figure 3.15 for free space and inside the building. There are several models for path loss in an urban area. The most popular model is the Hata model for urban areas. In cities, there is almost never a line of sight (LOS) between the transmitter and a receiver. The Hata model parameters are: • d: the distance from the transmitter to the receiver (1–20 km); 160 140 120 100 Free space L f (dB) In building 80 60 40 20 0 20 40 60 80 d (m) Figure 3.15 Path loss for 2.4 GHz.
  • 63. 52 Electromagnetic Properties of Communications Systems • f: the frequency in MHz (100–1,500 MHz); • hb: the base station height (30–200m); • hm: the mobile station height (1–10m). The mean path loss is given by empirical equation: L S = 6955 + 2616 log f − 1382 log( hb ) + . . . (3.64) [ 449 − 655 log( h )] log( d ) − a( h ) − L . . b m αx in an open, suburban, or medium-size city . [ . .] a( hm ) = 11 log( f ) − 07 hm − 156 log( f ) + 08 . (3.65) and in a big city ⎧ 829 log 2 (154hm ) − 11 f ≤ 300 MHz . . . a( hm ) = ⎨ (3.66) ⎩32 log 2 (1175 hm ) − 497 f ≥ 300 MHz . . . The correction factor is ⎧ 2 log 2 ( f 28) + 5.4 in the suburbs L cor = ⎨ (3.67) ⎩ 478 log ( f ) − 1833 log( f ) + 4094 in the open 2 . . . Figure 3.16 depicts an example of path losses for open space, a suburban medium sized city, and a large sized city, versus distance (in kilometers). The heights of the transmitter and receiver in this example are 50m and 2m, respectively. The frequency is 900 MHz. The medium and large city path losses are almost the same for this example. 180 160 140 Open area L S (dB) Suburbs Medium city 120 Big city 100 80 0 2 4 6 8 10 12 14 16 18 20 d (km) Figure 3.16 Hata model path loss.
  • 64. 3.3 Wave Generation and Propagation in the Terrestrial Atmosphere 53 The reflections from the obstacles or objects produce multiple paths or fading. Usually 30 dB is considered enough to raise the transmitted power (or some other means) for compensation. Spatial or frequency diversity is used to battle this prob- lem. Many other models are used to calculate the path loss, including the Irregular Terrain Model (also known as the Longley-Rice code), which is a model of radio propagation for frequencies between 20 MHz and 20 GHz. The Longley-Rice model predicts the median attenuation of a radio signal as a function of distance and the variability of the signal in time and in space. A more precise evaluation can only be obtained with test measurements in the field. This is more expensive but gives better insight into the problem.3.3 Wave Generation and Propagation in the Terrestrial Atmosphere In the atmosphere there are gases: mainly nitrogen, oxygen, and carbon dioxide. The atmosphere is also influenced by gravity. Near the Earth’s surface, density and pressure are higher than at higher altitudes away from the Earth. The influence on the propagation is the highest closest to the Earth. The atmosphere is divided into layers of which the most important are: the troposphere, stratosphere, and iono- sphere. The troposphere is about 11 km high, depending on geographical latitude. It is the warmest, wettest, and densest layer of the atmosphere, with the greatest influ- ence on communication systems, mostly due to rain. The next layer is the strato- sphere, which reaches up to 50 km high. It is much colder than the troposphere and does not influence microwave transmissions very much. The ionosphere is the next layer, and has three parts: the D, E, and F regions. The D region is 75 to 95 km away from the Earth’s surface and has weak ionization. The E region is 95 to 150 km away from the Earth’s surface. The F region is 150 to 6,000 km from the Earth’s surface and has the most electrons of all three regions. It is the most important region for communications. The density and refractive index, which change with altitude and weather con- ditions, and the curvature of the Earth can influence the communication links for satellite and microwave applications over large distances. This means that commu- nication is possible, even if there is no line of sight (LOS) between the transmitter and the receiver. 3.3.1 Absorption and Scattering Absorption and scattering are the main sources of losses in the troposphere. Absorption occurs because atmospheric gas molecules resonate at some frequencies (i.e., water vapor molecules resonate at 22.235 GHz and oxygen molecules at 60 GHz). The absorption is always present, although it can depend on the humidity. More about absorption in the atmosphere will be discussed in Chapter 4. Scattering occurs when an electromagnetic wave collides with atmosphere par- ticles. If these particles are smaller than the wavelength, Rayleigh scattering will happen. The particles reflect some of the energy, depending on the size and dielec- tric property. They can be dust, nitrogen, or oxygen molecules. In addition, Mie scattering occurs when the particles in the atmosphere are about the same size as the
  • 65. 54 Electromagnetic Properties of Communications Systems Ionosphere Reflection from ionosphere Troposphere Direct wave Transmitter Reflected wave Receiver Surface wave Earth Figure 3.17 Wave propagation in the atmosphere. wavelength. These particles are dust, pollen, water vapor, and smoke and are usu- ally present in the lower parts of the atmosphere, especially if there are clouds. The last type is nonselective scattering, which occurs when the particles, usually large dust or rain drops, are much larger than the wavelength. 3.3.2 Wave Propagation in the Atmosphere There are three possible types of wave propagation over the Earth (Figure 3.17): • Surface propagation along the surface of the earth; • Wave propagation through the troposphere; • Propagation by reflection from the ionosphere. The ionosphere refracts the wave back to the earth in a frequency range up to approximately 50 MHz. The surface wave can be used up to 5 MHz. The surface wave is attenuated more than the wave traveling through free space, so the transmit- ters in these bends must have a higher transmitting power. In free space, beside the direct wave there is usually at least one reflected wave.
  • 66. 3.3 Wave Generation and Propagation in the Terrestrial Atmosphere 55Selected Bibliography Barclay, L. W., Propagation of Radio Waves, 2nd ed., London, U.K.: Institution of Electrical Engineers, 2003. Deal, W. R. et al., “Guided Wave Propagation and Transmission Lines,” in RF and Microwave Handbook, M. Golio, (ed.), Boca Raton, FL: CRC Press, 2001. Lee, W. C. Y., Mobile Communications Engineering, New York: McGraw-Hill, 1982. Magnusson, P. C., et al., Transmission Lines and Wave Propagation, 4th ed., Boca Raton, FL: CRC Press, 2001. Rappaport, T. S., Wireless Communications: Principles and Practice, Upper Saddle River, NJ: Prentice-Hall, 2001. Rothwell, E. K., and M. J. Cloud, Electromagnetics, Boca Raton, FL: CRC Press, 2001. Sadiku, M.N.O., and K. Demarest, “Wave Propagation,” in Electrical Engineering Handbook, Dorf, R. C., (ed.), Boca Raton, FL: CRC Press, 2000. Solheim, F. S., et al., “Propagation Delays Induced in GPS Signals by Dry Air, Water Vapor, Hydrometeors, and Other Particulates,” Journal of Geophysical Research, Vol. D8, April 1999, pp. 9663–9670. Smrkic, Z., Mikrovalna Elektronika, Skolska Knjiga, Zagreb, 1986.
  • 67. CHAPTER 4Electromagnetic Interference4.1 Electromagnetic Interference with Wave Propagation andReception Electromagnetic interference exists in every communication link. It manifests itself as noise, which degrades the quality of the application. In analog systems, tradition- ally the signal-to-noise (S/N) ratio is used to show the quality of the communication link. In every case, the signal level should be above the noise for communication to be possible. How much above depends on the quality of the receiver used. In digital systems, especially where the spread spectrum is used, the ratio S/N is not the best parameter to evaluate link quality, since the signal is almost always buried in the noise—but this does not mean that communication will be impossible. In this case, other parameters such as the energy of the bit compared to the noise spectral density (Eb/N0), are much better to use regarding the quality of the communication. Any signal, although intentional and useful, is considered noise to other signals in the same channel or frequency band. This is why careful planning and good fre- quency allocation is necessary. In some cases even neighboring countries must work together, because electromagnetic signals are not bound to national borders. There are several types of interference or noise, which are either natural or man- made. Natural interference includes phenomena such as lightning or electrostatic discharge, atmosphere effects, sunspot activity, and reflections from the rough Earth surface. Manmade interference comes from both commercial and military communications such as radar, radio, television, and cell phone communications. Industry can also create interference. All this interference is unintentional, but there can also be intentional interference, especially during a war. 4.1.1 Additive White Gaussian Noise (AWGN) Additive white Gaussian noise (AWGN) is a statistically random noise in the wide frequency range (very low frequencies up to 1012 Hz) with constant spectral density. AWGN can come from many sources such as thermal noise, shot noise, noise from Sun radiation, and others. It is a background noise in the communication channel. If in the communication channel (Figure 4.1) a signal s(t) is introduced, it will be added by additive white Gaussian noise n(t): r(t ) = s(t ) + n(t ) (4.1) 57
  • 68. 58 Electromagnetic Interference r(t) s(t) Receiver n(t) Figure 4.1 AWGN channel communication model. At the receiver the signal r(t) will be received. There, in the process of detection (see Section 6.1), the decision about the value of the signal will be made. It is possible to use this model only in deep space communications (i.e., between satellites), where the only degradation in the channel is caused by the thermal noise in electronic devices. In real situations multipath, fading, dispersion, and other fac- tors must be included. 4.1.2 Thermal Noise Conductor resistivity used for the flow of electrons depends on temperature. Thus, temperature will have an influence on the noise in the communication channel. The thermal noise Pterm, in [W] (sometimes defined as Nt), is defined as Pterm = kTB (4.2) −38 where k is the Boltzmann 1.38 · 10 , T is the temperature in [K], and B is the fre- quency bandwidth in [Hz]. Thermal noise exists in every communication system and cannot be avoided. 4.1.3 Shot Noise Shot noise appears in electrical circuits where direct current (DC) flows. It repre- sents small variations of the current. This noise does not depend on temperature. The noise current In, in [A], is defined as I n = 2 qI DC B (4.3) −19 where q is charge of the electron, 1.6 · 10 C, IDC is a DC bias current in the electric circuit, and B is the frequency bandwidth in [Hz]. 4.1.4 Flicker (1/f ) Noise Flicker noise is proportional to the bias current and decreases with frequency. Its power density is proportional to 1/f, and falls by approximately 10 dB per decade. Flicker noise is weak above several kilohertz and is sometimes called pink noise.
  • 69. 4.2 Natural Sources of Electromagnetic Interference 59 4.1.5 Burst Noise Burst noise appears in semiconductors and is also called popcorn noise. It is caused by defects in the manufacturing process like heavy metal ion contamination or sur- face contamination. The noise increases with the bias current level and is propor- tional to 1/f2. 4.1.6 Noise Spectral Density Noise spectral density, N0, is the noise in the frequency range of 1 Hz: Pterm N0 = = kT (4.4) B In digital systems, energy per bit, Eb, is often used with noise spectral density for evaluating data (bit) error rate performance (BER). It can be found using the signal to noise ratio by: Eb S B = ⋅ (4.5) N0 N R where R is the data rate and B is the frequency bandwidth. 4.1.7 Effective Input Noise Temperature The effective input noise temperature, Te, is defined as the temperature at which the input impedance has to be placed in order to generate the observed noise power at the output of a two-port network or amplifier. It is calculated as Te = 290( NF − 1) (4.6) where NF is the noise factor [defined in (1.19)] at 290K. This parameter is often used for satellite communications where antennas are pointed to the cold sky, and the temperature of 290K (used for most terrestrial com- munications) is not applicable.4.2 Natural Sources of Electromagnetic Interference 4.2.1 Lightning and Electrostatic Discharge Lightning and electrostatic discharge are examples of transients. Transients can be created from guided or radiated emissions from electromechanical or electronic devices, or from natural interference or discharges. They often appear as a result of current changes in inductive loads such as engines or relays. They can be created by radar as well as isolators in high voltage conductors during bad weather and can be dangerous, as the semiconductor could burn, the capacitor could explode, and the wire or transformer isolation could break down. Transients rise quickly and fall slowly (ratio of one to hundred). The rise time ranges from a nanosecond to a milli-
  • 70. 60 Electromagnetic Interference second. The amplitudes can be from below one volt up to more than one hundred kilovolts. Lightning Lightning is a transient electric discharge, the path of which is measured in kilome- ters. It appears when a part of the atmosphere becomes electrically charged enough to allow electric breakdown in the air. It is the strongest natural force. In most cases, lightning appears in clouds, but it is also possible in snowstorms, desert storms, and above erupting volcanoes. It can very rarely appear on mountains or tall TV towers. It can strike the same place several times during the same storm. The lightning waveform is shown in Figure 4.2. An understanding of the wave- form is necessary to create the protection system. The pulse can be divided into three parts (I to III). The first component (initial stroke) is a pulse of strong DC current, which can reach more than 200 kA and last about 200 µs. The rise speed is about 3 · 1010 m/s. The second component is an intermediate phase with a current level of sev- eral kA. It lasts about 5 ms. The third component has a current of around 400A and lasts about 0.75 second. After that, the first component can appear again (restrike) with an intensity of half as much as the initial stroke and of the same length. Usually there are a few of restrikes, each of lower intensity. A lightning strike can cause potential difference between buildings. This poten- tial difference can be up to 1 MV. Figure 4.3 shows the potential difference between two buildings as the result of a lightning strike. When lightning strikes a high voltage post 150m from the building, is there going to be any damage to the cable connecting the two buildings? Let us assume that the other building is 75m away from the first building. If the resistivity of the ground, ρ, is 1 kΩ/m and the current is 200 kA, what will be the potential difference between the two buildings? The potential is calculated from the following equation: I, kA 200 I ~5 II 0.4 III t, ns 3 6 0 200 5×10 0.75×10 Figure 4.2 Lightning waveform.
  • 71. 4.2 Natural Sources of Electromagnetic Interference 61 Building 1 Building 2 Grounded metal conductor d1 d2 Figure 4.3 Potential difference between buildings from lightning. ρI ⎛ 1 1 ⎞ V = ⎜ − ⎟ (4.7) 2π ⎝ d 1 d 1 + d 2 ⎠ where d1 is the distance from the first building to the post and d2 is the distance between the two buildings. The potential for the above values will be 70.77 kV. This can be enough to dam- age the isolation of a communication cable between the buildings, which can be pre- vented by connecting the buildings with a conductor having a small impedance in the frequency range of 300 kHz (lightning strike). Inside this conductor all the com- munication cables are placed. Thus, the lightning currents will flow on the outer surface, which protects the interior and the communication cables. The previously mentioned skin depth (3.13) prevents the currents from going too deep into the conductor. Lightning protection grounding must be performed with care. Typically it con- sists of guides with small impedance. There can be several going in parallel from the top to the bottom of the building. Usually they are made of aluminum and copper, not only because of their electrical characteristics, but also because they are rust resistant. Transient voltages can enter a building through electrical, cable TV, phone, or internet lines. If the antenna on the roof is protected, the rest of the build- ing is not. The cables through which lightning current flows produces a magnetic field, and if it is large enough to encompass other conductors, magnetic coupling can occur. The cables can also conduct the lightning currents to other electronic systems and damage them. If a tree is close to a building and lightning strikes the tree, it can conduct cur- rents to the building. Trees have a relatively large impedance compared to the grounding protection. The diagram in Figure 4.4 shows the situation when light- ning hits a tree. The lightning is a current source, and the tree has impedance Z, so potential will occur on the tree. If it exceeds one million volts, the current can go to the objects in the vicinity. If the typical current is 20 kA and the impedance of the tree 100Ω, the voltage on the tree will be 2 · 106V, which can propel a person standing even 2m from the tree.
  • 72. 62 Electromagnetic Interference I Z d Figure 4.4 Lightning hitting a tree. Electrostatic Discharge (ESD) Electrostatic discharge is a fast spontaneous transmission of the electrostatic charge induced from the electrostatic field. The charge is transferred over the spark (static discharge) between two bodies with different electrostatic potentials when they are close to each other. Electrostatic discharge exists everywhere in our surroundings. How many times have we felt it when we touch a door handle or a metal chair? Even though this dis- charge cannot harm humans, it can be devastating to electronic equipment sensitive to ESD. Even the ESD that we do not feel at all is dangerous to equipment. Table 4.1 shows the typical sources of static electricity, and Table 4.2 gives the typical situa- tions that generate the electrostatic voltages. Table 4.1 Typical Sources of Static Electricity Object Material Floor PVC Concrete Clothes Shoes Worksuit Chair Wood Plastic Packaging Cathode rays room Table 4.2 Typical Situations that Generate Electrostatic Voltages Relative Humidity Relative Humidity Static Discharge Type 10%–20% 65%–90% Walking on the carpet 35,000V 1,500V Walking on the pvc floor 12,000V 250V Plastic foil 7,000V 600V Worker at the desk 6,000V 100V
  • 73. 4.2 Natural Sources of Electromagnetic Interference 63 Moisture is an important factor. It is best to have the relative humidity between 40% and 60% in the working area. The damage from ESD occurs when a person or an object comes into contact with an electronic device sensitive to electrostatic discharge. If this discharge has enough energy, overheating damage can occur. Generally, the more sensitive the equipment is, the more vulnerable it will be to ESD. The damage can be immediate, where the electronic device is damaged or destroyed right after ESD, or latent, where the electronic device appears to be working normally, but the circuitry is damaged and could stop operating at any moment. Protection can be done on several levels. The first is in the working area. Elec- tronic devices sensitive to ESD should be operated in places where there is no ESD. Antistatic wrist tape (Figure 4.5) should be worn if available. Additionally, an air ionizer can be used. Ions are created in nature by ocean waves, waterfalls, and so forth. They purify the air from dust, smoke, or pollen. If we had an EMC laboratory in the open near a waterfall, there would be no trouble with ESD. However, in big cities pollution is greater, so additional air ionizers could be an option, as they are commercially available. Normally, sources of static elec- tricity should be at least 1m away from sensitive equipment. Figure 4.6 shows a work area protected from ESD. The table is covered with material absorbing static charge through a 1 MΩ resistor, which protects the operator from shock if the Earth becomes electrically alive. The mat under the table is also of a similar material to the cover on the table. Ground points (Gp) can also have connectors for the antistatic wrist tape and for other electronic equipment that are being tested. The materials used for carpet and table surfaces should not be made of a metal, like stainless steel, because of their low resistivity, which could lead to transient dis- charges of electricity. Fast discharge is much more dangerous for electronic devices than discharge through static dissipative materials, which should have resistivity values in the range of 105 to 1011Ωm. Second, before operating sensitive equipment, a person should discharge him- self or herself from all static electricity. This can be done with the previously men- tioned antistatic wrist tape or by touching a conductive surface. There are also antistatic suits that can be worn. Last, devices sensitive to ESD should be placed in antistatic bags or containers during transportation and storage. Figure 4.5 Antistatic wrist tape.
  • 74. 64 Electromagnetic Interference Gp Figure 4.6 Working area protected from ESD. Figure 4.7 shows the symbol for ESD danger, which is placed on packages con- taining sensitive electronic equipment. Electrostatic field measurement equipment is also commercially available. It can usually measure voltages up to 30 kV. 4.2.2 Multipath Effects Caused by Surface Feature Diffraction and Attenuation The path of the electromagnetic wave between the transmitter and receiver is rarely direct. There is almost always a multipath. The propagation with the presence of a multipath is different from the propagation in ideal free space conditions. There are at least two paths: the direct path and the reflected path (from the atmosphere and Earth) as shown in Figure 4.8. The reflected component has two parts: coherent and noncoherent. The coher- ent part is determined in regards to amplitude, phase, and direction. It follows the Snell law. The noncoherent part is subject to random characteristics of the scatter- ing terrain and is not deterministic. It is not a plane wave and does not follow the Snell law. It does not come from a certain direction but from the continuum. Table 4.3 gives the electric properties of various types of terrain. The surface wave propa- gates best over sea water, and worst over dry terrain. Figure 4.7 ESD danger symbol.
  • 75. 4.2 Natural Sources of Electromagnetic Interference 65 Receiver d Direct path h2 Transmitter d2 Reflected path h1 d1 Flat earth Curved earth Figure 4.8 Multipath from Earth. The phase difference, ∆, between the direct and reflected path is calculated from 2π ∆= (d1 + d 2 − d ) (4.8) λ where d1 and d2 are the reflected paths and d is the direct path. Whether the terrain is smooth or not will depend on the Rayleigh criterion: λ h≥ (4.9) 8sin Ψ where h is the height of the terrain roughness, λ is the wavelength, and Ψ is the angle of wave incidence (Figure 4.9). If the above condition is fulfilled, the terrain is rough—otherwise it is smooth. In other words, if h is small enough, the dominant reflection will be coherent. 4.2.3 Attenuation by Atmospheric Water The effect of atmospheric hydrometeors is of major concern for satellite to Earth propagation. The main hydrometeors that exist are rain, snow, and dust particles. Rain is the major obstacle because it causes attenuation, phase difference, and depo- larization of radio waves. For analog signals, rain is most significant at frequencies above 10 GHz, and for digital signals above 3 GHz. The loss due to rain is given by Table 4.3 Dielectric Properties of Various Earth Types Earth Type Permittivity r Conductivity Sea water 80 5 Fresh water (river, lakes) 80 0.005 Moist Earth 15–30 0.005–0.01 Rocky terrain 7 0.001 Dry terrain 4 0.001–0.01
  • 76. 66 Electromagnetic Interference h Figure 4.9 Reflection from the rough terrain. L = γ(R ) ⋅ l c (R ) ⋅ p(R ) (4.10) where γ is the attenuation per unit length at rain rate R, le is the equivalent path length at rain rate R, and p(R) is the probability in percentage of rainfall rate R. The attenuation depends on the rain rate, size, temperature, and refractive index of the water. Attenuation is calculated from γ(R ) = a ⋅ R b [dB km] (4.11) where a and b are constants depending on the frequency. At 0°C, the values of a and b are obtained from a = Ga ⋅ f Ea (4.12) Eb b = Gb ⋅ f where the values for Ga, Ea, Gb, and Eb are given in Tables 4.4 and 4.5. The effective length le(R) is used because the rain intensity is not the same over the whole path. It depends on the local climate conditions. It can be approximated from [ ] −1 I e (R ) = 00007R 0. 766 + (0232 − 000018R ) sin θ . . . (4.13) Table 4.4 Values of Ga and Ea Frequency Ga Ea −5 f < 2.9 GHz 6.39 · 10 2.03 −5 2.9 GHz ≤ f ≤ 54 GHz 4.21 · 10 2.42 −2 54 GHz ≤ f ≤ 180 GHz 4.09 · 10 0.6999 f 180 GHz 3.38 −0.151 Table 4.5 Values of Gb and Eb Frequency Ga Ea f < 8.5 GHz 0.158 0.158 2.9 GHz ≤ f ≤ 54 GHz 1.41 −0.078 25 GHz ≤ f ≤ 164 GHz 2.63 −0.272 f > 164 GHz 0.616 −0.0126
  • 77. 4.2 Natural Sources of Electromagnetic Interference 67 where θ is the elevation angle. The probability of the rainfall rate R in percentage is determined by: p(R ) = M 87.66 [ ( 003 ⋅ β ⋅ e −0. 03 R + 02(1 − β) ⋅ e −0. 258 R + 186 ⋅ e −1. 63⋅R . . . )] (4.14) where M is the mean annual rainfall accumulation in [mm] and β is the Rice–Holmberg thunderstorm ratio. Other hydrometeors like snow, vapor, or ice have similar characteristics as rain, but are at least one order of magnitude smaller. Figure 4.10 shows the rain attenua- tion versus frequency and rainfall rate. The size of raindrops depends on weather conditions and rainfall rate. They are given in Table 4.6. The larger the size of the raindrops, the higher the attenuation. The effects of rain can be lessened by using the circle polarization. With circle polarization, the polarization of the electromagnetic wave changes during one period of the wave. It can be clockwise or counterclockwise. In both cases, the trans- mitter and receiver must be synchronized. Circle polarization is widely used in satel- lite communications. 4.2.4 Attenuation by Atmospheric Pollutants Beside rain and other hydrometeors, other particles can also influence the propaga- tion of the electromagnetic wave. The atmosphere consists of several gasses given in Table 4.7. As can be seen from the Table 4.7, the atmosphere mainly consists of nitrogen and oxygen. While hydrometeors influence the propagation of the electromagnetic 100 Rain rate 150 mm/hr t = 20 degrees Celsius 100 mm/hr 50 mm/hr 25 mm/hr 10 12.5 mm/hr Attenuation (dB/km) 5 mm/hr 2.5 mm/hr 1 0.1 1 10 100 300 Frequency (GHz) Figure 4.10 Rain attenuation versus frequency and rainfall rate.
  • 78. 68 Electromagnetic Interference Table 4.6 The Size of Raindrops for Different Types of Precipitation Condition Raindrop Size ( m Haze 0.01–3 Fog 0.01–100 Clouds 1–50 Light rain 3–800 Medium rain (4 mm/hr) 3–1,500 Heavy rain (16 mm/hr) 3–3,000 Table 4.7 Dry Atmosphere Constituents from Sea Level to 90 km High Volume Weight Particle Percentage Percentage Nitrogen 78.088 75.527 Oxygen 20.949 23.143 Argon 0.93 1.282 Carbon dioxide 0.03 0.0456 −3 −3 Neon 1.8 × 10 1.25 × 10 Helium 5.24 × 10−4 7.24 × 10 −3 −4 −5 Methane 1.4 × 10 7.75 × 10 Krypton 1.14 × 10−4 3.30 × 10 −4 −5 −5 Nitrogen oxide 5 × 10 7.60 × 10 Xenon 8.6 × 10−6 3.90 ×10 −5 −5 −6 Hydrogen 5 × 10 3.48 × 10 wave by attenuation and scattering, gasses mostly only attenuate the EM waves. Oxygen has a small permanent magnetic moment, which results in small attenua- tion above 30 GHz. Water vapor can also attenuate EM waves above 10 GHz because it has a permanent electric dipole. The amount of water vapor can vary from 1 mg/m3 in cold dry climates to 30 g/m3 in hot humid climates. This means that in deserts there is almost no water vapor present, whereas in rain forests it can make up about 4% of the atmosphere. Nitrogen, on the other hand, has no permanent electric or magnetic dipole, so it does not attenuate EM waves. Most gases have a negligible influence below 30 GHz. 4.2.5 Sunspot Activity A sunspot is a region on the Sun surface near the equator consisting of magnetic activity with reduced surface temperature (4,000K compared to the surrounding 5,800K). They are visible from the Earth without a telescope. Their numbers and size rise and fall every 11 years. The sunspots have influenced Earth’s climate throughout history. They do not influence solar radiation much, but their magnetic activity influences the ultraviolet and soft X-ray emission levels. They also emit ions, the amount of which depends on the sunspot activity. Both X-rays and ions are
  • 79. 4.3 Manmade Sources of Electromagnetic Interference 69 charged particles, which can interfere with radio electromagnetic waves near the surface of the Earth. They have a strong influence on the ionosphere. When the sun- spot activity is at its maximum, the attenuation in the atmosphere is very high and communication is very difficult to establish. It is then sometimes necessary to switch to higher frequencies for long distance communications. Since the Sun rotates around its own axis, there is also a 27-day sunspot cycle that can also influence the ionization density in the ionosphere. The sunspot activity is regularly observed by telescopes from Earth and satellites. The lower range of radio frequencies is more affected by sunspots than UHF fre- quencies or microwave communications.4.3 Manmade Sources of Electromagnetic Interference Manmade sources of electromagnetic interference can be intentional or uninten- tional. Intentional interference is used when one party wants to disrupt the commu- nication capability of the other party by transmitting an interfering signal in the same frequency band with a higher power than what is used by the second party. Most manmade interference however is unintentional. It comes from bad planning of mobile telephony, bad reuse of frequencies, intermodulation products, other ser- vices using the same frequency bands, and industrial sources (which may not be used for communications at all, but still interfere with useful communications). All communications that are not intended for a certain use are considered interference, regardless of the fact that it is a useful communication for some other users. If inter- ference is in the same communication channel, it will increase the noise in the sys- tem. It is important to know all the potential sources of interference in order to calculate the parameters of the communication link. 4.3.1 Commercial Radio and Telephone Communications Commercial radio and telephone communications are the most widely used com- munication systems. Radio and TV broadcasting are several decades old, and can be found in most undeveloped countries. Cell phone communications are also become more prominent in these locations. Pager networks and private communications mostly exist in developed countries and are not so widely used. Broadcast Systems Broadcast systems include radio, TV, satellite, and any other transmission of audio or video signals to a broad audience. Radio Broadcasting Historically radio broadcasting can be divided into amplitude modulation (AM) and frequency modulation (FM). AM is generally used for larger distances and is not as good in quality as FM radio stations. There is also digital radio, which has the highest quality signal.
  • 80. 70 Electromagnetic Interference AM radio can be divided into long wave (LW), medium wave (MW), and short wave (SW). • LW works in the frequency band from 148.5 kHz to 283.5 kHz. The channels are 9 kHz apart. • MW works in the frequency band of 526.5–1,705 kHz. It is most frequently used for AM radio. There are 117 carrier frequencies in 10 kHz intervals (out- side of United States 9 kHz). Each carrier frequency should not deviate more than ± 20 Hz from the allocated frequency. Modulation frequencies range from 50 Hz to 5 kHz; if it exceeds 5 kHz, the radio frequency bandwidth will exceed 10 kHz and therefore interfere with the adjacent channel. The classes of AM stations based on transmitting power are given in Table 4.8. There are several thousand AM radio stations in the United States alone. • SW uses frequencies above the ones of MW radio stations (i.e., from 2.3 MHz to 26.1 MHz). The channels are separated by only 5 kHz. They usually do not broadcast 24 hours a day, and they sometimes change frequency during the day to compensate for the deterioration of reception conditions. The range is not as large as with MW radio. SW is divided into frequency bands as given in Table 4.9. Table 4.8 MW Radio Stations Class Power (kW) Frequency (kHz) A 10–50 535–1,605 B 0.25–50 1,605–1,705 C 0.25–1 1,230, 1,240, 1,340, 1,400, 1,450, 1,490 D 0.25-50 535–1,605, 1,605–1,705 Table 4.9 SW Radio Bands Frequency Name (MHz) 120m 2.3–2.495 90m 3.2–3.4 75m 3.9–4.0 60m 4.75–5.06 49m 5.9–6.2 41m 7.1–7.35 31m 9.4–9.9 25m 11.6–12.1 21m 13.57–13.87 19m 15.1–15.8 16m 17.48–17.9 13m 21.45–21.85 11m 25.6–26.1
  • 81. 4.3 Manmade Sources of Electromagnetic Interference 71 Sunspots, weather conditions, and whether the station operates at day or night will influence the signal quality and propagation. The power of the transmitters can be from 1W (or less) to 500 kW. FM radio stations use the frequency band from 88 to 108 MHz. They have a much higher quality than AM radio stations. Unlike AM radio stations, which have a very large range (several hundred km), FM stations normally can only be heard up to approximately 100 km from the stations. If there is clear frequency (no other transmitter in the vicinity transmitting at the same or very near frequency), and if the transmitter is placed high on a mountain, this distance may be even larger. The frequency band of 20 MHz is divided into 100 carrier channels with 200 kHz in width, which are placed 200 kHz apart. The frequency deviation should not exceed ±75 kHz, while the stability of the carrier should be ±2 kHz. The maximum powers of the transmitter classes are given in Table 4.10. There are also many digital radio technologies in the world today, both terres- trial and satellite. This technology is still evolving and not a single one has gained acceptance. Most users are still listening to radio stations with their old and very cheap receivers. TV Broadcasting TV broadcasting can be either analog or digital. More and more countries in the world are switching to digital systems. Old TV receivers might still be used with a DVB-T receiver. Analog channels are divided as shown in Tables 4.11 to 4.13. TV standards are not the same in all countries. The National Television Systems Committee (NTSC) standard is used in the United States, Canada, Central America, most of South America, and Japan. NTSC has 525 horizontal lines. Phase Alterna- tion each Line (PAL) is used in Western Europe and China; it has 625 horizontal Table 4.10 FM Radio Stations Class Power (kW) A 6 B 25 B1 50 C3 25 C2 50 C1 100 D 100 Table 4.11 VHF-1 TV Channels Channel Frequency (MHz) 2 54–60 3 60–66 4 66–72 5 76–82 6 82–88
  • 82. 72 Electromagnetic Interference Table 4.12 VHF-3 TV Channels Channel Frequency (MHz) 7 174–180 8 180–186 9 186–192 10 192–198 11 198–204 12 204–210 13 210–216 Table 4.13 UHF TV Channels Frequency Frequency Frequency Frequency Channel (MHz) Channel (MHz) Channel (MHz) Channel (MHz) 14 470–476 32 578–584 50 686–692 68 794–800 15 476–482 33 584–590 51 692–698 69 800–806 16 482–488 34 590–596 52 698–704 70 806–812 17 488–494 35 596–602 53 704–710 71 812–818 18 494–500 36 602–608 54 710–716 72 818–824 19 500–506 37 608–614 55 716–722 73 824–830 20 506–512 38 614–620 56 722–728 74 830–836 21 512–518 39 620–626 57 728–734 75 836–842 22 518–524 40 626–632 58 734–740 76 842–848 23 524–530 41 632–638 59 740–746 77 848–854 24 530–536 42 638–644 60 746–752 78 854–860 25 536–542 43 644–650 61 752–758 79 860–866 26 542–548 44 650–656 62 758–764 80 866–872 27 548–554 45 656–662 63 764–770 81 872–878 28 554–560 46 662–668 64 770–776 82 878–884 29 560–566 47 668–674 65 776–782 83 884–890 30 566–572 48 674–680 66 782–788 — — 31 572–578 49 680–686 67 788–794 — — lines. Sequential Color ‘avec’ Memory (SECAM) is used in France, Russia, and some eastern European countries. It also has 625 horizontal lines. The maximum effective radiated power (ERP) for VHF-1 transmitters is 100 kW, for VHF-3 it is 316 kW, and for UHF transmitters it is 5 MW. While some analog TV systems cease to operate, new digital systems emerge. This trend is present around the world. High definition television is of much higher quality than analog television. The TV channels used for digital TV are the whole VHF-1 and VHF-3 bands, and a part of the UHF (14–36, 38–51) band. Satellite broadcasting operates in the L-band (1,452–1,492 MHz), S-band (2,310–2,360, 2,520–2,655), Ku-band (11.7–12.7 GHz), K-band (17.3–17.8 GHz, 21.4–22 GHz), and Ka-band (40.5–42.5 GHz). The frequency band from 11.7 to 12.2 GHz is used for the fixed satellite service.
  • 83. 4.3 Manmade Sources of Electromagnetic Interference 73 The maximum power flux density on the Earth’s surface should not exceed 2 −137 dBΩ/m for frequencies between 2.5 GHz and 27 GHz in order to not interfere with LOS terrestrial communications. Cell Phone and Pager Networks Cellular phones are part of everyday life in a manner so great that some people have more than one mobile phone. The base stations are everywhere around us. While broadcast transmitters are relatively rare, mobile base stations are much more pres- ent, whether they are placed on highways or inside offices. Cell Phone Networks The Global System for Mobile Communication (GSM) is the most widely used cell phone network in the world. It operates in the 450 MHz band, 900 MHz band, 1800 MHz band, 850 MHz band, and 1900 MHz band. The first three are most commonly used in Europe, Asia, and Africa, and the latter two in North and South America. In Japan, CDMA technology is used. Table 4.14 gives some characteristics for various GSM networks. The power of the transmitters is defined by international regulations as well as local country regulation. Up to 500W per channel (transmitter) of effective radiated power (ERP) is allowed in urban areas. In most cases, only 100W per channel is used. Usually there are 21 channels per sector, with three sectors totaling 63 chan- nels per base station. At the maximum, with a omnidirectional antenna, 96 channels are possible. Thus, 48 kW would be the total maximum power if all the channels were active, but that would be very rare. The power density drops very rapidly in accordance with the distance from the antenna. Pager Networks The pager system is a simplex communication system, which can send short mes- sages to a subscriber. This message can be numeric, alphanumeric, or a voice mes- sage. It is actually a warning or notice for further communications. The receivers are usually very simple and cheap, but the transmission system used for large distances is not. Transmitters are usually high in power (kW) for wide-area paging systems, but for local systems (e.g., the office), which work at 2.4 GHz, the power is in mW. The paging frequencies are given in Table 4.15. Table 4.14 GSM Networks GSM 400 900 1,800 850 1,900 Uplink 450.4–457.6 890–915 1,710–1,785 824–849 1,850–1,920 frequency (MHz) 460.4–467.6 Downlink 478.8–486 925(935)–960 1,805–1,880 869–894 1,930–1,990 frequency (MHz) 488.8–496 Frequency spectrum 7 MHz 35 (25) MHz 75 MHz 25 MHz 70 MHz Duplex separation 10 MHz 45 MHz 95 MHz 45 MHz 80 MHz Carrier spacing 200 kHz 200 kHz 200 kHz 200 kHz 200 kHz
  • 84. 74 Electromagnetic Interference Table 4.15 Paging Frequencies Region Frequency (MHz) United States 35–36, 43–44, 152–159, 454–460, 929, 931 European Union 47.0–47.25, 440–470 Japan 280 Australia 148 The maximum power of the transmitters for 152–153-MHz bands is 1.4 kW, for 153–159 MHz it is 150W, for 454–455 MHz it is 3.5 kW, and for 459–460 MHz it is 150W. Private Networks Private networks differ from public networks as they are open only to users of the network. Private networks are mostly used for corporate networks, which are not connected to global Internet networks for security reasons. Private networks can be established through an Internet connection or broad- band satellite. The Internet can also be wireless on 2.4 or 5.5 GHz (ISM frequency band). There is a lot of interference from other services working in this license-free band (Bluetooth, microwave oven, ZigBee). The Spaceway satellite operating in the Ka-band enables broadband services with up to a 16-Mbps connectivity rate. 4.3.2 Military Radio and Telephone Communications There is a vast range of military transmitters in every army in the world, including communication in HF and microwave ranges for both terrestrial and satellite appli- cations. The complete list of frequencies used by the military would be too large to fit into this book. They can be found in tables of frequency allocation for the specific frequency band of interest. The civil transmitter frequencies in the vicinity of mili- tary bases (both land and sea) should be carefully planned in order not to interfere with the military communications. Furthermore, communication signals from air- crafts can arrive from great distances. Military communications are often coded and encrypted. Portable radios have large power compared to cellular phones. They are robustly made to endure severe conditions such as changes in heat, humidity, mud, dust, and so forth. Radios can also be mounted on vehicles. They are usually higher in power and range than portable versions carried by individuals. As for propagation, the same laws apply for military as well as civilian communications. 4.3.3 Commercial Radar Systems Radar is a system that uses electromagnetic impulses to identify the position of an object. It also determines the altitude, direction, and speed. The object of interest can be an airplane, naval vessel, clouds, or terrain. The word “radar” is an abbrevi-
  • 85. 4.3 Manmade Sources of Electromagnetic Interference 75 ation of radio detecting and ranging. Radar transmits pulses, which are reflected from the objects. From the time of the reflected wave, the position can be calculated. It is used by the army, air traffic, in metrology and astronomy, and by the police. Air Traffic Control Air traffic control is placed on commercial (and also military) airports for prevent- ing collisions of airplanes when taking off or landing. Radar operates on frequencies in the L-band (1–2 GHz) for long distances, in the S-band (2–4 GHz) for medium distances, and the X-band (8–12 GHz) and Ka-band (24–40 GHz) for short distances. The power density from radars should not exceed 5 mW/cm2. Table 4.16 gives air traffic radar frequencies in the United States and European Union. Astronomy Radar in astronomy works on the same principle as radar for air traffic control. However, in this case the objects of interest are placed in the solar system. Radar is also used for weather control. The frequencies are given in Table 4.17. In metrology, radars in the S-band (2–4 GHz), K-band (18–24 GHz), and W-band (75–110 GHz) are used. The power density from radar should not exceed 5 mW/cm2 for this application as well. 4.3.4 Industrial Sources Dielectric heaters, neon signs, X-ray and welding machines, air conditioning, medi- cal devices, fluorescent lights, and lasers can interfere with communication systems. Most international standards for communication equipment in industrial sur- roundings have a limit for susceptibility of either 3 V/m or 10 V/m. The standards of interest are EN 50082-2 (Electromagnetic Compatibility—Generic Immunity Stan- dard—Part 2: Industrial Environment), EN 61000-6-2 (Electromagnetic Compati- bility—Generic Standards—Part 6-2: Immunity for Industrial Environments), and EN 50082-1 (Electromagnetic Compatibility—Generic Immunity Standard, Part 1: Residential, Commercial, and Light Industry, CENELEC). Table 4.16 Air Traffic Control Radar Frequencies Region Frequency (MHz) United States 1,300–1,350, 2,700–2,900, 3,500–3,650, 9,000–9,200, 13,250–13,400 European Union 1,215–1,350, 2,700–3,100, 3,300–3,500, 5,250–5,725 Table 4.17 Weather Radar Frequencies Region Frequency (MHz) United States 5,600–5,650, 9,300–9,500 European Union 5,250–5,570, 5,650–5,850
  • 86. 76 Electromagnetic Interference 4.3.5 Intentional Interference An example of intentional interference is jamming. The purpose of intentional inter- ference is to disable the communication of the adversary at a minimum cost. On the other hand, the goal of the party trying to communicate is to develop a system immune to interference. A total immunity from interference is impossible. It has to be assumed that the party trying to jam the communication knows the frequency but not the spreading codes. The shape of the signal should be chosen in such a way that it leaves the jammer with no other option except the broadband Gauss noise. There are several ways of possible interference. Figure 4.11 shows the spectral power densities of interference against communication systems. The width of the spread spectrum frequency band is B. If one party uses fre- quency hopping (S1 and S2) inside this band B, the interfering Gaussian noise spec- trum can be done in three ways. The first is the low spectral noise density in the whole frequency band B [Figure 4.11(a)]. In this way, there will be interference in the entire system—but this will not pose a great problem. The second method is increased noise in one part of the spectrum [Figure 4.11(b)]. Here the damage will be great in some cases, but in other cases there may be no damage at all. The last method [Figure 4.11(c)] is to have large power in just one small portion of band B, but at the same time hop the position of this band. The damage is devastating if the interference coincides with signal S2. The party that wants to protect its communication system from jamming should use frequency hopping or the time hopping spread spectrum and an antenna system with high directivity. This chapter is the last of the introduction chapters. The next chapters will deal with active and passive interference control. A A S1 S2 (a) f f B B A A S1 S2 (b) f f B B A A S2 S1 (c) f f B B Figure 4.11 (a–c) Various types of interfering spectral densities.
  • 87. 4.3 Manmade Sources of Electromagnetic Interference 77Selected Bibliography Freeman, R.L., Radio System Design for Telecommunication, New York: Wiley-IEEE, 2006. Lin, G., and Alvarado, M., “Reviewing EU EMC Generic Standards,” EE-Evaluation Engineer- ing, July 2000, pp. 50–57. MIL-STD-464 Electromagnetic Environmental Effects, Requirements for Systems, 18 March 1997, Morrison, R., Noise and Other Interfering Signals, New York: John Wiley & Sons, 1992. Kocharyan, V., and D. Tolman, “An Express Diagnostic Method for ESD Simulators and Stan- dardized ESD Test Stations,” Proc. 2003 IEEE International Symposium on Electromagnetic Compatibility, Vol. 2, August 18–22, 2003, pp. 708–712. Rakov, V. A., “Transient Response of a Tall Object to Lightning,” IEEE Transactions on Electro- magnetic Compatibility, Vol. 43, No. 4, November 2001, pp. 654–661. Riaziat, M. L., Introduction to High-Speed Electronics and Optoelectronics, New York: John Wiley & Sons, 1996. Van der Laan, P. C. T., and A. P. J. van Deursen, “Reliable Protection of Electronics Against Lightning: Some Practical Applications,” IEEE Transactions on Electromagnetic Compatibility, Vol. 40, No. 4, November 1998, pp. 513–520. Young, P. H., Electronic Communication Techniques, Columbus, OH: Merrill Publishing Co., 1985.
  • 88. CHAPTER 5Filter Interference Control5.1 Filters A filter is an electronic element with two ports that separates one frequency band from another. The input signal, or excitation, goes through the filter to the output port whose signal is called the response. The application of filters ranges from acoustics to atomic clocks. In telecommunications, bandpass filters are used in the audio frequency range for speech processing. Filters can be divided in several ways. One division is into passive and active fil- ters. A passive filter does not require an external power source in order to operate, while an active filter does. A passive filter is made of inductors and capacitors or their equivalent (microwave waveguides). Active filters are made of resistors, capac- itors, and amplifiers. Filters can be analog or digital, but analog filters are longer in use. They work with analog or continuous signals, and their response signal is a continuous signal. Digital filters use analog-to-digital converters (ADC) to process the analog signal. The output from the filter or response is represented in digital numbers. In order to obtain the analog signal, a digital-to-analog converter (DAC) is needed. Filters can be divided regarding the function they perform. There are lowpass, highpass, bandpass, and bandstop filters. A lowpass filter passes all frequencies up to the cutoff frequency and stops all of the frequencies above it. A highpass filter stops all frequencies up to the cutoff frequency and passes all of the frequencies above it. A bandpass filter passes all frequencies between two frequencies of interest and stops all of the frequencies below and above them. A bandstop filter stops the frequencies between two frequencies of interest and passes all of the other frequen- cies above and below them. The transfer function of the filter T(jω) [sometimes also called H(jω)] is a func- tion represented with gain (amplitude) and phase characteristics. In this chapter, only the amplitude will be dealt with. It shows how the filter changes the input sig- nal at the output response depending on the frequency. This response can be shown graphically or mathematically (the Bode plots). The transfer function of the filters depends on the type used and the required application. It is the response of the filter depending on whether the filter is a lowpass, highpass, bandpass, or bandstop and whether the type of filter is Butterworth, Chebyshev, Bessel, or another type. 79
  • 89. 80 Filter Interference Control 5.1.1 Lowpass Filter The response of an ideal lowpass filter is shown in Figure 5.1. It must pass all of the frequencies from 0 Hz up to the cutoff frequency fc without attenuation; at the same time, it must stop (attenuate) all the frequencies above the cutoff frequency to amplitude 0. Although the ideal response is desired in many situations, it is impossible to cre- ate it in the real world. The response of a real lowpass filter will be considered to pass all the frequencies up to the point where the amplitude (magnitude) drops by 2/2 or down to 0.707 (Figure 5.2)—in other words, by 3 dB. The 3-dB point determines the width of the communication channel. This means that the frequencies that are close to the cutoff frequency fc will still be passed (although somewhat attenuated), even though they are not supposed to be. They will however have a smaller amplitude. How much smaller, and at which frequen- cies this amplitude will reach 0, will depend on the type of the filter and its quality. The filters determine the quality and therefore the price of the electronic equipment. With good filters, the communication channels can be packed more closely to each other without interference between them. The goal in designing the filters is to have a curve as steep as possible from the cutoff frequency up to the amplitude of 0 (i.e., the filter should pass as little of the frequency band as possible above the cutoff frequency). Looking closely at Figure 5.2, it can be seen that not only are the frequencies passed above the cutoff frequency, but also the attenuation already starts below the cutoff frequency, which is certainly not intended. That is why, in designing the lowpass filter, it will be desirable to improve this setback or to have the attenuation start at the cutoff frequency and not before. 5.1.2 Highpass Filter A highpass filter response is shown in Figure 5.3. It should stop (attenuate) all the frequencies from 0 Hz up to the cutoff frequency fc, and at the same time pass (atten- uate) all the frequencies above the cutoff frequency to infinity. Again, this is the ideal response, which cannot be achieved. A 1 0 fc f Figure 5.1 Ideal lowpass filter response.
  • 90. 5.1 Filters 81 A 1 0.707 0 fc f Figure 5.2 Real lowpass filter response. A 1 0 fc f Figure 5.3 Ideal highpass filter response. The response of a real highpass filter will pass all of the frequencies higher than the cutoff frequency up to the infinity. The cutoff frequency starts at the point where the amplitude (voltage) is at 0.707 (−3 dB) of the input signal (Figure 5.4). If the power is used rather than voltage, then 3 dB is 50% less in power or 0.5. Figure 5.4 shows that the responses of lowpass and highpass filters are actually mirrored. This means that the highpass filter and the bandstop filter have similar responses. 5.1.3 Bandpass Filter The response of an ideal bandpass filter is shown in Figure 5.5. It should pass all of the frequencies from the first cutoff frequency fc1 to the second cutoff frequency fc2 without attenuation. At the same time, it should stop (attenuate) all of the frequen- cies below the first cutoff frequency and above the second cutoff frequency to the amplitude of 0. The ideal bandpass filter will pass the frequency band between the two cutoff frequencies (0.707 amplitude) as shown in Figure 5.6. This has to be kept in mind
  • 91. 82 Filter Interference Control A 1 0.707 0 fc f Figure 5.4 Real highpass filter response. A 1 0 fc1 fc2 f Figure 5.5 Ideal bandpass filter response. A 1 0.707 0 fc1 fc2 f Figure 5.6 Real bandpass filter response. when planning the channel spacing in a real communication system—otherwise, the channels might interfere with each other.
  • 92. 5.1 Filters 83 5.1.4 Bandstop Filter The response of an ideal bandstop filter is shown in Figure 5.7. It should stop all the frequencies from the first cutoff frequency fc1 to the second cutoff frequency fc2. At the same time it is supposed to pass (without attenuation) all the frequencies below the first cutoff frequency and above the second cutoff frequency. The real response of the bandpass filter will attenuate some of the frequencies below the first cutoff frequency (0.707 amplitude) and also some of the frequencies above the second cutoff frequency (Figure 5.8). This filter is used when only one band is to be attenuated and everything else is to be passed. 5.1.5 Resonator A resonator is the most basic filter, intended to pass only one frequency or resonate on only one frequency and filter out (attenuate) all other frequencies. The resonator response is shown in Figure 5.9. The quality of the resonator or selectivity is defined with the Q-factor as fc fc Q= = (5.1) ∆f f c 2 − f c1 A 1 0.707 0 fc1 fc2 f Figure 5.7 Ideal bandstop filter response. A 1 0.707 0 fc1 fc2 f Figure 5.8 Real bandstop filter response.
  • 93. 84 Filter Interference Control A 1 0.707 0 fc1 fc fc2 f Figure 5.9 Resonator response. where frequencies fc2 and fc1 are those where the voltage drops to 0.707 of the inci- dent value (power is at 50%). The more selective the resonator (filter), the smaller (narrower) the ∆f, and the larger the Q. The simplest resonance can be done with the serial and parallel resonant circuit (Figure 5.10), which consists of resistor R, inductance L, and capacitor C. The total impedance of serial resonance is 1 ⎛ 1 ⎞ Z = R + jX L − jXC = R + jωL − j = R + j ⎜ ωL + ⎟ (5.2) ωC ⎝ ωC ⎠ where ω = 2πf. This means that the impedance will have both a real and imaginary part. The latter can be either inductive or capacitive, depending on the frequency and values of L and C. The resonance will appear when Im{Z} = 0, that is, when XL = XC. Then, most of the energy will be transferred from the generator to the load, which can be an antenna in the transmitter system. To have this, 1 1 ωL − = 0 ⇒ ωL = (5.3) ωC ωC must be valid. The above expression will be true for serial resonant frequency fsr R I U L C Figure 5.10 Serial RLC resonance.
  • 94. 5.2 Analog Filters 85 ILC I IR IL IC R L C U Figure 5.11 Parallel RLC resonance. 1 1 ω sr = ⇒ f sr = (5.4) LC 2 π LC In serial resonance, the voltages on L and C will be equal, but of opposite directions. The serial resonance is not the only possible resonance. There is also a parallel resonance of R, L, and C. It is shown in Figure 5.11. The total admittance of the above circuit is Y = G + j( BC − B L ) (5.5) where BC (1/XC) and BL (1/XL) are capacitive and inductive susceptances, and G (1/R) is the conductance. The imaginary part must again equal zero, or Im{Z} = 0. This will be fulfilled again for 1 1 ω pr = ⇒ f pr = (5.6) LC 2 π LC5.2 Analog Filters There are many analog filter types; the following are used the most: Butterworth, Chebyshev, Bessel, and elliptic. Analog filters can be either passive or active. Passive filters use resistors, inductors, and capacitors, while active analog filters use resis- tors, capacitors, and operational amplifiers as mentioned before. Active filters have higher Q-factors, whereas passive filters with inductors do not. 5.2.1 Butterworth Filter Butterworth filters have flat attenuation in the passband region without any ripple. At the cutoff frequency, fc, the attenuation is 3 dB (50% power). The frequency response of an Nth-order Butterworth lowpass filter is obtained by the transfer function T(jω) as
  • 95. 86 Filter Interference Control 1 T( jω) = 2N (5.7) ⎛f ⎞ 1+ ⎜ ⎟ ⎝ fc ⎠ where fc is the cutoff frequency and N is the filter order. The response is flat both at 0 and infinity. The N number is related to the total number of reactive elements (inductors or capacitors) in the lowpass or highpass filter. For a bandpass or bandstop filter, the number of required reactive elements is twice as high as for lowpass or highpass filters. Therefore, the filter order determines the steepness of the slope in the bandstop part. The higher the filter order, the steeper the slope. The slope for the lowpass Butterworth filter is −20N dB/decade (or approximately 6N dB/octave) as shown in Figure 5.12. The frequency response is usually shown in logarithmic scale, where the ratio of 10 to 1 is called a decade (the ratio of 2 to 1 is called an octave). That is why (5.7) is used for drawing Figure 5.12 in decibels according to T( jω) = 20 log10 T( jω) (5.8) 5.2.2 Chebyshev Filters Chebyshev filters, in comparison to Butterworth filters, have a ripple in the passband region. They are more similar to the ideal filter except for the ripple. Above the cutoff frequency, Chebyshev filters have much higher attenuation than Butterworth filters. The transfer function is calculated from 1 T( jω) = (5.9) ⎛f ⎞ 1+ ε T ⎜ ⎟ 2 2 N ⎝ fc ⎠ where ε is a real constant whose value is less than 1; it determines the ripple of the filter calculated from 0 −5 −10 n=1 T, dB n=2 −15 n=3 −20 −25 −30 0.1 0.4 0.7 1.0 1.3 1.6 1.9 2.2 2.5 2.8 f Figure 5.12 Butterworth filter response function for different orders.
  • 96. 5.2 Analog Filters 87 ( ) 0.5 ε = 10 0.1 r − 1 (5.10) with r being the positive real number and TN(f/fc) the Nth order Chebyshev polyno- mial calculated from ⎧ ⎛ −1 ⎛ f ⎞ ⎞ f ⎪ cos ⎜ N cos ⎜ ⎟ ⎟ , ≤1 ⎛f ⎞ ⎪ ⎝ ⎝ f c ⎠⎠ fc TN ⎜ ⎟ = ⎨ (5.11) ⎝ fc ⎠ ⎪ ⎛ ⎛ f ⎞⎞ f cosh ⎜ N cosh −1 ⎜ ⎟ ⎟ ≥1 ⎪ ⎝ ⎝ f c ⎠⎠ f c ⎩ The Chebyshev filter has three parameters, ε, fc, and N. Figure 5.13 shows the Chebyshev filter response for r = 1, and a different order N in the logarithmic scale. A ripple is visible below the cutoff frequency. For different values of r, different slopes can be achieved. 5.2.3 Bessel Filters Bessel filters are used for reducing nonlinear phase distortion (i.e., they have a flat delay). The transition from passband to stopband is much slower than with other filters. It is the only one of the filters mentioned in this chapter where the phase response is important. The transfer functions for N = 1, 2, and 3 are given as 1 T( jω) = , N =1 f 2 +1 3 T( jω) = , N =2 (5.12) f 4 + 3f 2 + 9 15 T( jω) = , N =3 f 6 + 6f 4 + 45 f 2 + 255 1 0 −1 −2 −3 n=1 −4 n=2 T, dB −5 n=3 −6 −7 −8 −9 −10 0.1 0.4 0.7 1.0 1.3 1.6 1.9 2.2 2.5 2.8 f Figure 5.13 Chebyshev filter response for a different order and for r = 1.
  • 97. 88 Filter Interference Control where N is the order of the polynomial. The response functions (amplitude) are shown in Figure 5.14. 5.2.4 Elliptic Filters Elliptic filters have ripples in both passbands and stopbands. They also have a very fast transition between passbands and stopbands. They are also called Cauer filters, and are used in communications where multiple carriers exist. The transfer function is 1 T( jω) = , (5.13) ⎛f ⎞ 1+ ε F ⎜ ⎟ 2 2 N ⎝ fc ⎠ where ε is a ripple factor and FN is the Jacobian elliptic function. The calculation of this filter type is not easy. There are other types of filters such as Gaussian, Legendre, and Linkowitz-Riley filters, but the most frequently used ones are described above. The filters mentioned above can be either passive or active. Passive filters use R, L, and C components, while active filters use operational amplifiers instead of the L component. 5.2.5 Passive Filters Passive filters do not require external power. For low- or high-pass filters, RL or RC combinations can be used. For bandpass filters RLC combinations are used. 1 0 −1 −2 −3 n=1 −4 n=2 T, dB −5 n=3 −6 −7 −8 −9 −10 0.1 0.4 0.7 1.0 1.3 1.6 1.9 2.2 2.5 2.8 f Figure 5.14 Bessel filter response for a different order of N.
  • 98. 5.2 Analog Filters 89 Lowpass RL Filter The lowpass RL filter is shown in Figure 5.15. It consists of a resistor and imped- ance in an L shape. This is a first-order (N = 1) filter, because it has only one reactive element. The transfer function of this filter is given as 1 T( jω) = , (5.14) 2 ⎛f ⎞ 1+ ⎜ ⎟ ⎝ fc ⎠ where the cutoff frequency fc is obtained from R R fc = = (5.15) 2πL ωL Lowpass RC Filter The lowpass RC filter is shown in Figure 5.16. It consists of resistor and conductor, also in an L shape. It is a first-order (N = 1) filter as well. The transfer function of this filter is given as 1 T( jω) = , (5.16) 2 ⎛f ⎞ 1+ ⎜ ⎟ ⎝ fc ⎠ L R Figure 5.15 Lowpass RL filter. R C Figure 5.16 Lowpass RC filter.
  • 99. 90 Filter Interference Control where the cutoff frequency fc is obtained from 1 1 fc = = (5.17) 2 πRC ωRC Highpass RL Filter The highpass RL filter is shown in Figure 5.17. It consists of a resistor and imped- ance in an L shape. It differs from the lowpass RL filter in the position of the ele- ments—R and L exchanged places. The transfer function of this filter is given as 1 T( jω) = (5.18) 2 ⎛f ⎞ 1+ ⎜ c ⎟ ⎝f ⎠ where the cutoff frequency fc is obtained from R R fc = = (5.19) 2πL ωL Highpass RC Filter The highpass RC filter is shown in Figure 5.18. It consists of a resistor and conduc- tor with a changed position or R and C compared to the lowpass RC filter. R L Figure 5.17 Highpass RL filter. C R Figure 5.18 Highpass RC filter.
  • 100. 5.3 Digital Filters 91 The transfer function of this filter is given as 1 T( jω) = (5.20) 2 ⎛f ⎞ 1+ ⎜ c ⎟ ⎝f ⎠ where the cutoff frequency fc is obtained from 1 1 fc = = (5.21) 2 πRC ωRC It can be seen that the cutoff frequencies for the RC lowpass and RC highpass are the same. The same applies for RL highpass and RL lowpass filters as well. Bandpass or bandstop filters require two sets of L or C components (each for its cutoff frequency). 5.2.6 Active Filters Instead of inductors, active filters use both operational amplifiers and R and C com- ponents. Their transfer function can approach ideal filters more closely than passive filters, and there can be an amplification of the signal, which is compared only to the attenuation in passive filters. Figure 5.19 shows the lowpass active filter of the first order. The cutoff frequency is given with 1 1 fc = = (5.22) 2 πR1 C ωR1 C The gain in the passband is equal to −R1/R2, and in the stopband part it drops by 20 dB/decade. Figure 5.20 shows the highpass active filter of the first order. The cutoff frequency is given with 1 2 fc = = (5.23) 2 π R 2 C ωR 2 C The gain in the passband is also equal to –R1/R2. More complex filters can be achieved with higher orders (N > 1) (i.e., with more C elements).5.3 Digital Filters There is usually an analog-to-digital converter (ADC), a microprocessor acting as digital filter, and a digital-to-analog converter (DAC) in digital filters, as shown in Figure 5.21. They are used in modern communication systems.
  • 101. 92 Filter Interference Control C R1 R2 Figure 5.19 First-order lowpass active filter. R1 C R2 Figure 5.20 First-order highpass active filter. x(t) xn yn y(t) ADC Digital filter DAC Figure 5.21 Digital filters. There are two main types of digital filters: finite impulse response (FIR) and infinitive impulse response (IIR). FIR filters are sometimes called nonrecursive fil- ters and IIR filters are known as recursive filters. The advantage of digital filters is that they can be programmed and stored in the memory of the processor. They can also be reprogrammed without the change of hardware, whereas with analog filters this is not possible. Digital filters are also more stable than analog filters regarding temperature and time.
  • 102. 5.3 Digital Filters 93 5.3.1 FIR Filters The impulse response of FIR filters has a finite length. The response lasts N + 1 sam- ples for the Nth filter order and then becomes zero. If a time dependable analog signal is used at the input of a filter x(t), it must be converted into a digital signal. This is done by discretization or taking the samples in time intervals of ∆t. The sampled value of x at discretization time ti = i∆t will be x i = x (t i ) (5.24) The digital values from the analog to digital converter (ADC) will have the sequence x0, x1, x2, x3, ..., xn, where x0 is the sampled value at t = 0, x1 the sampled value at ∆t, x2 is the sampled value at 2 · ∆t, xn is the sampled value at n · ∆t and so forth. The digital output from the filter will have the sequence of values y0, y1, y2, y3, ..., yn. The exact values of y will depend on the values of x and the function of the digital filter. The values of y are then fed to the digital-to-analog converter (DAC) to obtain analog values again. The delay filter, yn = xn−1, can be realized by taking the output value at the time of (n − 1) · ∆t, or y 0 = x −1 y1 = x 0 y2 = x1 (5.25) y3 = x 2 y n = x n −1 If the same filter would take the output values at intervals n · ∆t, it would just be an all-pass filter. Usually, all values of x before t = 0 are considered to be zero. The filter sums the current value xn and the previous value xn−1: y 0 = x 0 + x −1 y1 = x 1 + x 0 (5.26) y2 = x 2 + x1 y3 = x 3 + x 2 The simple lowpass filter can be made by calculating the arithmetic mean of the current and the previous value, that is, yn = (xn + xn−1)/2, or y 0 = ( x 0 + x −1 ) 2 y1 = ( x 1 + x 0 ) 2 (5.27) y2 = ( x 2 + x1 ) 2 y3 = ( x 3 + x 2 ) 2
  • 103. 94 Filter Interference Control The order of the digital FIR filter is the number of previous inputs needed for the calculation of the current filter output. The filter mentioned above is of the first order, since only one previous input was used. Thus, depending on the order, the FIR digital filter can be written as yn = a0 x n 0 order y n = a 0 x n + a1 x n −1 first order (5.28) y n = a 0 x n + a1 x n −1 + a 2 x n − 2 second order The transfer function of the filter describes the filter function, which depends on current and previous values of input for FIR filters. For this purpose, the delay func- tion, z−1, must be introduced. It gives the previous value of the sequence or delay of the same. If the delay function is applied to the input value xn, the output will be the previ- ous value, that is, xn−1, or z −1 x n = x n −1 (5.29) If the input sequence is for example equal to x0 = 5, x1 = 3, x2 = 6, x3 = 2, then −1 −1 −1 −1 −1 z x0 = 0, z x1 = 5, z x2 = 3, z x3 = 6, and so forth. It is assumed that x = 0. The delay does not have to be restricted to the previous input only, so the fol- lowing applies: ( z −1 z −1 x n )= z −1 x n −1 = x n − 2 (5.30) or z −1 z −1 = z −2 (5.31) giving z −2 x n = x n − 2 (5.32) If needed, more delay can be used. The transfer function of FIR filters is as follows: ( y n = a 0 + a1 z − 1 + a 2 z − 2 x n ) (5.33) The general diagram of a FIR digital filter with delay functions is shown in Figure 5.22. 5.3.2 IIR Filters The impulse response has an infinite length of numbers. Usually it is best to design an analog filter (i.e., Butterworth, Chebyshev, or Bessel) and then convert it into a digital filter.
  • 104. 5.3 Digital Filters 95 x(n) a0 y(n) z−1 a1 z−1 a2 z−1 Figure 5.22 FIR digital filter. While for FIR or nonrecursive filters, the current output (yn) depends only on the current input current (xn) and/or previous (xn−1, xn−2, ...) inputs, the IIR or recur- sive filter’s current output also depends on previous outputs (yn−1, yn−2, ...). In the recursive digital filter the current output depends on the current input and the previous output y n = x n − y n −1 (5.34) or y 0 = x 0 − y −1 y1 = x 1 − y 0 (5.35) y 2 = x 2 − y1 where y−1 is usually taken to be 0. If the values of y−1 are used in the following expressions, the above becomes y 0 = x 0 − y −1 = x 0 y1 = x 1 − y 0 = x 1 − x 0 (5.36) y 2 = x 2 − y1 = x 2 − x 1 − x 0 It can be seen that the current output yn is equal to the difference of the current input and all previous inputs.
  • 105. 96 Filter Interference Control For example, in the fifth time interval, the IIR or recursive filter will have the expression y5 = x 5 − y 4 (5.37) In order to have the same function performed with the nonrecursive FIR filter, the function should be: y5 = x 5 − x 4 − x 3 − x 2 − x 1 − x 0 (5.38) This means that the nonrecursive filter would require much more time, opera- tion, and memory to operate as the recursive filter. The order of the recursive IIR digital filter is the largest number of previous input or output values needed to calculate the current output. The lowest order of the IIR filter is the first order; if it were of the zero order, it would not be a recursive filter! The IIR digital filter functions depending on the order can be written as b 0 y n + b1 y n −1 = a 0 x n + a1 x n −1 1st order (5.39) b 0 y n + b1 y n −1 + b 2 y n − 2 = a 0 x n + a1 x n −1 + a 2 x n − 2 2nd order The transfer function of IIR filters is similar to those of FIR filters. The same delay function is valid for output values as well: z −1 y n = y n −1 (5.40) If the second order filter is used, then we will have z −1 y n = y n −1 z −2 y n = y n − 2 (5.41) z −1 x n = x n −1 z −2 x n = x n − 2 and substituting the above expression in the second-order function will yield ( y n b 0 + b1 z − 1 + b 2 z − 2 ) = x (a n 0 + a1 z − 1 + a 2 z − 2 ) (5.42) The above expression can be then written as the transfer function of the second- order IIR filter, showing the dependence of the current output on the current input and delay coefficients: yn = (a 0 + a1 z −1 + a 2 z −2 )x (5.43) (b ) −1 −2 n 0 + b1 z + b2 z A general diagram of an IIR digital filter with delay functions is shown in Figure 5.23.
  • 106. 5.4 Microwave Filters 97 x(n) a0 y(n) z−1 z−1 a1 b1 z−1 z−1 a2 b2 z−1 z−1 Figure 5.23 IIR digital filter.5.4 Microwave Filters A microwave filter is a two-port network used for controlling the frequency response in a microwave system by passing the frequencies in a passband and atten- uating the frequencies in the stop band. The types are the same as in other filter types: lowpass, highpass, bandpass, and bandstop. The electric circuits are similar to those of analog filters. The only difference is that the L and C elements are realized differently at high microwave frequencies than at lower frequencies. Microwave filters can be realized as lumped element, waveguide cavity, and dielectric. 5.4.1 Lumped-Element Filters The inductor and capacitor at microwave frequencies (gigahertz) can be replaced with transmission lines; the inductor can be made with a short-circuited stub, whereas the capacitor can be made with an open stub as shown in Figure 5.24. This is done with the Richard’s transformation. Inductive reactance is given as jX L = jωL = jL tan βl (5.44) where βl = ωl/vp = 2π. The capacitive susceptance is given as jBC = jωC = jC tan βl (5.45) For the cutoff frequency, it must be tan βl = 1 or βl = π/4. With β = 2π/λ, it fol- lows that the stub length l should be l = λ/8, where λ is the wavelength of the line at the cutoff frequency ωc = 2πfc.
  • 107. 98 Filter Interference Control λ/8 at ωc L S.C. jXL jXL Z0 = L λ/8 at ωc C O.C. Z0 = 1/C jB C jB C Figure 5.24 Richard’s transformation of inductor and capacitor. In designing the microwave lumped filter, the first task is to convert the elec- tronic circuits with L and C elements (i.e., Chebyshev or Butterworth) into an open circuit or short circuit using Richard’s transformation. The exact normalized proto- type element values for each filter type can be calculated or found in tables. Next, the serial short circuit stubs must be converted into parallel open stubs because only they can be realized in microstrip lines. This is done with the Kuroda transforma- tion. Before that, a unit element must be added to both ends of the filter (i.e., 50Ω, if this is the characteristic impedance of the generator). Unit elements do not influence the filter since they are matched to the generator and load. The Kuroda transforma- tion is shown in Figure 5.25. The transformation is achieved using Z2 n2 = 1+ (5.46) Z1 In the end, the filter is realized in a microstrip line as shown in Figure 5.26. It can be done on a printed circuit board (PCB). The order of the filter is equal to the number of stubs. 5.4.2 Waveguide Cavity Filters The cavity notch filter was mentioned in Section 1.5. It is often used for FM radio stations. Waveguide filters are made from resonators coupled together by parallel induc- tive irises, which make the resonators. Their circular openings have an inductive character. They are shown in Figure 5.27. The irises are spaced along the waveguide at λg/2, where λg is the guide wave- length of each resonator. The equivalent circuit is shown in Figure 5.28.
  • 108. 5.4 Microwave Filters 99 S.C. Series stub Z 1 /n 2 l l l Z1 Z 2 /n 2 l Z2 Unit element Unit element O.C. Shunt stub Figure 5.25 Kuroda transformation. Z0 Z0 Z 01 Z 02 Z s1 Z s2 Z s3 Figure 5.26 Third-order microstrip lowpass filter. b a l1 l2 l3 Figure 5.27 Waveguide cavity filter. This type of filter can be used for the realization of bandpass or bandstop filters with a high Q factor. The size of the openings will determine the coupling strength, which will influence the bandwidth as well as the shape of the filter transfer function (Chebyshev or close to ideal filter). The openings can also be rectangular, or the inductive element can be created with a vertical screw (column) inserted inside the
  • 109. 100 Filter Interference Control l1 l2 l3 Z0 jX1 Z0 jX2 Z0 jX3 Z 0 jX4 Figure 5.28 Equivalent circuit of waveguide resonator. waveguide. If it is inserted a little, it will behave as a capacitor (the electric field is stronger at smaller distances from the top to the bottom of the waveguide), and when it connects the top and bottom, it will behave as an inductor (the electric field is equal to zero and hence the magnetic field is at a maximum, which is a character- istic of inductors). This is shown in Figure 5.29. When the screw is inserted only a little inside the waveguide, the electric field gets stronger and thus behaves like a capacitor [Figure 5.29(a)]. When the top and bottom of the waveguide are connected, the inserted screw behaves like an inductor, because the electric field is zero at this point [Figure 5.29(b)]. In this way, inserting a screw in or out can change the resonant frequency in some small bandwidth. By adding more resonators (screws) in a series or parallel, a different shape and bandpass or bandstop can be achieved. 5.4.3 Dielectric Resonator If instead of air a material with higher permittivity (εr 20) is used, the dimension of the resonator can be much smaller—up to two orders of magnitude. Dielectric reso- nators usually come in the shape of a small disc or cube (Figure 5.30). The electromagnetic field is concentrated inside the dielectric material, so the dimensions of the dielectric resonator are much smaller than the waveguide cavity resonator for the same frequency resonance (mode). Due to small tangent loss, the b “C” b “L” a a Figure 5.29 Capacitor and inductor in a waveguide. z a y d x εr >>1 Figure 5.30 Dielectric resonator.
  • 110. 5.4 Microwave Filters 101 Table 5.1 Values of (k0) for TEmn Modes n m 0 1 2 3 4 5 6 7 1 3.832 1.841 3.054 4.201 5.317 6.416 7.501 8.578 2 7.016 5.331 6.706 8.015 9.282 10.520 11.735 12.932 3 10.173 8.536 9.969 11.346 12.682 13.987 — — 4 13.324 11.706 13.170 — — — — — Table 5.2 Values of (k0) for TMmn Modes n m 0 1 2 3 4 5 6 7 1 2.405 3.832 5.136 6.380 7.588 8.771 9.936 11.086 2 5.520 7.016 8.417 9.761 11.065 12.339 13.589 14.821 3 8.654 10.173 11.620 13.015 14.372 — — — 4 11.792 13.323 14.796 — — — — — quality factor can be more than 10,000. Such resonators are used in the frequency range from 300 MHz to 300 GHz. As with the waveguide cavity, the resonant modes are the same as in circular cavity waveguides (TE and TM modes) as shown in Tables 5.1 and 5.2. The resonant frequency of the desired mode of propagation (TEmn or TMmn) of the dielectric resonator can be calculated from c ⋅ k0 fc = (5.47) 2π ε r ⋅ a where c is the speed of light, the values of k0 are taken from Tables 5.1 and 5.2, and a is the radius of the dielectric.Selected Bibliography Caviacchi, T. J., Digital Signal Processing, New York: John Wiley & Sons, 2000. Douglas, S. C., “Adaptive Filtering,” in Digital Signal Processing, V. K. Madisetti and D. B. Wil- liams, (eds.), Boca Raton, FL: CRC Press, 1999. Dunlop, J., and D. G. Smith, Telecommunication Engineering, 3rd ed., London, U.K.: Chapman and Hall, 1994. Di Paolo, F., Network and Devices Using Planar Transmission Devices, Boca Raton, FL: CRC Press, 2000. Haykin, S., “Adaptive Systems for Signal Process,” in Advanced Signal Processing Handbook, S. Stergiopoulos, (ed.), Boca Raton, FL: CRC Press, 2001. “Introduction to Digital Filters,” Massara, R.E., et al., “Active Filters,” The Electrical Engineering Handbook, R. C. Dorf (ed.), Boca Raton, FL: CRC Press, 2000. Paul, H., and P. E. Young, Electronic Communication Techniques, New York: Merril Publishing Co., 1985.
  • 111. 102 Filter Interference Control Pozar, D. M., Microwave Engineering, 2nd ed., New York: John Wiley & Sons, 1998. Rahman, J., et al., “Filters,” in Measurement, Instrumentation and Sensors Handbook, J. G. Webster, (ed.), Boca Raton, FL: CRC Press, 1999. Rosa, A. J., “Filters (Passive),”in Engineering Handbook, R. C. Dorf, (ed.), Boca Raton, FL: CRC Press, 2000. Whitaker, J. C., “Circuit Fundamentals,” in The Resource Handbook of Electronics, J. C. Whitaker, (ed.), Boca Raton, FL: CRC Press, 2001.
  • 112. CHAPTER 6Modulation Techniques6.1 Signal Processing and Detection At the receiving end of a digital communication system, the decision whether the bit value is 0 or 1 must be made. This decision takes place after the demodulation pro- cess and sampling of the waveform. This should be done with as little errors as pos- sible. The errors may come from filtering and noise in the communication signal. The most common noise in the radio channel is the Gaussian or white noise, which is present in every communication signal with the same spectral density from very low frequencies up to 1012 Hz. In a binary channel, the transmitted signal si(t) in the time interval (0, T) will have the following form: ⎧ s (t ) 0 ≤ t ≤ T "1 " s i (t ) = ⎨ 1 (6.1) ⎩ s 2 (t ) 0 ≤ t ≤ T "0" The received signal, r(t), will be degraded by the noise, n(t), and possibly by the pulse response, hc(t), as r(t ) = s i (t )∗ hc (t ) + n(t ) i = 1, 2, K, M (6.2) where n(t) is the mean white noise and * is the convolution operator. If the channel is ideal (i.e., there is no binary transmission distortion), hc(t) will not introduce the degradation, and the receiving signal r(t) can be written as r(t ) = s i (t ) + n(t ) i = 1, 2 0≤ t ≤ T (6.3) Figure 6.1 shows the demodulation and detection process at the receiver end. Demodulation determines the waveform, whereas detection is the procedure of determining the meaning of the waveform (i.e., decision making). The frequency downconverter translates the frequency to a lower frequency; then the receiving filter extracts the wanted frequencies and prepares the signal for detection. The filtering in the channel usually leads to intersymbol interference (ISI). This is why the equalizing filter is placed after the receiving filter. The sampling of the waveform is done before the actual detection. The pulse in the baseband is described as z(t ) = a i (t ) + n(t ) (6.4) 103
  • 113. 104 Modulation Techniques White noise Detection Demodulation Sampling si (t) r(t) z(T) Border Frequency Equalization comparison Filter down-converter filter ><γ z(t) Conversion of waveform into sample Decision making Figure 6.1 Demodulation and detection of digital signals. At the moment when t = T, the sample z(T) is taken. Its voltage is directly dependent on the energy of the received symbol and inversely proportional to the noise. If the input noise is Gaussian and the receiving filter is linear, the sample will be z(T ) = a i (T ) + n(T ), i = 1, 2, K (6.5) where ai(T) is the wanted part of the signal, and n(T) is the mean value of Gaussian noise. The next step will be detecting or decision making, depending on the digital meaning of the sample. The value of the random Gaussian noise n can be written as 1 ⎡ 1 ⎛n⎞ 2 ⎤ p(n ) = exp ⎢− ⎜ ⎟ ⎥ (6.6) σ 2π ⎣ 2 ⎝ σ⎠ ⎦ 2 where σ is the noise change. The two above expressions combined give the possibil- ities of waveforms for s1 and s2 as 1 ⎡ 1 ⎛z − a ⎞ 2 ⎤ p( z s1 ) = exp ⎢− ⎜ 1 ⎟ ⎥ (6.7) σ 2π ⎢ 2 ⎝ σ0 ⎠ ⎥ ⎣ ⎦ 1 ⎡ 1 ⎛z − a ⎞ 2 ⎤ p( z s 2 ) = exp ⎢− ⎜ 2 ⎟ ⎥ (6.8) σ 2π ⎢ 2 ⎝ σ0 ⎠ ⎥ ⎣ ⎦ The probability functions for s1 and s2 are shown in Figure 6.2. p(z/s 1 ) p(z/s 2 ) V2 V1 α1 γ α1 z(T ) Figure 6.2 Probability functions for p(z/s1) and p(z/s2).
  • 114. 6.2 Modulation and Demodulation 105 The curves represent the probability functions of z(T) for symbols s1 and s2. The abcissa axis z(T) represents all the possible sampled values. After the waveform is converted into the sample, the true shape of the waveform is no longer important. All of the waveforms that were converted into the same value of z(T) are equal as far as detection is concerned. Therefore, it is not the shape but the received energy that is the key parameter influencing the detection. Since the signal level z(T) depends on the energy of the bit received, the higher the value of z(T), the less errors in decision making.6.2 Modulation and Demodulation Modulation is a process of changing the electrical signal, which carries the informa- tion for its transmission. It changes one or more parameters of an auxiliary signal, depending on the signal that carries the information. This auxiliary signal is called the transmission signal or carrier. The signal that carries the information is called the modulating signal. It controls the changes of the transmission signal. The result of the modulation process is the modulated signal. Modulation is performed in an electronic device called the modulator, which is located in the transmitter. The reverse process, or demodulation, is the transformation of the received signal into the starting shape, which takes place in the demodulator at the receiving side. The modulation can be either analog or digital. Table 6.1 gives the most used basic types of modulation in communication systems. Analog modulations include amplitude modulation (AM), frequency modula- tion (FM), and phase modulation. Digital modulation includes amplitude shift key- ing (ASK), frequency shift keying (FSK), phase shift keying (PSK), and pulse code modulation (PCM). Quadrature amplitude modulation (QAM) can be both analog and digital. From the above mentioned basic types of modulation, several other combinations have been developed. They will be mentioned in Sections through and through 6.2.1 Analog Modulations In analog modulations, the message or information in analog form is superimposed on a carrier, which usually has a sinusoidal form. The three quantities that can be changed regarding the modulating signal are amplitude, frequency, and phase. Table 6.1 Modulations Analog Digital Amplitude modulation (AM) Amplitude shift keying (ASK) Frequency modulation (FM) Frequency shift keying (FSK) Phase modulation (PM) Phase shift keying (PSK) Pulse code modulation (PCM) Quadrature amplitude modulation (QAM)
  • 115. 106 Modulation Techniques Amplitude Modulation (AM) Amplitude modulation is the oldest modulation method. It works by changing the strength of the transmitted signal depending on the information being sent. The function of the amplitude of the AM modulated signal is linear and depends on the modulating signal. It can be written as [ ] u AM (t ) = U cm + u m (t ) cos( ω c t + ϕ) (6.9) where Ucm is the amplitude of the nonmodulated carrier, um is the modulating sig- nal, ωc is the frequency of the sinusoidal carrier signal, and ϕ is the phase of the car- rier signal. The principle of AM modulation is shown in Figure 6.3. Since the phase, ϕ, of the carrier does not influence the modulation process, it will be assumed that ϕ = 0 in further analysis. If the modulation signal is also sinusoidal it can be written as u m (t ) = U m cos ω m t (6.10) The modulated signal can then be written as uc Ucm 0 t um 0 t u AM Ucm 0 t Figure 6.3 Waveforms of carrier, modulating, and modulated signals of AM modulation.
  • 116. 6.2 Modulation and Demodulation 107 u AM (t ) = [U cm + U m cos ω m t ] cos ω c t (6.11) or as ⎡ U ⎤ u AM (t ) = U cm ⎢1 + m cos ω m t ⎥ cos ω c t (6.12) ⎣ U cm ⎦ The amplitude of the modulated signal changes around the mean value of Ucm. The maximum amplitude of the modulated signal is Ucm + Um. The minimum ampli- tude is Ucm − Um, accordingly. The ratio of the modulating signal and nonmodulated carrier signal is called the modulation index ma. Um ma = (6.13) U cm The modulation index is sometimes also called modulation depth. The expres- sion for the modulated signal can then be written as u AM (t ) = U cm [1 + m a cos ω m t ] cos ω c t (6.14) The above expression using the cosine product yields ⎡ m m ⎤ u AM (t ) = U cm ⎢cos ω m t + a cos( ω c + ω m )t + a cos( ω c − ω m )t ⎥ (6.15) ⎣ 2 2 ⎦ The modulation index should be ma ≤ 1. For ma > 1, correct demodulation is not possible. To determine the modulation index, a modulation trapezoid can be used. The modulation signal is on the horizontal axis and the modulated signal on the ver- tical axis, as shown in Figure 6.4. The modulation index is calculated as A−B ma = (6.16) A+B Demodulation of AM signals can be done with envelope detection or with syn- chronous detection. u AM u AM u AM A B 0 um 0 um 0 um ma <1 ma =1 ma >1 Figure 6.4 Modulation index depending on the trapezoid.
  • 117. 108 Modulation Techniques Envelope detection is the simplest procedure of AM signal demodulation. The envelope amplitude of the modulated signal with modulation index ma ≤ 1 is pro- portional to the modulation signal. Normally, a peak-amplitude detector or recti- fier is used. Figure 6.5 shows the AM demodulation procedure. The diode conducts when the input voltage is higher than the diode cut-in volt- age, which can range from 0.2V to 0.7V. The capacitor is used for filtering the demodulated signal. It also increases the efficiency of the demodulator by increasing the peak value of the carrier pulses while the diode is conducting. When the diode is not conducting, the capacitor is holding its charge. Additionally, an amplifier might be added to the demodulator. Demodulation with synchronous or coherent detection requires an additional signal whose frequency and phase match the carrier frequency. They also must be in phase, thus the name coherent. This type of detection is shown in Figure 6.6. The additional signal ua(t) (whose frequency matches the carrier frequency) will have the following form: u a (t ) = U a cos ω c t (6.17) Mixing this additional signal with the modulated signal gives u AM (t ) ⋅ u a (t ) = kAM ⎡ m m ⎤ ⋅U cm ⎢cos ω c t + a cos( ω c + ω m )t + a cos( ω c − ω m )t ⎥ (6.18) ⎣ 2 2 ⎦ ⋅ cos ω c t Further, it follows that Diode AM Demodulated signal signal R C Figure 6.5 AM demodulation with envelope detection. u AM (t) [u AM (t)xua (t)] [u AM (t)xua (t)]LPF LPF u a (t) Figure 6.6 Synchronous detection.
  • 118. 6.2 Modulation and Demodulation 109 ⎧1 + m a cos ω m t + ⎫ 1 ⎪ ⎪ u AM (t ) ⋅ u a (t ) = kAM ⋅ U cm ⎨ m ⎡cos(2 ω c + ω m )t +⎤⎬ (6.19) 2 ⎪ cos(2 ω c )t + a ⎢ ⎥ ⎩ 2 ⎣cos(2ω c − ω m )t ⎦⎪⎭ With the use of a lowpass filter, the components around 2ωc are filtered out: 1 [u (t ) ⋅ u (t )] AM a LPF = 2 kAM ⋅ U cm (1 + m a cos ω m t ) (6.20) Frequency Modulation (FM) With frequency modulation, the frequency of the carrier is changed, unlike analog modulation where the amplitude is changed. The most widely known use of FM is in radio station broadcasting. The modulation of frequency happens when the frequency of the carrier is changed according to the modulation signal. The frequency of the modulated signal will be ωFM (t ) = ω c + kf ⋅ u m (t ) (6.21) If the modulating signal has the cosine form: u m (t ) = U m cos ω m t (6.22) then the modulated frequency can be written as ωFM (t ) = ω c + kf ⋅ U m cos ω m t (6.23) Factor kf determines the largest frequency change at a certain amplitude of the modulating signal. The largest frequency deviation of the carrier frequency is ∆ωFM kf ∆fFM = = Um (6.24) 2π 2π The carrier frequency of the modulated signal will be f (t )FM = f c + ∆fFM cos ω m t (6.25) The waveform of frequency modulation is shown in Figure 6.7. The waveform of the FM signal is determined with ⎡ ∆ωFM ⎤ uFM (t )U cm cos ⎢ω c t + ⎣ ωm ⎦ [ sin ω m t ⎥ = U cm cos ω c t + m f sin ω m t ] (6.26) where the modulation index mf is calculated from ∆ωFM ∆f kf U m kf U m mf = = FM = = (6.27) ωm fm ωm 2 πf m
  • 119. 110 Modulation Techniques uc Ucm 0 t um 0 t u FM Ucm 0 t Figure 6.7 Frequency modulation waveforms. The modulation index is therefore the ratio of the frequency deviation and mod- ulating frequency. It can be higher or lower than 1. The modulation index depends on both the frequency and amplitude of the modulating signal. With AM, only the amplitude determines the modulation index. Demodulation of the FM signal is performed with slope detection as shown in Figure 6.8. The slope detector consists of an FM to AM converter and an AM envelope detector (described in Section The first part of the detector is actually a lowpass RC filter whose cutoff frequency is determined from 1 fc = (6.28) 2πRC and is chosen to be the carrier frequency of the FM signal. The signal is next processed with the envelope detection circuit, as if it were an AM modulated signal.
  • 120. 6.2 Modulation and Demodulation 111 AM FM Demodulated envelope signal signal R detector C Figure 6.8 FM demodulation with slope detection. Phase Modulation (PM) Phase modulation, along with frequency modulation, is a form of angle modula- tion. Here, the phase of a carrier signal is changed depending on the modulating sig- nal. The phase modulation is closely related to the frequency modulation because the frequency cannot be changed without varying the phase. Thus, with phase mod- ulation there is always parasitic frequency modulation and vice versa. The change of the carrier phase is determined with the following expression: ϕFM (t ) = ϕ 0 + kp ⋅ u m (t ) = ϕ 0 + ∆ϕ(t ) (6.29) Since the relative phase of the carrier signal, ϕ0, does not influence the modu- lated signal, it can be 0. The angle of sinusoidal function is called the phase of mod- ulated signal and is: Φ PM (t ) = ω c t + kp ⋅ u m (t ) (6.30) The phase modulated signal can be described as [ u PM (t ) = U m cos ω c t + kp u m (t ) ] (6.31) and if the modulating signal has a sinusoidal waveform, u m (t ) = U m sin ω m t (6.32) the phase of the modulated signal becomes Φ PM (t ) = ω c t + kp ⋅ U m sin ω m (t ) = ω c t + ∆Φ PM sin ω m (t ) (6.33) where factor kp determines the largest phase change at some amplitude of the modu- lating signal. The largest phase shift of the modulated signal is called the phase devi- ation or ∆ΦPM. This phase deviation is also the modulation index, or m p = ∆Φ PM = kpU m (6.34) The same as with frequency modulation, mp can be higher or lower than 1. Figure 6.9 shows the waveforms of the carrier signal, modulating signal, and modulated signal for phase modulation.
  • 121. 112 Modulation Techniques Figure 6.9 Phase modulation waveforms. As can be seen, the phase deviation changes with the angle of the modulating signal. It can be positive or negative. Demodulation of PM is performed using frequency coherent demodulation (i.e., using a reference signal with a fixed phase reference). The circuit used is the phase detector, whose principle is beyond the scope of this book. PM has a better demodulated S/N ratio than FM, but since coherent demodula- tion was not as simple as envelope detection in the past, FM was spread more than PM, especially in broadcasting. Although today this is not a problem anymore, FM still stays more in use than PM. 6.2.2 Digital Modulation The difference between analog and digital modulations is in the modulating signal, which is analog in analog modulations and digital in digital modulations. The digi- tal modulating signal will be either 0 or 1. The three main types of modulations are amplitude shift keying (ASK), frequency shift keying (FSK), and phase shift keying (PSK). In this chapter, pulse code modulation and quadrature amplitude modulation will also be covered.
  • 122. 6.2 Modulation and Demodulation 113 Amplitude Shift Keying (ASK) Amplitude shift keying (ASK) is a modulation method where the amplitude of the analog signal carrier, usually sinusoidal, is changed according to the digital modu- lating signal. This type of modulation is also called on-off keying (OOK), where the signal exists when the digital modulating signal is equal to 1 and there is no signal when the digital modulating signal is equal to 0. ASK is used in optical communica- tions. The principle of ASK modulation is shown in Figure 6.10. The modulated signal is obtained by modulating the carrier signal having fre- quency fc with the modulating signal having frequency fm: ⎡1 2 2 ⎤ u ASK (t ) = U cm cos ω c t ⋅ + cos ω m t − cos 3ω m t + K (6.35) ⎢2 π ⎣ 3π ⎥ ⎦ The ideal ASK signal has an infinite spectrum. It is therefore necessary to shape the modulating pulses. The disadvantage of the ASK modulation is that the ratio S/N does not apply for 1 and 0 of the modulating signal. There is also a large differ- uc Ucm 0 t um 1 0 t u ASK Ucm 0 t Figure 6.10 ASK waveforms.
  • 123. 114 Modulation Techniques ence in the power consumption between these two states. Another disadvantage is that the loss of link is read as 0. Since the ASK modulated signal has a very definitive envelope, an envelope detector can be used for the first step in the demodulation process. Further process- ing is usually required because the shaping of modulating pulses has to be done before transmission in order to limit the frequency bandwidth. Frequency Shift Keying (FSK) Frequency shift keying is a modulation method where the frequency of the analog signal carrier, usually sinusoidal, is changed discretely according to the digital mod- ulating signal. The simplest FSK modulation is BFSK or binary FSK modulation with two carrier frequencies. Another derivative of FSK is minimum shift keying (MSK) where the modulation index is smallest or 0.5. One type of MSK is called the Gaussian MSK, which is used in mobile telephone standards. The principle of FSK modulation is shown in Figure 6.11. Usually a higher level of the modulating signal is associated with a higher frequency. uc Ucm 0 t um 1 0 t u FSK Ucm 0 t Figure 6.11 FSK waveforms.
  • 124. 6.2 Modulation and Demodulation 115 The modulation index is equal to the ratio of frequency deviation ∆f and modu- lation frequency fm: ∆f mF = (6.36) fm where frequency deviation is defined as f1 − f 0 ∆f = (6.37) 2 and current frequencies of FSK signal being f0 = fc − ∆f and f1 = fc + ∆f , with fc being the carrier frequency. FSK modulation is a nonlinear procedure whose frequency spectrum has many components. It can be obtained with two ASK signals, with carrier frequencies f1 and f2. The ASK modulation is a linear procedure with a much simpler frequency spectrum. If FSK is obtained with two ASK signals, envelope detection can be used for demodulation. In other cases, synchronous or asynchronous demodulation is necessary. Synchronous demodulation of FSK signals is shown in Figure 6.12. As mentioned before, the demodulator requires two local oscillators which gen- erate the carrier frequency and must be synchronized. There are two low pass filters, tuned to f1 and f2. At the end, a decision is made as to which of the two signals is the right one. This type of demodulator actually has two receiving channels. Asyn- chronous demodulation of the FSK signal is shown in Figure 6.13. Asynchronous demodulation uses the advantage of the fact that the FSK modu- lation can be achieved with two ASK signals. The FSK signal is separated with band pass filters 1 and 2. The filter outputs can be demodulated like ASK signals with envelope detectors. At the end, the decision is made as to which of the two signals is correct. FSK signal LFP Demodulated Decision signal f1 FSK signal LFP f2 Figure 6.12 Synchronous demodulation of FSK signals.
  • 125. 116 Modulation Techniques FSK signal Envelope BPF 1 detector Demodulated Decision signal FSK signal Envelope BPF 2 detector Figure 6.13 Asynchronous demodulation of FSK signals. Phase Shift Keying (PSK) Phase shift keying is a modulation method where the phase of the analog signal car- rier, usually sinusoidal, is changed according to the digital modulating signal. With PSK the relative phase of the modulated signal can have two or more different phases from the previously defined set of phases. The waveforms of PSK are shown in Figure 6.14. uc Ucm 0 t um 1 0 t u PSK Ucm 0 t Figure 6.14 PSK waveforms.
  • 126. 6.2 Modulation and Demodulation 117 The PSK modulation can also be obtained from two ASK signals, just like the FSK modulation. The difference is that with PSK these two signals must be in a quadrature relation. The PSK signal will be u PSK (t ) = U cm cos( ω c t + ϕ m ) = U cm [cos ϕ m ⋅ cos ω c t − sin ϕ m ⋅ sin ω c t ] (6.38) where ϕm is the modulating phase obtained from π(2n + c ) ϕm = , n = 0, 1, 2, K, M − 1; c = 0, 1 (6.39) M For M = 2, and c = 0, the modulating phases can be ϕm = 0, π, and for c = 1, another set of modulating phases will be ϕm = π/2, (3π)/2. This modulation is called BPSK or binary phase shift keying. Usually phase 0π or 0° is dedicated to state 1 and π or 180° to binary state 0. Their relation is shown in Figure 6.15. BPSK modulation is very resilient to interference, but its spectral efficiency is not very high. It is possible to use a set of phases much higher than M = 2. For M = 4, there will be four different phases. This type of PSK modulation is called QPSK. The possible set of phases will be for c = 0, ϕm = 0, π/2, π, 3π/2, and for c = 1, ϕm = ±π/4, ±, 3π/4. With QPSK, there will be two bits necessary for each state instead of one bit for BPSK. This requires more memory and increases the possibility of error com- pared to BPSK. The spectral efficiency of QPSK is increased compared to BPSK, which means that the amount of information that can be carried in a communica- tion channel is doubled. QPSK phases are shown in Figure 6.16. Each state is assigned binary digits according to the Gray code where the neighboring states dif- fer only by one digit. The QPSK signal can be described with uQPSK (t ) = I(t ) cos ω c t − Q(t ) sin ω c t (6.40) The QPSK signal can be achieved by combining two BPSK signals. There are other types of PSK modulations with a higher number of phases (8, 16, ...). However, by increasing the number of phases, the possibility of error also increases, since the phases are coming closer to each other. This can be compensated Q “0” “1” I 180° 0° Figure 6.15 BPSK phases in I-Q plane.
  • 127. 118 Modulation Techniques Q 01 11 I 00 10 Figure 6.16 QPSK phases in I-Q plane. for by increasing the power; however, in some cases this is not possible because of international power levels according to standard regulations. PSK modulated signals cannot be demodulated using envelope detection. Demodulation is possible only with synchronous demodulation, which means that the receiver will have to include a referent signal, which will provide the signal on the carrier frequency. The quadrature demodulator shown in Figure 6.17 is used for demodulating PSK signals (i.e., for obtaining I and Q components). Pulse Code Modulation (PCM) Pulse code modulation is a process where the amplitude of an analog signal is sam- pled and quantized into a binary code. The principle is given in Figure 6.18. PCM requires analog-to-digital conversion (ADC). The sampling rate of an analog signal must be at least twice the frequency of an analog signal, that is f s ≥ 2f c (6.41) FSK signal LFP Demodulated Decision signal f1 FSK signal LFP f2 Figure 6.17 Quadrature demodulator.
  • 128. 6.2 Modulation and Demodulation 119 uc 0 t um 1 0 t u PCM 7 111 6 110 5 101 4 100 3 011 2 010 1 001 0 000 Figure 6.18 PCM modulation. The level of quantization will determine how closely the analog signal is sam- pled. More levels will result in a closer resemblance of the sampled and quantized signal to the analog signal, but will require more processor memory. The demodulator works in a similar way. It takes the numbers (bits) and assigns them the analog voltage accordingly. The receiver will require an analog-to-digital conversion (DAC) circuit to perform this operation. Quadrature Amplitude Modulation (QAM) The quadrature amplitude modulation is a process where the amplitude of two car- riers is changed. The two carriers are usually sinusoidal and have a phase difference of 90°. QAM can be either analog or digital. The analog QAM works similar to AM. The difference is that QAM uses two carriers instead of one, which both have the same frequency, but their phase differ- ence is 90°. They are modulated with two different modulating signals and then combined before transmission. The QAM signal has the following form: uQAM − u m1 (t ) cos ω c t + u m 2 (t ) sin ω c t (6.42) Modulation changes the amplitude of carrier signals, but not the phase differ- ence between them. Analog QAM is demodulated using synchronous detection. Digital QAM is used more than analog QAM, so it is enough to call it just QAM. Digital QAM has two modulating signals, I(t) and Q(t), whose relation is as follows uQAM I(t ) cos ω c t − Q(t ) sin ω c t (6.43)
  • 129. 120 Modulation Techniques Q Q I I 8-QAM 8-PSK Figure 6.19 8-QAM and 8-PSK modulations. PSK is actually a special case of QAM. The difference is that QPSK has no amplitude modulation while QAM has. QPSK has a constant amplitude. Depending on the quantization level, QAM can be 4 QAM, 16 QAM, 64 QAM, and so forth. Figure 6.19 shows the 8-QAM and 8-PSK states. The more states there are, the higher the possibility of error.6.3 Control of System Drift If there is a difference between the carrier frequency in the transmitter and the car- rier frequency in the receiver (which is necessary for the synchronous demodulator), a frequency offset will occur. ∆f = f cr − f ct (6.44) International standards require that the frequency offset be kept under a certain level. For example, for the HIPERLAN/2 transmitter carrier frequency fc, the ratio ∆f/fc must be less than 0.002%. If the same demand were required for the receiver, ∆f/fc would need to be 250 kHz. This means that the majority of the power from the transmitted subcarrier will be received in the neighboring channel, which will lead to a large bit error rate (BER). The frequency offset must be measured in the receiver system and corrected. This can be done with a voltage controlled oscillator and phase-locked loop.Selected Bibliography ETSI EN 300 910 V.8.5.1. (2000-11), Digital Cellular Telecommunications System, (Phase 2+); Radio transmission and reception (GSM 05.05 version 8.5.1 Release 1999) Feher, K., Wireless Digital Communications, Upper Saddle River, NJ: Prentice-Hall, 1995. Modlic, B., and I. Modlic, Modulacije i modulatori, Skolska knjiga Zagreb, 1995. Sklar, B., Digital Communications—Fundamentals and Applications, 2nd ed., Upper Saddle River, NJ: Prentice-Hall, 2001.
  • 130. 6.3 Control of System Drift 121 Van Hoesel, L. F. W., et al., “Frequency Offset Correction in a Software Defined Hiperlan/2 Demodulator Using Preamble Section A,” Proceedings of the Third International Symposium on Mobile Multimedia Systems & Applications, Delft, the Netherlands, December 6, 2002, pp.51–62. Xiong, F., Digital Modulation Techniques, Norwood, MA: Artech House, 2000.
  • 131. CHAPTER 7Electromagnetic Field Coupling to Wire7.1 Field-to-Wire Coupling Every wire that acts as a conductor carries charge. When a wire is placed in the elec- tromagnetic field, it will be under its influence. This influence will depend on the type of wire, the intensity of the field, and the quality of the shield. If a conductor is used for carrying information, a foreign electromagnetic field may cause errors in the receiving system. Any cable can also act as an antenna (i.e., it can radiate or receive unwanted interfering signals). The flow of the current largely depends on the frequency. As the frequency increases, the current tends to flow more closely to the surface due to the skin effect. The resistance R of the wire depends on the length, l, cross-section, S, and resistivity of the material, ρ: 1 R= ρ (7.1) S The resistivity of some materials is given in Table 7.1. 7.1.1 Skin Effect At high frequencies the resistance is not the same as it is on DC current. The surface resistivity of the wire dependant on the frequency can be calculated from 1 ρ Rs = πfµρ = = (7.2) δσ δ where f is the frequency, µ is the permeability of the wire, and ρ is the resistivity 1 ( ρ = , with σ being the conductivity and δ being the skin depth). The higher the fre- σ quency, the smaller the skin depth—and the resistivity increases. This is shown in Figure 7.1. The current flows only in the dashed area. At DC, the wire cross-section is used for the current flow. At higher frequencies, the current tends to stay closer to the surface. At RF frequencies, the current is concentrated only at the surface and the penetration depth is very small. This characteristic is used for building shields from electric fields. The AC resistance can be written as 123
  • 132. 124 Electromagnetic Field Coupling to Wire Table 7.1 Resistivity of Different Materials at 20° Conductor Resistivity (n m) Aluminum 28.2 Copper 17.2 Gold 24.4 Iron 97.1 Platinum 106 Lead 220 Silver 15.9 Zinc 58 Semiconductor Resistivity (m m) Carbon 0.015 Germanium 460 Silicon 250,000 Insulator Resistivity (T m) Glass 0.01–100 Rubber 10–10,000 Wood 0.01 δ r r´ DC HF Figure 7.1 Skin effect. 1 R= ρ (7.3) S ef where the effective area, Sef, is obtained by S ef = πr 2 − πr ′ 2 = πr 2 − π( r − δ) 2 (7.4) Further analysis gives ( S ef = π r 2 − r 2 + 2 rδ − δ 2 ) = π(2rδ − δ ) 2 (7.5) The resistance of a conductor at radio frequencies will then be 1 R= ρ (7.6) [π(2rδ − δ )] 2
  • 133. 7.1 Field-to-Wire Coupling 125 The resistance of a copper wire (1m long, 1.5 mm in diameter) at different fre- quencies is shown in Figure 7.2. It can be seen that the resistance rises quite fast with the frequency. At 200 MHz all of the current will flow in about 5 outer micrometers of the cable. As frequencies increase, cables have difficulty carrying the signals properly. Additionally, it is difficult to prevent them from leaking. Regarding EMC problems, nonmetallic conductors—which include wireless, fiber-optic, microwave, or laser links—are better for carrying signals. Coupling paths between transmitters and receivers can be either radiated or conductive and include antenna-to-antenna coupling, cable-to-cable coupling, antenna-to-wire (cable) coupling, or cable-to-antenna coupling as shown in Figure 7.3. Crosstalk or cable-to-cable coupling depends on frequency and bandwidth. 7.1.2 Unshielded Twisted Pair (UTP) When current flows through the wire, electromagnetic radiation is inevitable. When two cables (forward and return) are needed, it is possible to twist them into a 0.9 0.8 0.7 Resistance (ohms) 0.6 0.5 0.4 0.3 0.2 0.1 0 0 1 2 5 10 20 50 100 200 f (MHz) Figure 7.2 Resistance of 1-m-long copper wire versus frequency. Radiated T coupling R Transmitter Receiver Signal line equipment equipment Conducted coupling Figure 7.3 Coupling paths.
  • 134. 126 Electromagnetic Field Coupling to Wire twisted pair for the cancellation of electromagnetic interference. This old technique is called UTP or unshielded twisted pair. Here, the signals flowing in a pair have opposite directions and the fields tend to cancel each other out. This solution can sometimes effectively cancel crosstalk between neighboring pairs. This method has been used for telephone lines for many years now. An unshielded twisted pair usually comes in two colors (Figure 7.4). It is used for the Internet, telephone cables, and video. There are usually between 4 to 25 pairs inside a sheath. While the UTP cable has no shield, there are other designs of wires with shielding, which will be covered in Section 7.4.3. Whether the conductors are used inside or outside a certain electronic device will determine the type of wire used. If used inside a product with a good shield, the choice of wire is not so important in regards to interference, although signal perfor- mance might be of interest. It would be best not to have internal cables at all, and instead have PCB traces. This simplifies the shielding and reduces its cost. If conductors are used outside of an electronic device, a shield of any sort is desirable regardless of whether the communication is analog or digital. 7.1.3 Ferrite Filter Filtering the signals with ferrite filters (Figure 7.5) can sometimes help protect against interference. A ferrite filter can, if properly used, suppress interference. It is possible to use an in-line filter or onboard suppression circuits, but they are usually more expensive solutions. Ferrites have a concentrated homogenous magnetic structure with high permeability. Their characteristics do not change with time and temperature, and their application depends on the frequency of use. Prior to the usage of a ferrite filter, the cable impedance has to be known. It is usually 50Ω, but it can also vary from a few ohms to several hundreds of ohms. Ferrite impedance depends on dimensions (length, outer and inner diameter). The most important thing to have in mind is that the ferrite diameter should be as close to the wire diameter as possible. Isolation 4 pairs UTP Conductor Sheath Figure 7.4 Unshielded twisted pair (UTP).
  • 135. 7.1 Field-to-Wire Coupling 127 Connector Ferrite filter Cable Figure 7.5 Ferrite filter. The insertion loss (a measure of the filter effectiveness at a certain frequency), which is usually described as the ratio of voltage with and without a filter, can be calculated from ZC + ZF Insertion loss = 20 log10 (7.7) ZC where ZC is the circuit impedance and ZF is the ferrite impedance. If, for instance, circuit impedance is 50Ω and ferrite impedance at 40 MHz is 100Ω, the insertion loss will be 9.5 dB. If circuit impedance is 75Ω instead of 50Ω, the insertion loss will be 7.4 dB. This means that the insertion loss will be higher when the circuit imped- ance is lower. The effectiveness of the ferrite can be enhanced if cable or wire is passed through the ferrite more than once. The ferrite impedance, ZF, increases geometri- cally with the number of loops (Figure 7.6). For two loops, the ferrite impedance is four times bigger. The disadvantage of multiple loops is that the frequency band of ferrite becomes narrower. If, for the previous example, there are two loops instead of one, ferrite Figure 7.6 Multiloop ferrite.
  • 136. 128 Electromagnetic Field Coupling to Wire impedance ZF will be 400Ω at 40 MHz. The insertion loss in this case will be 19.1 dB and for three loops it will be 25.6 dB. Ferrite placement is also important. It is best to place the ferrite close to the end of cable where it leaves the electronic equipment. If the cable connects two pieces of the electronic equipment, two ferrites might be necessary. Large ferrites generally have higher impedance, but since their size increases the total weight and space, this has to be kept in mind when choosing the ferrite. Filtering only attenuates the interfering signals—it does not remove them. For some cases, cable shielding is necessary. The coupling to wires can be either capacitive (electric field) or inductive (mag- netic field).7.2 Electric Field Coupling to Wires An electric field can be coupled to wire by stray capacitance. Figure 7.7 shows how an electric field is coupled into a wire (circuit) carrying the useful signal. This elec- tric field can come from several sources such as another parallel cable or circuit, or from the electric field of an antenna. An interfering electric field is coupled through stray capacitance in an equiva- lent circuit, as shown at the bottom of Figure 7.7. The signal source impedance, RS, and load impedance, RL, are the same. The useful signal is characterized with VS, and the source of noise, VN, is coupled via stray capacitance, CN. Stray capacitance occurs when two conductors are close to each other and there is no shield or grounding present. It usually occurs between parallel traces on a PC board. Bad planning can lead to lower stability, greater noise, and reduced fre- quency response. The stray capacitance is proportional to the area of interlap, S, and inversely proportional to the distance between two circuits, d, as: S C = ε0 εr (7.8) d Increasing the distance or minimizing the overlap will minimize the capacitance and thus the coupling of noise into the signal. The coupling also depends on noise level, frequency, and load impedance. Coupled voltage, VC, will be equal to RL VC = VN (7.9) 1 RL + ωC N Capacitive coupling will be smaller for lower values of noise voltage, frequency, and circuit impedance. The voltage induced on a wire from the electric field using Faraday’s law is: r r d r r Vi = ∫ E ⋅ dl C =− dt ∫ ∫ B ⋅ ds S (7.10)
  • 137. 7.2 Electric Field Coupling to Wires 129 E-field VS h RL RS l CN VN VS RL RS Figure 7.7 Capacitive coupling. r r where E is the electric field intensity vector and B is the magnetic flux density vector. The above expression can be simplified to ⎛ βl ⎞ Vi = 2 Eh sin ⎜ ⎟ (7.11) ⎝2⎠ where h and l from Figure 7.7 define the loop area. If the phase constant β = 2π/λ is introduced in the above expression, the equation for induced voltage is obtained: ⎛ πl ⎞ Vi = 2 Eh sin ⎜ ⎟ (7.12) ⎝ λ⎠ For low frequencies, where wire length l is short relative to the wavelength (l < λ/2), sinx = x, in which case the above expression becomes Vi = 2πlh E λ (7.13) Above a frequency at which l = λ/2, (7.13) can be simplified: Vi = πhE (7.14)
  • 138. 130 Electromagnetic Field Coupling to Wire This is a high-frequency asymptote, overinduced voltage at all frequencies above the one where l = λ/2. If, for example, we take a case with l = 1m and h = 0.5 cm, the normalized elec- tric field-to-wire coupling, Vi /E will depend on the frequency as shown in Figure 7.8. The dashed line shows the approximation for low and high frequencies dis- cussed above. The maximums of the coupling will appear when (2n + 1)c f = , n = 0, 1, 2, K (7.15) 2l and the minimums will appear when nc f = , n = 0, 1, 2, K (7.16) l where c is the speed of light. Electric fields can be coupled using common mode coupling or differential mode coupling. With common mode coupling the currents in cables run in the same direction, while with differential mode coupling the currents run in opposite direc- tions. Both examples are shown in Figure 7.9. 0 −10 −20 Vi /E −30 −40 −50 −60 1 10 100 1000 f (MHz) Figure 7.8 Electromagnetic field-to-wire coupling versus frequency. ic id ic id Figure 7.9 Common mode coupling and differential mode coupling.
  • 139. 7.3 Magnetic Field Coupling to Wires 1317.3 Magnetic Field Coupling to Wires If current is flowing through a wire it will create a magnetic field around the wire. The magnetic field creates an electric field perpendicular to the magnetic field. This electric field can cause a current flow in the wire in its vicinity. This characteristic of mutual inductance is the basis of how transformers actually operate. Magnetic coupling of interference is unwanted inductive coupling from one loop to another. Typical noise sources are motors, transformers, relays, and so forth. Inductive coupling is the result of a magnetic field in the area enclosed by the signal circuit loop. The magnetic field is generated from the current flowing in the adjacent noise circuit as shown in Figure 7.10. The induced voltage VN in the signal circuit is calculated from: VN = 2 fBS cos ϕ (7.17) IN VN R VS RL RS IN VN R M VS RL RS Figure 7.10 Inductive coupling.
  • 140. 132 Electromagnetic Field Coupling to Wire where f is the frequency of the noise signal, B is the magnetic flux density, S is the area of the signal circuit loop, and ϕ is the angle between the flux density, B, and the area, S. The induced voltage according to the equivalent circuit shown at the bottom of Figure 7.10 is VN = 2 fMI N (7.18) where IN is the current value in the noise circuit. Mutual inductance, M, is proportional to the area of the receiver circuit loop, frequency of the source, and current level of the source, and is inversely propor- tional to the distance between the signal and loop circuit. Thus, the coupling can be minimized by separating these two circuits. The coupling can also be minimized by lowering the frequency and current of the noise circuit or twisting the wires of the noise source. Twisting the wires reduces the circuit loop area, S. Finally, magnetic shielding of both the noise circuit as well as the signal circuit can further lower the magnetic (inductive) coupling. The ratio of noise voltage to signal voltage is lowest when the circuit impedance is highest. This condition is exactly the opposite for the case of minimum electric field coupling for the lowest circuit impedance. There is an optimum circuit imped- ance where the overall coupling will be the smallest.7.4 Cable Shielding Cable shielding is a procedure of protecting the wires (conductors), which are carry- ing information or are used as power lines, with an outer protective layer. The shield protects the wire from the electric field coming from the noise source and at the same time protects the surrounding electronic equipment or other cables from the wire’s own electromagnetic radiation. In Section 7.1.2 the twisted pair was mentioned as a method for protection of the wire from unwanted electromagnetic coupling. In some cases this is not suffi- cient. An additional metal shield provides better noise suppression. The parasite, or stray capacitance, CN, can be reduced by applying the capaci- tive shielding. The coaxial cable is a good example of this method. The idea is to provide another path for the induced current rather than the wire carrying the sig- nal. The shield is placed between the capacitively coupled conductors and con- nected to the ground only at the source end. If the shield is connected to the ground at both ends, a large ground current may exist, so this is generally not recommended. If a cable shield is thick (several skin depths), and has no holes in it, it will have a shielding effectiveness (SE) above 200 dB. Shielding effectiveness will be discussed in depth in Chapter 8. The conducted currents flowing on the surface of the shield will not penetrate the shield and interfere with the wires inside, which carry useful signals. This type of shield does not require grounding at all. If, on the other hand, the shield is thin (less than skin deep) and leaky (i.e., with holes), the absorption loss will be small. There will only be the reflection loss of the shield; the holes might further lessen the reflection loss. The only protection this
  • 141. 7.4 Cable Shielding 133 shield can give to the cable would occur if it were grounded. In this case it must be connected to the closest metal reference to form a Faraday cage. If the interfering noise field is uniform, it does not matter which end of the shield is grounded. In other cases, it is better to ground the end closer to the interfering noise source. Grounding the shield at both ends generally is not a good idea because this will cre- ate a new ground loop, resulting in an even greater surface current. Since the shield has holes, the greater surface current would induce high voltages in internal wires. Thus, if a bad cable shield is used, grounding only one end is desirable. 7.4.1 Tri-Axial Cable Shielding cables differentiates shielding and the screening. Screening is actually shielding more wire pairs and not just one pair. Usually more screening and shield- ing will raise the cable cost. The cable screen should cover the entire length of cable with 360° coverage. The screen (shield) carries both the return signal and external interference on opposite sides of the cable (due to the skin-depth effect), which is possible for solid copper screens. Flexible screened cables cannot keep the two currents separated, so the return currents leak out and the interfering currents leak in. The solution for this problem is a tri-axial cable shown in Figure 7.11. A tri-axial cable is quite similar to the coaxial cable, except with the addition of one more shield. One center conductor is surrounded by two shield layers insulated from each other. Usually the outer shield is grounded and the inner shield is used for the return signal. The shielding properties of such a cable are better than those for a simple coaxial cable. The outer shield lowers the ground loop interference and elim- inates the radiated noise or crosstalk. 7.4.2 Cable Termination The shield, as mentioned before, must be terminated (or matched). Cable screens should always be connected to their enclosure shield, and should be terminated in 360° to the skin of the screened enclosure they are going into. The choice of the con- nector at the end of the cable is essential. The most common connection is the pigtail connection. It is the connection where the screen is brought down to a single wire and extended through a connector pin to the ground point. It is easy to assemble. At high frequencies this connection Foil Copper shield shield Copper conductor Dielectric insulator Plastic coating Figure 7.11 Tri-axial cable.
  • 142. 134 Electromagnetic Field Coupling to Wire becomes inappropriate because of its inductance, which is serially connected to the cable screen. The inductance will introduce a voltage when interference currents flow along the screen to the ground. This voltage can be coupled from the screen into the inner conductors. The same goes for the emission of radiation from the inner conductors to the ambient area. The impedance of such a connection increases with the frequency and may completely degrade the use of a good cable screen. The pigtail connection should be avoided at frequencies above 10 MHz if possi- ble, or kept very short. A much better solution is crimping and employing backshell in combination with stress relief bolts as shown in Figure 7.12. Connector backshells can be potted and molded. Even the best cable shield will be useless if the bonding to the connector is done poorly. Some sort of enclosure shielding at the end of the cable is always necessary, especially at higher frequencies. 7.4.3 Shielded Twisted Pair Cables The previously mentioned unshielded twisted pair (UTP), which is the most simple and cheapest cable available, is sometimes not appropriate because of its lack of shielding. There are other types of twisted pairs with screening and shielding. The shielding can be applied to individual pairs or several pairs in a cable. The first con- sidered is the screened unshielded twisted pair (S/UTP). The S/UTP cable is a screened UTP cable, which means that it has a single shield beneath the sheath for all of the twisted pairs together as shown in Figure 7.13. Since the shield is a metal foil, it is sometimes called the foiled twisted pair (FTP). The next type of shielded twisted pair is the shielded twisted pair (STP), which has metal shielding over each individual pair of wires instead of just one shield for all the twisted pairs together like S/UTP. It provides better protection from electro- magnetic interference and prevents crosstalk between the pairs as well. The cross-section of STP is shown in Figure 7.14. The last shielded cable considered is the screened shielded twisted pair (S/STP), also known as the screened fully shielded twisted pair (S/FTP) cable. This cable has shielding of the pairs as well as a shield beneath the sheath. It has the best protection from interference of all the cables mentioned in this section. The cross-section is shown in Figure 7.15. Backshell Connector Cable Stress relief bolts Figure 7.12 Bonding cable to connector.
  • 143. 7.4 Cable Shielding 135 Sheath Conductor Insulator Screen Pair Figure 7.13 Screened unshielded twisted pair (S/UTP) cable. Pair shield Sheath Conductor Insulator Figure 7.14 Shielded twisted pair (STP).
  • 144. 136 Electromagnetic Field Coupling to Wire Pair shield Sheath Conductor Insulator Screen Figure 7.15 Screened shielded twisted pair (S/STP).Selected Bibliography “Crimping, Interconnecting Cables, Harnesses, and Wiring,” NASA-STD-8739.4, February 1998, EMC and Compliance Yearbook 2003, CD-ROM, Nutwood UK Ltd, Eddystone Court, De Lank Lane, St Breward, Bodmin, Cornwall, PL30 4NQ, United Kingdom. Javor, K., “On Field-to-Wire Coupling Versus Conducted Injection Techniques,” Proceedings of IEEE International Symposium on Electromagnetic Compatibility, August 18–22, 1997, Austin, TX, pp. 479–487. Martin, L., and A. Kamiens, “Magnetic Shielding Theory and Practice,” ITEM 2001, pp. 1–3. May, J., “Filtering Out Interference Signals with Cable Ferrites,” Compliance Engineering Maga- zine, November 2002. Trout, D., “Investigation of the Bulk Current Injection Technique by Comparison to Induced Currents from Radiated Electromagnetic Fields,” 1996 IEEE EMC Symposium, Santa Clara, CA, 1996, pp. 412–417. Worshevsky, A., and R. Patlaty, “Low Frequency Common Mode Voltages in Electrical Cable Runs,” Proceedings of IEEE 6th International Symposium on Electromagnetic Compatibility and Electromagnetic Ecology, June 21–24, 2005, Saint Petersburg, Russia, pp. 224–225.
  • 145. CHAPTER 8Electromagnetic Field-to-ApertureCoupling8.1 Field-to-Aperture Coupling The aperture is a hole or an opening in an enclosure of an electronic device through which electromagnetic fields may enter or leak out. Since immunity of an electronic device is defined as its resistance to ambient electromagnetic fields, any holes or apertures in the shielding may compromise the operation of such a device. At the same time, other electronic devices in the vicinity of the above mentioned electronic device, which has holes or apertures in its shield, will be in danger from its emis- sions. The example of this is shown in Figure 8.1. Electronic device emissions are not a problem if they are kept below levels pre- scribed by international standards. Most of the emissions will stay inside the shield and diminish with multiple reflections. Some emissions may leak out from the shield and reach other electronic equipment in the vicinity. A shield aperture can be poten- tially dangerous to other electronic equipment as shown in Figure 8.1(a). Similarly, the aperture in the shield can leak in the electromagnetic radiation. The shield pro- vides good protection from most of the ambient radiation, but some might still pen- etrate the shield, as shown in Figure 8.1(b). Some of the radiation that enters the shield will be lessened through multiple reflections (and absorptions) from the shield, but not all. The best shield will be the one without any holes or apertures; no electromag- netic fields can enter the shield and interfere with the electronic equipment. Also, this kind of equipment will not produce any ambient fields, provided that the shield is designed in such a way as to protect from both the electric and magnetic fields. This type of shield would be beneficial even though it is impossible to achieve total shielding against electric or magnetic fields; however, such a shield is impractical. This is because there has to be some sort of power cord going in or out of the shield. Also, apertures for ventilation, sensors, antennas, and connectors to other equip- ment all must be taken into consideration. These are just some of the reasons why a perfect shield is impossible to achieve. Since the apertures in the shield cannot be avoided, there are still methods avail- able to keep the influences of these apertures on the shield effectiveness at a minimum. 137
  • 146. 138 Electromagnetic Field-to-Aperture Coupling Emission Immunity (a) (b) Figure 8.1 (a) Emission and (b) immunity depend on the shield apertures. 8.1.1 Shielding Effectiveness (SE) Apertures in a metal shield can be considered as half-wave resonant slot antennas. This means that the geometrical dimensions of the apertures will determine which frequencies can travel through the slot or aperture and which cannot. If there is only one aperture in the shield, the shielding effectiveness (SE) can be obtained from ⎛ λ⎞ SE = 20 log10 ⎜ ⎟ (8.1) ⎝ 2d ⎠ where λ is the wavelength and d is the maximum dimension of the aperture (diago- nal). Thus, for a desired SE, there is a maximum aperture, with the largest d allowed for a given frequency: 150 d ( mm) = (8.2) f ( MHz ) Figure 8.2 shows the maximum allowable aperture diameter d for SE = 60 dB. For 1 MHz this value is 150 mm, and it becomes smaller, reaching only 0.15 mm for 1 GHz. 8.1.2 Multiple Apertures It is much better to have several smaller apertures than one large one, if possible. Induced currents (from the magnetic field) in the shield will flow as long as there is no obstruction in their path. Currents flowing in a shield coming to an aperture will create the magnetic fields. There will be a voltage difference on the aperture sides, Aperture dimensions for SE = 60 dB 150 d (mm) 100 50 0 1 10 100 1000 f (MHz) Figure 8.2 Aperture dimensions for SE = 60 dB.
  • 147. 8.1 Field-to-Aperture Coupling 139 which will result in an electric field. It is desirable that the apertures interfere with those currents as little as possible, as can be seen from Figure 8.3. As can be seen in Figure 8.3, smaller apertures will not stop the currents as much as a large one will. The resonant frequency of the smaller aperture will be much higher than that of the larger aperture. Since every aperture in the shield lessens its effectiveness, the shielding effective- ness of multiple apertures is degraded compared to a single aperture by 20logn, where n is the number of apertures. That means that two apertures will degrade the shield by 6 dB, four apertures by 12 dB, and so forth. This works only up to the point where the wavelength becomes comparable with the size of the small aperture arrays, or in the case when the apertures are not close to each other compared to the wavelength. If the apertures are placed at a distance more than half of a wavelength apart, they can be considered individual apertures. At a frequency of 1 GHz, half of a wavelength will be 15 cm. The smaller the aperture, the less the electromagnetic fields penetrate inside the shield. This is shown in the Figure 8.4. The electromagnetic field will not penetrate very deep if the aperture dimension is small compared to the wavelength. The effec- tiveness of the shield at the distance l from the shield, depending on the diameter d of the aperture, will be 1 20 dB if ≈1 d 1 40 dB if ≈ 2 d 1 60 dB if ≈ 5 d where λ > d. The rule of thumb is that d/ ≤ 30. For the frequency f = 1 GHz or λ = 30 cm, d can be 1 cm at most. According to the above rule, the shielding effectiveness of 60 dB will be achieved at the distance of l = 5 cm from the aperture. One large aperture Several smaller apertures Figure 8.3 Multiple aperture currents. Figure 8.4 Penetration of the electromagnetic field through the apertures.
  • 148. 140 Electromagnetic Field-to-Aperture Coupling The above rules are only an approximation. In reality they will depend on the thickness of the shield, the type of the material used, and on the distance of the inner cables or wires to the shield aperture. It is recommended to have an additional shielding effectiveness of 20 dB. 8.1.3 Waveguides Below Cutoff When the aperture dimension becomes too small, waveguides below the cutoff fre- quency should be used (Figure 8.5). There can be just one or multiple waveguides forming the honeycomb. They are often used for ventilation. Placing waveguides inside the aperture holes can reduce the emission of the waves through the aperture. Multiple reflections inside the waveguide’s walls will reduce wave strength. The size of the aperture can be much larger when using waveguides below cutoff than when not using them. The characteristics of the waveguide are determined by its geometrical dimen- sions: gap (g) and height (h). A waveguide will allow all the waves to pass when its internal diagonal (g) is half of the wavelength. Thus, the cutoff frequency of the waveguide is determined as 150,000 fc = (8.3) g where fc is in megahertz and g in millimeters. Below its cutoff frequency, the wave- guide does not leak very much, and it will provide sufficient shielding for f fc/2. The attenuation (SE) of the waveguide dependent on the frequency is given by 2 ⎛f ⎞ 1− ⎜ ⎟ ⎝ fc ⎠ SE = 27.2 h g [dB] (8.4) where both h and g are given in mm. Figure 8.6 shows the shielding effectiveness (SE) depending on the frequency for h/g = 3. For f fc/2, according to the above graph, the SE will be about 70 dB, which is sufficient for most purposes. A smaller g results in higher cutoff frequency, while a larger height h increases the value of shielding effectiveness (SE). h g Figure 8.5 Waveguides below cutoff.
  • 149. 8.2 Reflection and Transmission 141 90 80 70 60 50 SE 40 30 20 10 0 0,2 0,0 0,4 0,6 0,8 1,0 f/f c Figure 8.6 SE of waveguide below cutoff versus frequency for h/g = 3. If multiple waveguides similar to multiple apertures are used, there will be a reduction in shielding by 10logn, where n is the number of apertures. Thus, 10 aper- tures will have 10-dB less attenuation than a single waveguide. If conductors are placed inside the waveguides below cutoff, they will not be equally effective, and the shielding effectiveness will be greatly reduced, so this should be avoided.8.2 Reflection and Transmission The shielding theory is based on two mechanisms: reflection and transmission (absorption) losses. When an electromagnetic wave in free space—with electric and magnetic fields perpendicular to each other (TEM mode of propagation)—hits a metal wall, one part will be reflected depending on the angle of incidence, and the other part will travel through the metal wall with attenuation (Figure 8.7). The attenuation of the electromagnetic wave will be exponential, depending on the skin depth, δ, and the distance from the border of the two mediums (free space and metal), d. The absorption loss through the metal shield at distance d can be cal- culated from ( S A = 20 log10 e d δ ) (8.5) or S A = 8686( d δ)[dB] . (8.6) where δ is the skin depth at which the field intensity drops to the value of 1/e: δ = 2 ωµ 0 µ r σ (8.7)
  • 150. 142 Electromagnetic Field-to-Aperture Coupling E −d/δ E*e P P −d/δ H*e H Metal wall Figure 8.7 Absorption loss in the shield. with f being the frequency, µr being the permeability of the material, and σ being the conductivity of the material. Absorption loss SA increases with frequency. The necessary shield depth, d, is getting smaller with the rise of frequency, which means that the shield will be more effective on a higher frequency than on a lower frequency. Figure 8.8 shows the alu- minum shield depth, d, for SA of 100 dB and 60 dB versus frequency. Figure 8.8 shows that at higher frequencies a good shield can be achieved with a very thin aluminum foil. The results will be similar for any other metal. A part of the electromagnetic field will not be absorbed but reflected. The free space has an impedance of 10 1 Sa = 100 dB 0,1 Sa = 60 dB 0,01 0,001 d (m) 0,0001 0,00001 0,000001 0,0000001 1,E+00 1,E+01 1,E+02 1,E+03 1,E+04 1,E+05 1,E+06 1,E+07 1,E+08 1,E+09 f (Hz) Figure 8.8 Necessary aluminum shield depth versus frequency.
  • 151. 8.2 Reflection and Transmission 143 µ Z0 = = 120π = 377 Ω (8.8) ε The metal shield has a smaller impedance than free space: ωµ Zs = (8.9) σ Since the two impedances are different, there will be a reflection. The transmis- sion coefficient rt1 at the border of the air and the shield is given as 2Z s rt 1 = (8.10) Z0 + Zs The electromagnetic wave must exit the shield again into free space (air) as shown in Figure 8.9, and the transmission coefficient at this second border of the two mediums will be: 2Z 0 rt 2 = (8.11) Z0 + Zs The total transmission rttot is given as a product of rt1 and rt2: 4Z 0 Z s rttot = (8.12) (Z0 + Zs ) 2 Total reflection and transmission is equal to the incident wave: rt + rr = 1 (8.13) Since a metal shield has a much smaller impedance than free space, the reflec- tion loss, SR, is given as d Ei Et Er E tr E tt Air Air Metal Figure 8.9 Reflection and transmission of an electromagnetic wave at a metal shield.
  • 152. 144 Electromagnetic Field-to-Aperture Coupling Z0 Z0 SR = = (8.14) 4Z s 4 ωµ r µ 0 σ or in [dB], Z0 S R = 20 log10 (8.15) 4 ωµ r µ 0 σ The reflection loss decreases with the frequency. Figure 8.10 shows the reflec- tion loss of the aluminum shield depending on the frequency. The total shield loss will be the combined reflection and absorption loss: S = S A S R = S A [dB] + S R [dB] (8.16) Total shield loss for aluminum foil 0.1 mm in depth is shown in Figure 8.11. Total shield loss for aluminum foil that is 0.1 mm in depth is shown in Figure 8.11. The figure shows that the reflection loss is dominant on lower frequencies, up to 10 MHz; at 100 MHz their contribution is about the same, and on higher fre- quencies absorption loss becomes dominant. At even higher frequencies, the shield thickness has almost no influence at all. Total shielding loss is the combination of both contributions (full line); it stays above 100 dB on all frequencies. Other metals give similar results. If the shield consists of several laminate layers, the total reflection and absorp- tion losses will be a sum of reflections between each layer and the attenuation in every layer. The above discussion is valid for far field conditions where E/H = Z0 = 377 ohms. In the near field, the ratio of the electric and magnetic field is different—it changes depending on the distance from the electromagnetic source. Therefore, the shield effectiveness should be considered separately for electric and magnetic fields. 170 160 150 140 130 S R (dB) 120 110 100 90 80 70 1,E+00 1,E+02 1,E+04 1,E+06 1,E+08 1,E+10 f (Hz) Figure 8.10 Aluminum shield reflection loss.
  • 153. 8.2 Reflection and Transmission 145 450 400 350 300 SR 250 S SA 200 S 150 100 50 0 1,E+00 1,E+02 1,E+04 1,E+06 1,E+08 1,E+10 f (Hz) Figure 8.11 Reflection and absorption loss for an aluminum shield versus frequency. 8.2.1 Electric Field Shielding against the electric field is usually made with the Faraday cage (Figure 8.12). The Faraday cage can be a sphere, rectangle, or any other shape. When placed inside an electric field, it will not absorb the field, but rather produce electric potential of different polarity along the edge of the cage. This will create an opposite electric field, which will result in no electric field inside the cage. The earlier discus- sion proved that the thickness of the cage (shield) is not very important. If the shield has an aperture, the electric field will penetrate inside the shield, and with a wire in the vicinity of the shield, there will be an induced voltage along the wire inside the shield as is shown in Figure 8.13. This figure shows that the elec- tric field intensity falls with the distance from the aperture. It is therefore advisable to place unshielded wires carrying information as far away from the apertures as possible, or to place the apertures away from the critical areas. Charged metal wall E Faraday cage E=0 Figure 8.12 Faraday cage.
  • 154. 146 Electromagnetic Field-to-Aperture Coupling Charged metal wall Wire Shield with aperture E E≠0 E Figure 8.13 Shield with an aperture. 8.2.2 Magnetic Field The Faraday cage does not work for the magnetic field. The magnetic field (tangen- tial to the shield) will penetrate the shield that works for the electric field. The atten- uation of the magnetic field can be done with material that has a permeability much larger than 1 (µ > 1) as is shown in Figure 8.14. The magnetic field stays in the shield; there will be no magnetic field inside the shield if the permeability of the material is great enough. If the shield has an aperture, the magnetic field will pene- trate the shield the same way an electric field penetrates the Faraday shield. The magnetic shield can also be made with a thin conducting material of small permeability for AC. The alternating magnetic field will create the eddy currents, which flow along the shield. These currents create a magnetic field of the opposite Wire B>0 Magnetic shield with µ >> 1 H B=0 Figure 8.14 Magnetic shield.
  • 155. 8.3 Equipment Shielding 147 direction, which will cancel out the outer magnetic field. For higher frequencies, this effect will be stronger. Such an aluminum shield will be sufficient protection against the magnetic field at 50/60 Hz for power lines. It is harder to make the shield for low-frequency magnetic fields than it is for high frequency magnetic fields. The magnetic shield should have as few apertures/holes as possible to sustain shield efficiency. All the apertures (doors, windows, ventilation, cable openings, and so forth) will compromise the shield and allow the tangential magnetic fields to enter the shield. If there should be a cable or wire in the vicinity of the aperture, there can be an induced current, which will be carried further inside the shield and interfere with other equipment (Figure 8.15).8.3 Equipment Shielding As can be concluded from the above discussion, it is easier to make a shield against the electric field that it is for the magnetic field. Even a very thin metal sheet will provide good shielding effectiveness, especially at high frequencies. Generally it is advisable to keep the wires and equipment as far as possible from the shield walls and the apertures in it. However, today the industry is constantly trying to make electronic devices smaller, and big shields are not practical. If the shield has parallel walls, there will be standing waves between them and thus reso- nances. It would be best if the shield would be of an irregular shape, which is impractical. It is better to use the rectangular than the cubic shape for the shield in order to lower the number of resonances. Another problem arises when apertures are to be placed in the shield. 8.3.1 Gasketing Gaskets are elements that are placed inside the apertures to ensure continuity of the shield (Figure 8.16). This can also include doors, windows, or other apertures. Gas- Wire H Shield Figure 8.15 Magnetic field entering through aperture in the shield.
  • 156. 148 Electromagnetic Field-to-Aperture Coupling Gasket Shield Figure 8.16 Gasket in the shield. kets should be able to withstand environmental conditions such as temperature, salt, moisture, and heat for a long time. After a certain period of time they should be replaced. If the gasket is made of the same material as the shield, theoretically the currents in the shield could flow without interruption. This is hard to achieve due to mechanical constraints. Gasket types include spring fingers, metal meshes, and conductively wrapped polymers. Spring fingers (Figure 8.17) are usually made of beryllium copper. They are placed on frequently used doors and must be pressed tightly to achieve a good impedance match with metal walls. As the frequency rises, the finger size should get smaller. Metal meshes consist of elastomer with impregnated metal particles in them. Such a gasket can also function as an environmental seal. Conductively wrapped polymers are a combination of polymer foam or tube and an outer conductive coat- ing. They are flexible and do not require high contact pressure, which makes them vulnerable to environmental conditions. Gaskets should be painted with conductive paint only. They have a wide area of use, from wireless communications (also in mobile phones) to facilities used for test- ing immunity and emission. Gasket attenuation of an electric field is between 40 dB and 60 dB. The frequency of use is from 10 kHz up to 20 GHz. 8.3.2 PCB Protection Complete PCB shielding will have a six-sided metal box around it and shielded con- nectors and filters for power and signal cables going in and out. Another possibility is to have a five-sided metal box attached to the ground plane in several places to create a Faraday cage. There will still be the problem of apertures in the ground plane, the connections between ground plane and metal shield, and in the shield itself (ventilation). Figure 8.17 Spring fingers.
  • 157. 8.3 Equipment Shielding 149 Generally it is better to have as many tracks or striplines as possible instead of wires and cables. The striplines can be filtered with feed-through filters, or at least with ferrite beads or capacitors. Unfiltered signals should not be close to the filtered ones. The cables entering the PCB should also at least have ferrite beads if they are not filtered. Figure 8.18 shows the PCB shielding. Besides shielding, the design of PCB can improve electromagnetic interference suppression. Every cable is a possible antenna. The PCB traces (striplines) do not radiate as much as cables or wires because of much smaller dimensions. However, they might still radiate, especially when combined with cables going out of the shield. With a proper design, this effect can be minimized. The goal in designing PCB traces is to have a return signal trace close to the one going in the opposite direction. In this way, the electromagnetic fields of both traces will cancel each other out. Figure 8.19 shows a typical PCB connected to the power cable (this can be a sig- nal cable as well). A PCB with a power cable connected can be seen in Figure 8.19(a). The PCB trace line in combination with the power cable can form a struc- ture, which resembles a dipole [Figure 8.19(b)]. The currents flow in the same direc- tion and the radiation is strong and increases with the frequency. This can be prevented with the mains filter (or ferrite beads) on the power cable. On the other hand, another PCB trace going in the opposite direction will form a structure similar to the transmission line [Figure 8.19(c)]. Here, the currents will cancel each other out and the total radiation will be small. The efficiency of the antenna is higher when the antenna is large; cables repre- sent the largest possible antennas, so the currents in the cables are the greatest possi- ble sources of interference. Therefore, the PCB design should be used for transmission line structures and not dipole structures. 8.3.3 Magnetic Shield Magnetic shielding is needed in protecting computer hard disks, speakers in audio engineering, power sources, and so forth. Shielding from the magnetic field can be Mains filter Shield Ventilation Ground plane Signal filters Figure 8.18 PCB shielding.
  • 158. 150 Electromagnetic Field-to-Aperture Coupling I Power cable I Z (a) I I Z Dipole I I Transmission line (b) (c) Figure 8.19 Dipole and transmission line structures in the PCB. (a) PCB structure, (b) dipole, and (c) transmission line. done with magnetic materials of high permeability. Materials for magnetic shield- ing differ in saturation (in Gauss) and permeability (Table 8.1). Other characteristics of magnetic materials include the loss factor, Curie tem- perature, density, and resistivity. Amumetal is used for high attenuation in a small space. It is available in the thickness from 50 µm to 3 µm. For smaller attenuations, a more economic material like UCLS can be used. S1 and L8 are used for ferrite toroids in the EMI suppression. The attenuation of a circular shield can be calculated from µ⎛ r2 ⎞ S= ⎜1 − o2 ⎟ (8.17) 4⎝ rt ⎠ where µ is the material permeability, and ro and ri are the outer and inner shield radiuses. Table 8.1 Magnetic Material Properties Material Saturation Permeability r Amumetal 8,000 400,000 Amunickel 15,000 150,000 ULCS 22,000 4,000 L8 2,550 1,500 J70 2,500 620 M7 2205 160 S1 1625 120
  • 159. 8.3 Equipment Shielding 151Selected Bibliography Gooch, J. W., and J. K. Daher, Electromagnetic Shielding and Corrosion Protection for Aerospace Vehicles, New York: Springer, 2007. Kaires, R. G., “Stopping Electromagnetic Interference at the Printed Circuit Board,” Conformity, November 2003, pp. 12–21. Moongilan, D., and E. Mitchell, “EMI Gasket Shielding Effectiveness Evaluation Method Using Transmission Theory,” Proc. IEEE International Symposium on Electromagnetic Compatibility, August 18–22, 2008, pp. 1–6. Pothapragada, P., “Selecting Material for Shielding Enclosures,” Conformity, November 2003, pp. 37–39. Raza, I., “Faraday Cage Enclosures and Reduction of Microprocessor Emissions,” Compliance Engineering, 2001. Strauss, I., “Shielding Review,” Conformity, April 2004, pp. 24–32. Vasquez, H., et al., “Simple Device for Electromagnetic Interference Shielding Effectiveness Mea- surement,” IEEE EMC Society Newsletter, Winter 2009, pp. 62–68.
  • 160. CHAPTER 9Electrical Grounding and Bonding The electronic equipment’s safety should be dealt with prior to the electromagnetic interference problem. This is why grounding and bonding are important in a proper communication system design. For every signal current sent to the load there will be a return current path (or several of them). Knowing where return currents flow can help minimize interfer- ence and improve safety. Grounding and bonding are methods of connecting equip- ment or cables to each other or to the Earth in order to ensure safety and current flow. Grounding is a procedure in which conductive equipment is connected to the Earth for safety reasons. The conductor that connects equipment to the Earth is called the grounding electrode. If there is an unintentional connection between the equipment and the ungrounded conductor, there will be a ground fault. In this case, a ground-fault current may exist from the ground fault to the electrical supply source and not to the Earth. Not every conductor through which the current flows can be used for grounding. Grounding consists of the following electrically inter- connected subsystems: the Earth electrode subsytem, the fault protection system, the lightning protection subsystem, and the signal reference subsystem. All conductive objects or equipment that are not grounded or electrically iso- lated from the ground by nonconductors or gaskets should be bonded. An untreated isolated conductive object, if charged, can cause static electricity discharge. Bonding is a process of making a low impedance path for the flow of electric current between two metallic objects. The two conductive surfaces are electrically connected to each other to prevent electric potential between metal surfaces, which can cause interference or sparking. There are various bonding methods, which will be discussed in Section 9.4. To provide a mechanically strong and low impedance path for the current flow and to achieve grounding, different metallic objects must be connected (bonded) together. The bonds should be made in such a way that the junction itself does not determine the electric and mechanic properties of the junction. The bond properties should primarily be determined by the metallic objects (members) that are to be connected. Grounding and bonding cables should have low resistance and the ability to withstand environmental conditions and the passage of time. Contact with the metal surface must be kept at all times, regardless of paint loss, corrosion, and sur- face contamination. If the resistance between bonded objects and the grounding conductor (i.e., 25Ω) is specified, the bonding can be easily tested with simple instruments. 153
  • 161. 154 Electrical Grounding and Bonding9.1 Grounding for Safety Grounding is a process of connecting metal parts to the Earth for protecting person- nel and facilities, as well as for lightning and electrostatic discharge protection. A person can touch the metal enclosure and experience a shock. Electronic devices with AC currents should have grounding incase a current flows between the power source and enclosure, which is usually the biggest conductor or metal plate available. Lightning protection includes low impedance conductors (low resistance and low inductive reactance), which are used to prevent arcing between nearby metallic objects. Grounding must also be performed for protection against electrostatic dis- charge (ESD) from people, furniture, or other objects that can be charged. The grounding should have a path back to the ground for the discharge current. Grounding can also be used for signal reference. To have an unchanging reference voltage, stray currents must be kept away from the reference ground. As long as the signal reference wire is connected only to one place of another ground, there will be no stray currents. If, however, the signal reference wire is tied to some other ground at two or more points, there will be noise currents causing interference. Short, wide wires are better for grounding than long, thin wires because of lower inductance. Round wires should be avoided for grounding, because they have the highest inductance. If the wire is grounded at more than one point, so called ground loops will be created with interference voltages between them. A connection to the Earth can be through capacitive coupling, accidental contact, and intentional contact. The ground is a direct path of low impedance between the Earth and differ- ent communication or electronic equipment. The fault protection subsystem ensures that personnel are protected from shock hazard. In addition, equipment should be protected from damage resulting from faults in the electric system. This is usually done with a green wire placed inside the device, which represents the Earth. The fault protection wire should be separated from the signal reference ground, except at the Earth electrode subsystem. 9.1.1 Shock Control Human resistance is between 1 kΩ and 10 kΩ, depending on the person’s moisture and wetness levels. This means that voltages up to 50-V AC cannot harm humans. However, higher voltages might cause damage to or even kill a person. This is why electronic equipment is grounded with a metal shield. Personnel operating communication equipment should follow safety measures when working, repairing, installing, and operating dangerous equipment. They must make sure that high-voltage devices have been grounded. Figure 9.1 shows a typical case of a shock hazard. Figure 9.1(a) shows an example of no ground protec- tion. When operating normally, the current will flow only through the intentional resistance R on the return path. If there is an accidental short circuit to the casing of the electronic equipment, the casing will become a shock hazard. If a person touches the frame, the current will flow through the connection of the frame and the equip- ment (RS) and then through the person (RP) touching the casing to the ground. A current of 75 mA through the body can be fatal. This can be avoided by adding a
  • 162. 9.1 Grounding for Safety 155 Accidental short Electronic device Electronic device RS 220/110 V RS 220/110 V R R RP RP 0V 0V IS IS Ground Ground No ground protection Ground protection (a) (b) Figure 9.1 (a, b) Shock hazard. grounding wire to the equipment casing [Figure 9.1(b)]. The person touching the casing will be protected from electric shock. If there is a fault in the grounding, further protection can be ensured by using fil- ters for electromagnetic interference. Capacitors (0.1 µF for 50/60 Hz) or filters can be placed from hot and neutral wires to the ground wire. If there is a ground fault in the electronic equipment, but there is additional protection with EMI filters or capacitors, the person touching the casing will not experience currents exceeding 5 mA through his or her body, which is safe. 9.1.2 Fault Protection Fault protection includes a path with low resistance between the location of the fault and the power source. The low resistance path in the building is provided by the green wires. If there is contact between the ground wire and energized conduct- ing objects, the fuses will blow and protect the power source from further damage. The danger from the fault (shock) depends on its duration. That is why fast fuses must be used. The longer the time of fault exposure, the higher the temperature, which can cause a fire. Faults can occur either as a direct short or as an arc. The cause of direct short can be: rodents, water, moisture combined with dirt on insulator surfaces, overload, deterioration from age, and damage from improper installation. The currents in the ground can result in the casing having a higher potential than the ground. The energy from the fault can result in high temperatures, which can damage the equip- ment or the personnel, or result in a fire. For single phase AC power distribution (Figure 9.2), the ground conductor (green wire) must be one of the four wires. The other three wires are the two phase hot black and red wires and the neutral white wire. The ground wire will carry cur- rent only if there is a fault. The hot wires are connected to the high sides of the distri- bution transformer secondary. The neutral white wire is grounded at the service disconnecting means. The ground green wire is grounded at the supply side of the first service disconnect to the Earth electrode and also to the ground terminal at the distribution transformer. All metal parts should be connected to the green ground wire. The three phase system is similar to the single phase system. There are three phase conductors, one neutral, and one ground. The ground (safety) wire must be
  • 163. 156 Electrical Grounding and Bonding Distribution Disconnecting transformer means Hot 115 V Neutral 230 V 115 V Hot Ground Figure 9.2 Single phase AC power ground connection. connected to the Earth electrode both at the supply side of the first service discon- nect of the facility, as well as at the distribution transformer. The neutral wire must be grounded at both locations as well. More explanations on ground connections can be found in the literature at the end of this chapter.9.2 Grounding for Voltage Reference Control Signal loss and hardware malfunction in the communication link between two elec- tronic devices that use the Earth ground as a voltage reference can occur due to unwanted noise in the ground loop. The cause can lie in bad grounding, a lightning strike, electrostatic discharge, or ground faults. Signal reference has a common reference for all of the equipment for minimization of currents between the equipment and elimination of the noise volt- ages on signal paths. It can also be a bus or a conductor for an internal circuit refer- ence of the electronic device. Signal circuits are referenced to the ground for establishing signal return paths between a load and a source, providing fault protection, and controlling electro- static discharge. Grounding at low frequencies depends on the surface, length, resis- tance, inductance, and capacitance of the conductor. At higher (RF) frequencies, the conductor must be considered a transmission line. The signal reference in equipment can be a single reference plane or a grid, or more precisely a floating ground, single point ground, or multipoint ground, which is actually an equipotential plane. 9.2.1 Floating Ground The floating ground shown in Figure 9.3 is used as signal reference for a number of electronic devices in a facility. The floating ground is isolated from the building or facility ground and conductive objects. The noise currents will not be conductively coupled to the signal circuits. How effective the floating ground will be depends on its isolation from the con- ductors in the vicinity. In large facilities it is hard to achieve and maintain a com- plete floating ground. The problem lies in the electric static, which can occur in isolated signal circuits, especially if placed near voltage power lines. Electric static
  • 164. 9.2 Grounding for Voltage Reference Control 157 Electronic devices Fault protection Fault protection Signal reference Ground Figure 9.3 Floating ground. can also cause sparks or shock. Most electronic equipment is referenced to the Earth ground, so the power faults in the signal system can cause electronic devices to rise to dangerous voltage levels relative to other conductive objects in the facility. In addition, in the case of lightning, since the conductors are not coupled together the whole system can have an increase in voltage resulting in the breakdown of insula- tion and arcing. Therefore, the floating ground solution for the signal reference voltage is not the best possible solution. 9.2.2 Single Point Ground For signal reference, a better solution than the floating ground is the single point ground (Figure 9.4). In this design, the signal paths are referenced to a single point, which in turn is connected to the ground. Ideally, each of the electronic devices as well as circuits inside the devices should have separate ground conductors. This requires a very large number of long wires, which is impractical. The above solution is intended for frequencies up to 300 kHz. The noise coupled in the single point ground is not conductively coupled into the signal circuitry over signal ground wires. However, at higher frequencies, single point grounds become transmission lines, where the ground is the other side of the line. In addition, every part of the equipment bonded to the transmission line is actually a tuned stub (see Section 3.1). At different frequencies, the single point Electronic devices Fault protection Fault protection Signal reference Ground Figure 9.4 Single point ground.
  • 165. 158 Electrical Grounding and Bonding ground will have a different impedance appearing either as an inductor or capacitor and therefore not as a ground. Large facilities also require long wires, which can act as antennas. There is also a problem of stray capacitance between the wires. With all of the above in mind, the single point ground is not recommended for communication systems. 9.2.3 Multipoint Ground A multipoint ground (Figure 9.5) has many conductive paths from the ground to various electronic devices. Inside each electronic device, circuits are multiply con- nected to the ground. Multipoint grounding is effective for high frequency signal circuits. Coaxial cables can be easily interfaced since their outer conductor or shield does not have to be floated relative to the equipment casing. If the length of conductors is longer than λ/8 for the highest frequency of use, a multipoint ground will require an additional equipotential ground plane for effective ground. Care must be taken to prevent 50/60-Hz power currents, as well as any low fre- quency currents of high amplitudes, flowing through the ground system to be con- ductively coupled to the signal circuits and introduce noise voltages. 9.2.4 Equipotential Plane Grounding does not always reduce all interference. In some cases it can increase interference by providing conductive coupling paths as well as inductive loops—or it can radiate. This can be countered by placing an equipotential plane on the floor beneath electronic devices being grounded. The equipotential plane is a large con- ducting material with negligible impedance. The equipotential plane can also be placed above the electronic devices if it is impossible to install it below them. Why is an equipotential plane better than a grounding wire? The characteristic impedance is a function of L/C , so if the capacity C increases, the characteristic impedance decreases. The capacity of a large metallic sheet is much larger than that of a wire. In addition, the inductance L is decreased with width, which decreases the characteris- tic impedance even more. An equipotential plane with large dimensions has a very Electronic devices Fault protection Fault protection Figure 9.5 Multipoint ground.
  • 166. 9.3 Bonding for Current Control 159 low characteristic impedance for a wide frequency range; it represents a reference plane for all the electronic devices bonded to it. Compared to other grounds, the equipotential plane represents a safer protec- tion for personnel, since there is no need for long grounding wires. If placed below an antenna, the equipotential plane protects the antenna from radiating cables or wires that might be placed below it. The equipotential plane can be built with a copper grid placed in a concrete floor, an aluminum (or copper) screen placed under the carpet/floor tile, or on the ceiling. The equipotential plane must be bonded to the Earth electrode at several points.9.3 Bonding for Current Control Bonding is a procedure that permanently joins two metallic parts to form a low impedance connection, which will ensure electrical continuity as well as capacity to conduct any current. Bonding also ensures equal potential between separate con- nections to the ground. In every communication system there are many interconnections between metallic parts in order to provide lightning and fault protection, reference signals, and power. These connections must be made in such a way as not to change the elec- tric path’s properties. The connection or junction must be strong and durable, have low impedance, and be resilient to corrosion. Bonding is used for prevention of static accumulation, lightning, shock, fault current return path, and minimization of RF potentials on casings. Improper bonds can cause load voltage drops or heat, which can result in a fire. In signal paths, bad bonds can cause noise and lower the signal level. Bonding can also influence shield effectiveness. Bonding must be done carefully for interference reduction using a lowpass π fil- ter for a power line as shown in Figure 9.6. The interfering high frequency current I1 will reach the ground through the ZB and should not reach the load, ZL. If the casing is not bonded properly to the ground reference plane, the bond impedance, ZB, could be relatively large compared to the reactance, XC, at the interfering frequency, and the interference current, I2, could reach the load and thus lower the filter efficiency. L C/2 C/2 I2 U ZL I1 ZB Figure 9.6 Improper bonding for a lowpass filter.
  • 167. 160 Electrical Grounding and Bonding 9.3.1 Bonding Classes Table 9.1 gives bonding classes. The class A bond is used for bonding antenna installations. Radiating elements have to be installed with a ground plane of negligi- ble impedance for operating frequencies of the antenna, but should not interfere with the antenna radiation pattern. The RF currents must have a low impedance path of a small length. For the coaxial antenna there must be continuity between outer conductors or the shield and ground plane of antennas. The class C bond reduces power and voltage losses. This bond type requires low impedance and low voltages in joints for assuring adequate power to the user. The class H bond protects against fire and shock to the personnel. It is applied to electronic devices required to carry the fault current. Bonding resistance for this class must be 0.1Ω or lower. The class L bond is used for lightning protection and must be able to endure very high currents (200 kA) and magnetic forces. Low inductance of the bond is required. The class R bond is applied in cases of RF noise, which is present in a wide frequency range. The R bond requires low RF impedance at high frequencies. The bonding resistance must be 5 mΩ or lower. Low inductance is also required. This bond type will be discussed in more detail below. The class S bond protects against electro- static discharge. The bonding resistance must be 1Ω or lower. All isolated conduc- tors with dimensions greater than 7.5 cm must be bonded with the S bond because they can be charged. 9.3.2 Strap Bond for Class R Isolated metallic elements whose linear dimensions are close to half of the wave- length (λ/2) associated with the operating frequency can act as an antenna and receive RF signals as well as produce enough voltage to cause discharge to other electronic devices or circuits. The strap bond discussed in this section is an indirect bonding type. The inductance L of a thin metal strap is given with ⎡ ⎛ 2l ⎞ ⎛ w + t ⎞⎤ L = 0002l ⎢ln ⎜ . ⎟ + 05 + 02235 ⎜ . . ⎝ l ⎠⎥ ⎟ [ µH] (9.1) ⎣ ⎝w + t⎠ ⎦ where l is the length, w is the width, and t is the thickness of the strap in centimeters. If the strap is round, the inductance is given as ⎡ ⎛ 4l ⎞ ⎤ L = 0002l ⎢ln ⎜ ⎟ − x ⎥ . ⎝d⎠ [µH] (9.2) ⎣ ⎦ Table 9.1 Bonding Classes Class Application A Antenna installation C Current path return H Shock hazard L Lightning protection R RF potential S Static charge
  • 168. 9.3 Bonding for Current Control 161 where d is the diameter of the strap in cm. For low frequencies x = 0.75 and for high frequencies x = 1. The inductive reactance of the strap is X L = ωL = 2 πfL (9.3) where f is the frequency in hertz and L is the inductance in H. If, for example, the round strap has the following characteristics: resistance (l = −6 2 20 cm) R = 32.2 · 10 Ω, length 20 cm, diameter 1 cm , thickness 0.1 cm, and width 6 cm, its inductance will be according to (9.2): L = 0.1453 µH. The capacitance of the bond can be found from S C=ε (9.4) d where ε is the dielectric constant, d is the distance between mating surfaces, and S is the area of mating surfaces. The capacitive reactance is 1 XC = (9.5) 2πfC where f is the frequency and C is the capacitance. If the square area of the bond is 200 cm2 and the bond is covered with a 0.02 cm layer of nonconductive paint, the bond capacitance will be according to (9.5): C = 8,852 pF. The bond impedance depends on the frequency. The capacitive reactance is high at low frequencies and decreases as the frequency increases. The inductive reactance increases with frequency. At a resonant frequency, the impedance will reach its maximum, which can be more than a thousand Ωs. The resonant frequency is found from 1 fr = (9.6) 2π LC The resonant frequency for the above example is 4.44 MHz. The inductive and capacitive reactance will be XL = 4.05Ω and XC = 4.05Ω. The impedance at the reso- nance is found from X2 Zr = (9.7) R where X is the inductive or capacitive reactance at the resonant frequency and R is 5 the resistance of the strap. For the above example, Zr will be 5.1 · 10 Ω.The obtained value is much higher than the required 5 mΩ. That is why bonding straps must be checked for their resonance.
  • 169. 162 Electrical Grounding and Bonding 9.3.3 Resistance Requirements The most important requirement for bonding is the low resistance path between the two parts that are to be joined. For the antistatic discharge, even the relatively high resistance of 50 kΩ is sufficient. This value, however, is not sufficiently low enough for lightning protection or fault currents. If bonding is used for voltage reference, the necessary resistance will depend on the estimated voltage and current levels. If the surfaces of the two parts to be joined together are properly cleaned, and the pressure used for bonding the mating surfaces is continuous, the bonding resis- tance achieved can be as low as 1 mΩ. Any attempts to achieve lower bonding resis- tance than intrinsic resistance of the conductors are unnecessary. The surfaces have to be cleaned to prevent corrosion. This issue will be dealt with in Section 9.5. All the bond resistances to the ground equipotential plane should have the same resistances, which will ensure minimum voltage drops and lower the noise in the system. Low bond resistance at DC does not necessarily mean that it will stay low at high frequencies. The bond resistance at high frequencies will depend on path resonances, transmission line effects, stray capacitance, and con- ductor inductance.9.4 Types of Electrical Bonds Bonding two metal parts regardless of whether they are intended for lightning pro- tection, Earth electrodes, or mating of the equipment front panels to the equipment racks can be done with different direct or indirect electrical bonds. The best bond is achieved by welding and brazing. Silver soldering also makes a very good bond. Round wires used as jumpers do not make very good bonds because of high induc- tance. A metal strap has a much lower impedance than a round wire and is a better solution. The strap length to width ratio should equal 5 to 1, while the strap width to thickness should be 10 to 1 or more. If the bond length is equal or above 0.1λ for the corresponding frequency, the bond will not be effective. Whatever bonding method is applied, it is important to obtain electric continu- ity and keep the DC resistance and RF resistance as low as possible. Direct bonding (welding, brazing) is the best type and is used if the two members have no relative movement and the bond will be permanent. If the two members must be separated, indirect bonding (bolting, clamping, straps, or other auxiliary conductors) can also be used. Figure 9.7 shows the direct bond of two members, by both butt joint and lap joint. The current flowing between the two members will depend on the resis- Member 2 Member 1 Member 2 Member 1 I I Butt joint Lap joint Figure 9.7 Butt and lap joint.
  • 170. 9.4 Types of Electrical Bonds 163 tance of the two conductors and on bond resistance. The bond resistance increases the total resistance of the path, and it must be much smaller than the conductors’ resistances so that the path resistance depends primarily on the conductor resistances. Both the direct and indirect bonding methods will determine the bonding resis- tance, which will depend on the type of metal used, surface cleanness, contact pres- sure at the surfaces, and cross-section area of the mating surfaces. 9.4.1 Welding and Brazing Welding is the best bonding method in view of the electrical properties of the bond. Heat over 2,000°C cleans the metal surfaces from any contamination. The bond resistance is close to 0 due to a very short bond length compared to member lengths. The bond strength is equal to if not stronger than the strength of the members. Intensive heat prevents moisture penetration into the bond, so there is almost no corrosion. The longevity of the bond depends on the duration of its members. Although welding is expensive, it should be utilized for permanent bonds. Brazing (Figure 9.8), including silver soldering, is also a metal flow process sim- ilar to welding for permanent bonding. The temperature used in brazing is above 800°C, which is above the melting point of the brazing filler metal, but below the melting point of the bond members. The filler metal is used to make contact between the two metal members. The brazed bond resistance is close to 0, but since the filler metal is different from the bond members, corrosion is more probable than in welding. 9.4.2 Bolting In some cases permanent bonds are not desirable. This includes moving equipment from one place to another (e.g., for repairing) or disconnecting the connections. Less permanent bonds are also easier to achieve and are more flexible. One of the most used semipermanent bonds is the bolting connection (Figure 9.9) with bolts, screws, or similar fasteners, which should be able to sustain shock and vibrations. The bolts provide the necessary pressure between contact surfaces. The primary purpose of the bolts is not to be conductive; they do not even have to be metallic. The necessary pressure between the mating surfaces, which should be over 8 MN/m2, will determine the number of bolts. It is better to use more bolts for large connecting surfaces, or even to use rigid backing plates or clamps. Brazing filler metal Member 1 Member 2 Figure 9.8 Brazing.
  • 171. 164 Electrical Grounding and Bonding Member 2 Member 1 Bolt Bonding area Figure 9.9 Bolting. 9.4.3 Conductive Adhesive Conductive adhesive is a direct low resistance bond without the use of heat. Con- ductive adhesive is made of two-component silver-filled epoxy resin, which pro- vides good electrical conductivity. It is used in places where heat might damage the equipment or cause a fire. In combination with bolts, the conductive adhesive low- ers the danger of corrosion, at the same time maintaining high mechanical strength. The problem with conductive adhesive is that it is not easy to disassemble.9.5 Galvanic (Dissimilar Metal) Corrosion Control Corrosion is the deterioration of conductive material due to environmental influ- ences. Almost all environments are corrosive. This is especially true for industrial areas. Corrosion raises the required low resistance connection up to the point where it becomes unusable. Corrosion is actually a chemical process in metals (Figure 9.10). The metal surface will form an anode and a cathode due to impurities in contact through the metal. If there is an electrolyte or conducting fluid in the environment above the surface of the metal, the circuit will allow the currents to flow from the anode into the cathode and thus make corrosion possible. This will result in oxida- tion (i.e., the transfer of electrons from the metal into the environment/oxidizing agent). The oxidation will cause an electromotive force (EMF) between the metal and oxidation agent. Metal in contact with an oxidizing solution will cause a fixed potential difference compared to any other metal in the same condition. This set of Anode I Electrolyte Cathode I Metal Figure 9.10 Corrosion.
  • 172. 9.5 Galvanic (Dissimilar Metal) Corrosion Control 165 potentials, or EMF series, depends on the temperature and ion concentration in the solution, and is given for some metals in Table 9.2. Table 9.2 shows the relative tendencies of materials to corrode; higher values represent more of a chance for corrosion, meaning that aluminum is more likely to corrode than gold. The higher the potential is between two metals, the more dissimi- lar the metals and the more chance for corrosion. The metal with the higher voltage will be the anode and the metal with the lower voltage will be the cathode. The cur- rent flowing between the two metals will result in metal loss. This phenomenon is called galvanic corrosion. Metals in a bond should be of the same material or as sim- ilar as possible (close electrode potentials) to prevent galvanic corrosion. Corrosion can exist only if current can flow from anode to cathode. If there is water on the surface, the impurities of the water will support corrosion. Therefore, the metal parts have to be painted (coated), since paint prevents moisture from reaching the metal and thus provides the electrolytic path for the current between the anode and the cathode. If bonding is to occur between different materials, the member representing the anode (higher voltage) should always be larger than the one representing the cath- ode (lower voltage) as shown in Figure 9.11. If, for example, a copper strap is bonded to an iron plate, the iron will not corrode much because of the large anodic area. In reverse, an iron strap in contact with a copper plate will corrode very fast due to a small anodic area. It is desirable to cover both the anodic and the cathodic member with paint. Never should only the anode be painted, since this will speed up the corrosion—the small breaks in the paint actually become a small anodic area. Before bonding, the metal members should be cleaned with a wire brush, steel wool, or other means to achieve a bright metal finish. Sometimes chemical cleaning Table 9.2 Standard EMF Series Electrode Metal Potential (V) Aluminum 2.37 Iron 0.440 Tin 0.136 Lead 0.126 Copper −0.337 Silver −0.799 Gold −1.5 Moisture Anode Cathode Figure 9.11 Dissimilar junction.
  • 173. 166 Electrical Grounding and Bonding will be necessary. It is a good idea to use washers for bonding as well, because they can be easily replaced in case of corrosion. At the end, a protective coating should be applied for anticorrosion, and the bonds should be tested periodically to ensure that their performance is satisfactory.Selected Bibliography Military Handbook, Grounding, Bonding and Shielding For Electronic Equipment and Facilities, MIL-HDBK-419A, December 29, 1987. MIL-STD-1542B(USAF), Electromagnetic Compatibility and Grounding Requirements for Space System Facilities, November 15, 1991. MIL-STD-188-124B, Grounding, Bonding and Shielding for Common Long Haul/Tactical Com- munication Systems Including Ground Based Communications—Electronics Facilities and Equipments, February 1, 1992. MIL-STD-1310G, Standard Practice for Shipboard Bonding, Grounding, and Other Techniques for Electromagnetic Compatibility and Safety, June 28, 1996. NASA-STD-P023, Electrical Bonding for NASA Launch Vehicles, Spacecraft, Payloads, and Flight Equipment, August 17, 2001. NSTS 37330, Space Shuttle Bonding, Electrical, and Lightning Specifications, December 2, 1999.
  • 174. CHAPTER 10Emissions and Susceptibility—Radiatedand Conducted10.1 Control of Emissions and Susceptibility—Radiated andConducted Electromagnetic interference (EMI) can be regarded as a sort of environment pollu- tion with consequences similar to those of poison chemicals, automobile gas exhaust, and so forth. The electromagnetic spectrum is a natural source that has been devastated to a large extent in the last hundred years. The spectrum is full, which is why new technologies operate on increasingly higher frequencies. Modern life depends on the systems using the electromagnetic spectrum, and its protection is a priority. Uncontrolled electromagnetic radiation can lead to equipment damage, loss of money, injuries, and even death. Electromagnetic interference introduces unwanted voltages and currents into the equipment—the victim. This can lead to audio noise in radio receivers, as well as snow or picture loss on TV receivers. When the victims are communication links or computer systems operating industry facili- ties, the damage is even greater. Interference can find its way to the victim in two ways: through cables (conducting) and by electromagnetic radiation. 10.1.1 Sources of Electromagnetic Interference Every electric or electronic device that changes voltage or current can be an EMI source. Electric devices can introduce interference by electromagnetic radiation or through the cables. Electric shavers and dish washers can cause interference on TV sets, not only directly via radiation but by power cables as well (Figure 10.1). Generally, a faster change of voltage or current will result in a wider interfer- ence spectrum. Similarly, a higher voltage or current level will cause a higher con- ducted or radiated interference level. Therefore, electric machines generating high voltages and currents with a fast rise will be strong EMI sources. Table 10.1 shows the EMI sources regarding usage. EMI sources can also be divided into continuous and transient. Radar, although an impulse system, has a wide but stable RF spectrum and is considered a continu- ous EMI source. The radiation from continuous EMI sources can best be analyzed with the spectrum analyzer. On the other hand, lightning or nuclear electromag- netic pulse occurs unpredictably and is called transient radiation. Transient signals are very short and have a wide spectrum. They are analyzed much easier in the time 167
  • 175. 168 Emissions and Susceptibility—Radiated and Conducted Figure 10.1 EMI from an electric shaver and dish washer to the TV receiver.Table 10.1 EMI Sources Regarding UsageManmade Sources Vehicle Engines and Industrial/ Customers ESDCommunication Power Lines Systems Tools CommercialBroadcasting Generators Automobiles Compressors Dielectric heaters MicrowavesRadar Converters Ignitions Saws Air conditioners OvensWalkie-talkies Amplifiers Mobile RF heaters Fluorescent lights PCs devicesAmateur radio Transmissions Ultrasound Lasers Blenders cleanersELF/VLF Line noise Welding Neon signs Vacuumnavigation machines cleanersMobile devices Cranes Medical devices Hair dryersRemote controls Refrigerators Electric shaversNatural SourcesStatic Noise Atmosphere Solar Noise Lightning Space Radio Noise domain. It is very hard to monitor the fast transient spectrum. Table 10.2 shows some continuous and transient EMI sources. An EMI problem exists only when the EMI source can exchange electromag- netic energy with its victim. The general EMI problem is shown in Figure 10.2. The coupling between the EMI source and EMI receiver (victim) can be either radiated or conducted. The energy exchange between the EMI source and victim equipment can hap- pen through the metal guides of the victim. If the primary coupling is radiated, and the EMI receiver has an input cable (which is not connected to the EMI source), the currents in the cable can appear and go directly to the victim, as shown in Figure 10.3, even if the victim is protected from radiated interference with a shield.
  • 176. 10.1 Control of Emissions and Susceptibility—Radiated and Conducted 169 Table 10.2 Sources of Continuous and Transient EMI Continuous EMI Sources— Transient EMI Sources— Frequency Domain Analysis Time Domain Analysis Broadcasting Lightning Ship radar (pulsed output) Nuclear electromagnetic pulse Electric machine noise Power line sparks Fixed and mobile communications Relays and switches PCs, printers Welding machines Solar and space radio noise Human electrostatic discharge (ESD) Radiated EMI source EMI receiver (victim) Conducted Transmitter Coupling Receiver Figure 10.2 EMI problem. Radiated EMI Shield protects radiated coupling EMI source Victim input connector I Conducted RF currents The cable works as unwanted antenna to other equipment RF EMI current into the cable Figure 10.3 Conducted EMI from radiated EMI. Similarly, if the primary coupling is conducted, and the victim has a good filter connected to the cable entrance, the currents in the cable can still radiate EM energy, which can be coupled into the victim if it does not have an appropriate shielding (Figure 10.4). These two examples show that both conducted and radiated interference (direct and indirect) must be considered to prevent EMI energy going from the EMI source to the victim equipment. It is, therefore, necessary to shield and filter the equipment simultaneously.
  • 177. 170 Emissions and Susceptibility—Radiated and Conducted Radiation from the cable Weak or no shield reaches PCB EMI source Victim I PCB RF current source Strong conducted EMI Circuits directly exposed to radiated EMI Good filter prevents conducted EMI Figure 10.4 Radiated EMI from conducted EMI. Some examples of intentional and unintentional EMI receivers are in Table 10.3. EMC is divided according to Figure 10.5. The main activities are conducted emission, conducted susceptibility, radiated emission, and radiated susceptibility. Other activities include transients [electrostatic discharge (ESD), nuclear electro- magnetic impulse, and lightning]. Electroexplosive devices and nonionizing effects of electromagnetic radiation fall into general activities of electromagnetic compati- bility and are beyond the scope of this book. Emission and susceptibility are the two most common tests of electromagnetic interference (EMI) that a device or piece of equipment should undergo, whether it operates on low or RF frequencies. Emission is the unintentional or undesired exit- ing of potentially interfering electromagnetic energy from electrical or electronic sources (devices, modules, equipment, and systems). Emission can also be inten- tional, such as from a transmitter, although it is not intended to cause interference to other devices or equipment. Emission can be conducted (carried along cables) or radiated (via propagation). Conducted emission (CE) is the potential EMI that is generated inside the equipment and is carried out of the equipment over I/O lines, control leads, or power mains. Radiated emission (RE) is the potential EMI that Table 10.3 EMI Receivers Intentional Receivers Unintentional Receivers Radio receivers Airplane control systems TV receivers Military systems, guided missiles Mobile phone receivers Ship electronic systems Microwave relay systems Computer equipment Air system receivers Signalization systems Navigation Pacemakers Radar Explosives
  • 178. 10.1 Control of Emissions and Susceptibility—Radiated and Conducted 171 EMC activities Electroexplosive Nonionizing effects of device safety electromagnetic radiation Radiated EMI Transients Conducted EMI Radiated emission Radiated sensitivity Conducted sensitivity Conducted sensitivity Cables 10 Hz–40 GHz 10 Hz–40 GHz 20 Hz–100 MHz 20 Hz–400 MHz E field 14 kHz–40 GHz 1–200 V/m (CW) antenna −20 dBm VA H field 10 Hz–30 MHz 100 MHz–40 GHz level to −110 dB V/m levels to −120 dB V/m Cable interference Human ESD Nuclear EMP lightning 15 kV transients 50 kV/m RS (10/400 ms) 10 kV transients direct and indirect 100 A harmonic distortions Figure 10.5 EMC activities. radiates from escape-coupling paths such as cables, leaky apertures, or inadequately shielded housings. Susceptibility is the characteristic of electronic equipment that permits undesir- able responses when subjected to electromagnetic energy. It is sometimes also called immunity. There are two types of susceptibility: conducted and radiated. Con- ducted susceptibility (CS) is EMI that couples from the outside of a piece of equip- ment to the inside over conductors (I/O cables, control and signal leads, or power mains). Radiated susceptibility (RS) is the undesired potential EMI that is radiated into a piece of equipment or system from a hostile outside electromagnetic source. 10.1.2 Test Requirements for Emission and Susceptibility Table 10.4 indicates the military and commercial tests that are required for the test sample. Regardless of the EMC test standard, the product must be set up in a con- trolled environment. This includes providing the test sample with standardized power to compare results from one lab to another. Commercial equipment is some- times used in a military environment. In some cases it is possible to compare mili- tary and commercial standards, and in other cases it is not. The reason is the different frequency bands specified for different tests, as well as injection methods used. Commercial standards such as EN 50081, EN 50082, IEC 60533, and IEC 945 can be compared with military standard Mil-Std-461 to some extent. For conducted emission both military and commercial standards use the Line Impedance Stabilization Network (LISN) for injection of interference. The military
  • 179. 172 Emissions and Susceptibility—Radiated and Conducted Table 10.4 Commercial and Military Tests of Emissions and Susceptibility— Radiated and Conducted Requirement Commercial Military CE Power Line (30 Hz to 10 kHz + Harmonics) Yes Yes CE Power Line (fluctuations) Yes No CE Power Line (10 kHz/150 kHz to 10 MHz/30 MHz) Yes Yes CE Antenna (10 kHz to 40 GHz) No Yes CS Power Line (30 Hz to 150 kHz) Yes Yes CS Structure CM (60 Hz to 100 kHz) No Yes CS Bulk Cable (10 kHz/150 kHz to 200 MHz/230 MHz) Yes Yes CS Bulk Cable (impulse) Yes Yes CE Cables P/S (damped sine 100 kHz to 100 MHz) No Yes RE Magnetic Field (30 Hz to 100 kHz) No Yes RE Electric Field (10 kHz/30 MHz to 18 GHz/40 GHz) Yes Yes RE Antenna (10 kHz to 40 GHz) No Yes RS Magnetic Field (30 Hz to 100 kHz) No Yes RS Electric Field (10 kHz/26 MHz to 40 GHz/1 GHz) Yes Yes RS Transient EM Field (impulse) No Yes RS ESD (up to 8 kV) Yes No Legend: CE: conducted emission; CS: conducted susceptibility; RE: radiated emission; RS: radiated susceptibility. standard (Mil-Std-461E) measures the current in the frequency range from 30 Hz to 10 kHz. In the frequency range from 10 kHz to 10 MHz, the voltage is measured. However, the commercial standards (IEC 60533 and EN 50081-2) measure only voltages between 150 kHz and 30 MHz. For radiated emission, in the harmonized commercial standards, only the elec- tric field is measured. Commercial standards require an open test site environment (OATS), while military standards require a shielded room. The cage effect is respon- sible for a deviation to open site measurement results of maximum of 6 dB. The measuring distance in the commercial standards is 30m, and sometimes 10m—only in some cases is 3m allowed. However, IEC 60533 has a distance of 3m in all cases. Mil-Std 461 has a measuring distance of only 1m, which can sometimes be a prob- lem because the antenna is placed in near field. Most commercial standards have a measuring range from 30 MHz to 1 GHz, with the exception of IEC 60533 having a starting frequency of 150 kHz. Mil-Std 461 has a frequency range from 10 KHz to 18 GHz. For conducted susceptibility commercial and military standards are not really comparable because of different injection methods. For radiated susceptibility the frequency range of commercial standards is from 30 MHz to 1 GHz (80 MHz to 2 GHz in IEC 60533), while military standards cover the frequency range from 10 kHz to 40 GHz. While military standards use nonmodulated test signals, commercial tests use a modulation of 80% with 1 kHz. In some cases like with electromagnetic pulse (EMP) where equipment should withstand 50 kV/m, only military tests exist.
  • 180. 10.1 Control of Emissions and Susceptibility—Radiated and Conducted 173 10.1.3 Standard Organizations Today’s products must conform to regulations and standards of both private and gov- ernment regulatory agencies. Their number is increasing, and updates are constantly being made. There are several international standard agencies like ISO, IEEE, IEC, ITU, and so forth. Each country has its own laws and standard organizations. Accredited testing laboratories issue certifications for products. A standard is a document established by a consensus and approved by a recog- nized body, which provides (for common and repeated use) rules, guidelines, or characteristics for activities or their results aimed at the achievement of the opti- mum degree of order in a given context. Standards cover several disciplines, dealing with all technical, economic, and social aspects of human activity and covering all basic disciplines such as language, mathematics, physics, and so forth. Standards are developed by technical committees. There are several interested parties such as: producers, users, laboratories, public authorities, and consumers. Standards are based on actual experience and lead to material results in practice (products—both goods and services, test methods, and so forth). They are a compromise between the state of the art and the economic constraints of the time. Standards are documents that are recognized as valid nationally, regionally, or internationally; they are reviewed periodically and evolve with technological and social progress. Standards are available to everyone, and can be consulted and purchased without restriction. Generally, standards are not mandatory, but voluntary. In certain cases their imple- mentation may be obligatory (e.g., safety requirements, electrical installations, pub- lic contracts, and so forth). Standards are used more and more by jurisprudence. For the user, the standard is a factor for production rationalization, which makes it pos- sible to master technical characteristics of products, satisfy the customer, validate the manufacturing methods, increase productivity, and give operators and installation technicians a feeling of security. There are four major types of standards: 1. Fundamental standards concerning terminology, metrology, conventions, signs and symbols, and so forth; 2. Test methods and analysis standards, which measure characteristics; 3. Standards defining product characteristics (product standard), specification standards (service activities standard), and standards for performance thresholds to be reached (fitness for use, interface and interchange ability, health, safety, environmental protection, standard contracts, documenta- tion accompanying products or services); 4. Organization standards dealing with the description of functions of the company and their mutual relationships, as well as the modeling of activities (quality management and assurance, maintenance, value analysis, logistics, quality management, project or systems management, production management). A national standard is programmed and studied under the authority of the national standards body, which publishes it. It is therefore protected, as early as at the draft standard stage, by a copyright belonging to the national body. Interna- tional standards are protected by a copyright of the international standards body
  • 181. 174 Emissions and Susceptibility—Radiated and Conducted (ISO, IEC). The exploitation right of this copyright is automatically transferred to the national standard bodies that are members of ISO or IEC, for the purpose of cre- ating national standards. The national standards body is obliged to take all appro- priate measures to protect the intellectual property of the ISO and IEC on national territory. The International Organization for Standardization (ISO), founded in 1947, is a worldwide federation of national standards bodies currently comprised of over 125 members—one per country. The mission of the ISO is to encourage the devel- opment of standardization and related activities in the world in order to facilitate international exchanges of goods. Its work concerns all of the fields of standardiza- tion except for electrical and electronic engineering standards, which fall within the scope of IEC. ISO counts over 2,800 technical work bodies (technical committees, subcommittees, working groups, and ad hoc groups). To date, ISO has published over 16,000 international standards. The International Electrotechnical Commission (IEC) was founded in 1906, and is responsible for international standardization in the fields of electricity, elec- tronics, and related technologies. It deals with all electrotechnologies including elec- tronics, magnetism and electromagnetism, electroaccoustics, telecommunication, energy production and distribution, as well as associated general disciplines such as terminology and symbols, measurement and performance, dependability, design and development, safety, and environment. The IEC currently has over 50 members (national committees), one for each country, which are required to be fully repre- sentative of all electrotechnical interests in the country concerned. National com- mittees are largely supported by the industry and are recognized by their respective governments. The IEC has published over 10,000 standards. Both the ISO and IEC have their central offices in Geneva, Switzerland, and operate according to similar rules. The incorporation of ISO and/or IEC standards into national collections is voluntary—it can be complete or partial. The birth of the International Telecommunication Union (ITU) can be traced back to 1865. A specialized agency of the United Nations since 1947, ITU member- ship currently includes some 180 member states and over 400 sector members. ITU international recommendations are developed in the fields of both telecommunica- tions and radiocommunications. ITU headquarters are located in Geneva, Switzerland. The Institute of Electrical and Electronics Engineers (IEEE) is the world’s largest technical professional society. It was founded in 1884; today it has over 380,000 members in more than 150 countries and has created about 2,000 standards. The IEEE Standards Association (IEEE-SA) is the newly founded organization under which all IEEE Standards Activities and programs will be carried out. In the United States there are: • ANSI—The American National Standards Institute was founded in 1918. It is the official U.S. representative to the International Organization for Standardization (ISO) and, via the U.S. National Committee, the Interna- tional Electrotechnical Commission (IEC). ANSI is also a member of the Inter- national Accreditation Forum (IAF).
  • 182. 10.1 Control of Emissions and Susceptibility—Radiated and Conducted 175 • FCC—The Federal Communications Commission is an independent United States’ government agency. The FCC was established by the Communications Act of 1934 and is in charge of regulating interstate and international commu- nications via radio, television, wire, satellite, and cable. The FCC’s jurisdic- tion covers the 50 U.S. states, the District of Columbia, and U.S. possessions. • NIST—The National Institute of Standards and Technology, founded in 1901, is a nonregulatory federal agency within the U.S. Commerce Depart- ment, which develops and promotes measurement, standards, and technology to enhance productivity, enable trade, and improve the quality of life. In the European Union there also several standard organizations: • CEN—The Comité Européen de Normalisation (European Committee for Standardization) was founded in 1961. It draws up European standards and consists of 27 European standards’ institutes. The CEN has witnessed strong development with the construction of the European Union. Its headquarters are located in Brussels, Belgium. A technical board is in charge of coordina- tion, planning, and programming of the work conducted within the work bodies (technical committees, subcommittees, working groups); the secretari- ats of which are decentralized in the different EU member states. CEN, which counts over 250 technical committees, has published several thousand docu- ments. • CENELEC—Comité Européen de Normalisation Électrotechnique (Euro- pean Committee for Electrotechnical Standardization) was founded in 1959 and is located in Brussels, Belgium. CENELEC fulfils the same functions as CEN within the electrotechnical sector. • ETSI—The European Telecommunications Standards Institute, develops European standards in the telecommunications field (ETS, European Telecom Standard). Its headquarters are at Sophia Antipolis, France. ETSI has 400 members (administrations, operators, research bodies, industrialists, users) representing over 30 countries (EU, Eastern Europe). • ECMA—The European Association for Standardizing Information and Com- munication Systems is an international, Europe-based industry association founded in 1961 and dedicated to the standardization of information and communication systems. ECMA standards and technical reports are made available to all interested persons or organizations, free of charge and copy- right, and can be obtained in printed form. • EBU—The European Broadcasting Union was created in 1950, initially with the aim of solving technical and legal problems and then to develop news and program exchanges. The result is that today the EBU assists its members in all areas of broadcasting, briefs them on developments in the audio-visual sector, provides advice and defends their interests with international bodies. Head- quartered in Geneva, Switzerland, the EBU is the world’s largest professional association of national broadcasters. Following a merger with the EBU on January 1, 1993, the International Radio and Television Organization (OIRT)—the former association of Socialist Bloc Broadcasters—expanded
  • 183. 176 Emissions and Susceptibility—Radiated and Conducted the EBU to 75 active members from 56 countries in and around Europe, and 45 associate members around the world. • CEPT—The Conference Européen des Administrations des Postes et des Télécommunications [European Post, Telephone, and Telegraph Agencies (PTT)] recommends communication specifications to the International Tele- communication Union Standardization Sector (ITU-T). In South America there are: • COPANT—The Pan American Standards Commission is a civil, nonprofit association with complete operational autonomy. The basic objectives of COPANT are to promote the development of technical standardization and related activities in its member countries with the aim of promoting the indus- trial, scientific, and technological development for the benefit of an exchange of goods and provision of services, while facilitating cooperation in the intel- lectual, scientific, and social fields. The commission coordinates the activities of all institutes of standardization in Latin American countries and develops all types of product standards, standardized test methods, terminology, and related matters. COPANT headquarters are in Buenos Aires, Argentina. • MERCOSUR—The Common Market of the South (Portuguese acronym MERCOSUL), is a common market made up of the economies of Argentina, Brazil, Paraguay, and Uruguay. Its principal objectives are to improve the economies of its member countries by making them more efficient and com- petitive, and by enlarging their markets and accelerating their economic devel- opment by means of a more efficient use of available resources Further objectives are to preserve the environment, improve communications, coordi- nate macroeconomic policies, and harmonize the different sectors of South American economies. MERCOSUR’s permanent headquarters are in the city of Montevideo, Uruguay. Each national standards body manages its own collection of standards and has access to the collections of other institutes. The collections can be either free infor- mation tools or services for identifying standards or announcing new standards. This can include catalogs, newsletters, Web servers, or chargeable services provid- ing access to the normative texts in different forms (subscription, hardcopy form, CD-ROM). National members of the ISO and IEC maintain links to related national organi- zations and, when applicable, to national standards–related networks. Information about standards can also be found in the ISO/IEC Directory of International Stan- dardizing Bodies. Normally, information on standardization and certification sys- tems, the identification of information concerning standards, and products and services is free. The publications of the standards bodies (standards, handbooks, hardcopy catalogues) are chargeable, and each body has its own tariffs. Every country has its national body for issuing its own standards or they use international ones.
  • 184. 10.2 Commercial Requirements 17710.2 Commercial Requirements The United States, European Union, Japan, Canada, Australia, and other countries have set up requirements for the emission and susceptibility of commercial equip- ment. These requirements are included in the standards given below. The FCC (Fed- eral Communications Commission) requires manufacturers of most types of electronic products to test the emission specifications. The requirements are speci- fied in the Code of Federal Regulations. Certain types of equipment require special testing by FCC. The IEC 60533 is a standard applicable for electromagnetic compatibility of electrical and electronic installations in merchant ships. Electrical installations of ships with electric and/or electronic systems need to operate under a wide range of environmental conditions. The control of undesired electromagnetic emission ensures that No other device on board is influenced by the equipment under test. On the other hand, the equipment needs to function without degradation in a normal electromagnetic environment (immunity). Special risks (e.g., lightning strikes), transients from the operation of circuit breakers, and electromagnetic radi- ation from radio transmitters are also covered. This standard also gives guidelines and recommendations on the measures to achieve EMC in electrical and electronic installations of the following equipment groups: • Group A: radio communication and navigation equipment; • Group B: power generation and conversion equipment; • Group C: equipment operating with pulsed power; • Group D: switchgear and control systems; • Group E: intercommunication and signal processing equipment; • Group F: nonelectrical items and equipment; • Group G: integrated systems. The IEC 945 (now IEC 60945) is an interference standard for navigation equip- ment installed in a ship’s environment. IEC 945 was originally produced to provide test methods and, where appropriate, limit values for electronic navigational aids. The two European EN standards are generic standards for equipment installed in an industrial environment and do not deal with product standards. EN 50081 is an emission standard and EN 80082 an immunity standard. EN 50081 and EN 50082 provide limits for emission and immunity of electromagnetic disturbances from electrical and electronic apparatus (for which there are No dedicated prod- uct-family standards) intended for use in the industrial environment. “EMC Testing and Measurement Techniques Section 3: Radiated, Radio Fre- quency, EM Field Immunity Test” (IEC 61000-4-3) has been used for many years as the basic test standard for radiated electromagnetic field immunity testing in order to fulfill one of many EU requirements for the CE mark. The IEC 61000-4-3 stan- dard is usually used together with a product standard that will specify this and other test standards, detailing the requirements the product must meet. The aim of this standard is to establish a common reference for immunity to radio frequency (RF)
  • 185. 178 Emissions and Susceptibility—Radiated and Conducted radiation caused by any source. Electronic products need to be designed and tested to have immunity from these sources. “EMC Testing and Measurement Techniques Section 6: Immunity to Con- ducted Disturbances by Radio Frequency Fields” (IEC 61000-4) relates to the con- ducted immunity requirements of electrical and electronic equipment to electromagnetic disturbances from intended radio-frequency (RF) transmitters in the frequency range of 9 kHz to 80 MHz. Equipment without at least one conduct- ing cable (such as a mains supply, signal line, or earth connection), which could cou- ple the equipment to the disturbing RF fields, is excluded. The objective of this standard is to establish a common reference for evaluating the functional immunity of electrical and electronic equipment when subjected to conducted disturbances induced by radio-frequency fields. The test method docu- mented in this part of IEC 61000 describes a consistent method to assess the immu- nity of a piece of equipment or system against a defined phenomenon.10.3 Military Requirements The military environment is different from the commercial in the areas of radar transmissions, communications in a wide frequency range, electromagnetic pulses, and inner deck situations because of high equipment density. Until the publication of 461E, MIL-STD-461 documented the test limits and levels while MIL-STD-462 specified the test methods and procedures that were to be used for conducting the tests. The E version of this standard combined both stan- dards into one document. Previous versions were A, B, C, and D. Table 10.5 shows the tests of Mil-Std 461E. Mil-Std 461 deals with two basic areas of electromagnetic effects: conducted and radiated. Each area is represented in two different modes—emission and sus- ceptibility—which include conducted emissions, conducted susceptibility, radiated emissions, and radiated susceptibility. Different devices and components such as ships, weapons, aircraft, ground and support equipment, and electrical and electronic systems that are used in a military or aerospace application are subject to meeting the requirements established in Mil-Std 461. Each branch of the armed services—Army, Navy, Air Force, and NASA—have identified specific requirements that are applicable to the specific needs and applications. Not all tests are required for every application. Mil Std 461 uses shielded enclosures for testing. The shielded enclosure should be large enough to hold the equipment being tested and be equipped to handle the requirements needed to perform simulation tests. Mil-Std 461E is free of charge and can be downloaded for free on the Internet. 10.3.1 Specific Conducted Emissions Requirements Mil-Std 461E CE101—Power Leads (30 Hz to 10 kHz) This level of low frequency testing is most applicable to the following platforms: submarines, Army, and Navy aircraft.
  • 186. 10.3 Military Requirements 179 Table 10.5 Tests of Mil-Std 461E MIL-STD-461E Specific Conducted Emissions Requirements CE101 Power Leads (30 Hz to 10 kHz) CE102 Power Leads (10 kHz to 10 MHz) CE106 Antenna Terminals (10 kHz to 40 GHz) Specific Conducted Susceptibility Requirements CS101 Power Leads (30 Hz to 150 kHz) CS103 Antenna Port-Intermodulation (15 kHz to 10 GHz) CS104 Antenna Port Rejection of Undesired Signals (30 Hz to 20 GHz) CS105 Antenna Port-Cross Modulation (30 Hz to 20 GHz) CS109 Structure Current-Spike (60 Hz to 100 kHz) CS114 Bulk Cable Injection (10 kHz to 200 MHz) CS115 Bulk Cable Injection-Impulse Excitation CS116 Damped Sinusoidal Transients (10 kHz to 100 MHz) Radiated Emissions Requirements Mil-Std 461E RE101 Magnetic Field (30 Hz to 100 kHz) RE102 Electric Field (10 kHz to 18 GHz) RE103 Antenna Spurious and Harmonic Outputs (10 kHz to 40 GHz) Radiated Susceptibility Requirements Mil-Std 461E RS101 Magnetic Field (30 kHz to 100 kHz) RS103 Electric Field (2 MHz to 40 GHz) CE102—Power Leads (10 kHz to 10 MHz) This is a similar test to CE101, but for higher frequencies. Additionally, this test has a much wider application and is required on the following platforms, systems, and subsystems: submarines; Army and Navy aircraft; air force aircraft; space systems; Army, Navy, and Air Force ground systems; and equipment surface ships. CE106—Antenna Terminals (10 kHz to 40 GHz) The CE106 testing is applicable to most antenna terminals, receivers, transmitters, and amplifiers, with the exception of equipment designed with the antenna perma- nently mounted to the equipment undergoing testing. CE106 has been recently modified to include amplifiers and is widely applicable and required on: subma- rines; Army and Navy aircraft; space systems; Army, Navy, and Air Force ground systems; and equipment surface ships. 10.3.2 Specific Conducted Susceptibility Requirements Mil-Std 461E CS101—Power Leads (30 Hz to 150 kHz) The CS101 test is applicable to equipment and subsystems of AC and DC power leads, excluding returns. If EUT is operating under DC power, testing is required only between 30 Hz and 150 kHz. If EUT is operating under AC power, testing is
  • 187. 180 Emissions and Susceptibility—Radiated and Conducted required to start at the second harmonic of the power frequency, and extends up to 150 kHz. These requirements are applicable on the following platforms: subma- rines; surface ships; space equipment and systems; Army, Navy, and Air Force air- craft; army, navy, and air force ground systems. CS103—Antenna Port-Intermodulation (15 kHz to 10 GHz) The CS103 test is required for communications receivers, RF amplifiers, transceiv- ers, radar receivers, acoustic receivers, and electronic ware receivers. The aim of this test is to control the response of the antenna connect receiving subsystems to in-band signals resulting from potential intermodulation products of two signals outside the intentional passband of the subsystems produced by the nonlinearity in the subsystem. The EUT should not exhibit any modulation beyond specified toler- ances. The CS103 test is applicable to the following platforms: submarines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems. CS104—Antenna Port Rejection of Undesired Signals (30 Hz to 20 GHz) The CS104 test controls the response of antenna connected receiving devices or sub- systems to signals outside the intentional passband produced by nonlinearity. The applications are similar to CS103, and are required for communications receivers, RF amplifiers, transceivers, radar receivers, acoustic receivers, and electronic ware receivers. CS104 can be applied to the following platforms: submarines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; Army, Navy, and Air Force ground systems. CS105—Antenna Port-Cross Modulation (30 Hz to 20 GHz) The CS105 test controls the response of antenna connected receiving subsystems to modulation being transferred from an out-of-band signal to an in-band signal. This can be caused by a strong out-of-band signal near the operating frequency of the receiver that modulates the gain in the front end of the receiver and adds amplitude varying in formation to the desired signals. The applications are similar to CS103 and CS104 and are required for communications receivers, RF amplifiers, trans- ceivers, radar receivers, acoustic receivers, and electronic ware receivers. CS105 is applicable to the following platforms: submarines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; Army, Navy, and Air Force ground systems. CS109—Structure Current-Spike (60 Hz to 100 kHz) CS109 has limited applications and is intended to simulate a spike in voltage, according to which the EUT must continue to operate without malfunction, degra- dation of performance, or deviation, even beyond the accepted tolerance range. Most applications are within the area of submarines.
  • 188. 10.3 Military Requirements 181 CS114—Bulk Cable Injection (10 kHz to 200 MHz) CS114 is widely applied to all interconnecting cables, including power cables. According to CS114, the EUT should not malfunction when subjected to a bulk injection probe drive level. CS114 is a requirement on the following platforms: sub- marines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems. CS115—Bulk Cable Injection-Impulse Excitation The CS115 test serves to protect equipment from fast rise and fall time transients that may be present due to platform switching and external transient environments, such as an electromagnetic pulse (EMP). The test will verify the ability of the EUT to withstand the impulse signals that are coupled onto the EUT associated cabling. This test replaces the old chattering relay test, referenced in Mil Std 461 C-RS 106. CS115 is applicable on the following platforms: submarines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems. CS116—Damped Sinusoidal Transients (10 kHz to 100 MHz) The concept of CS116 is to simulate electrical current and voltage waveforms occur- ring in platforms from natural resonances. CS116 is applicable to all electrical cables interfacing with the EUT and individually on each power lead. The testing should verify the EUT’s ability to withstand damped sinusoidal transients coupled onto the EUT associated cables and power leads. Power returns and neutrals need not be tested individually. The CS116 test is applicable to the following platforms: submarines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems. 10.3.3 Radiated Emissions Requirements Mil-Std 461E RE101—Magnetic Field (30 Hz to 100 kHz) The RE101 testing requirement is intended to control magnetic fields for applica- tions in which the equipment presents an installation potentially sensitive to mag- netic induction to lower frequencies. This test verifies that the magnetic field emissions from the EUT and its associated electrical interfaces do not exceed speci- fied requirements. A common example for this test is a tuned receiver that operates within the frequency range of the test parameters. The applications for this test are: submarines, surface ships, and Army and Navy aircraft. RE102—Electric Field 10 kHz to 18 GHz The RE102 test is one of the most widely required tests for electrical and electronic equipment. Its aim is to protect sensitive receivers from interference coupled through antennas associated with a receiver, and to verify that the electric field emissions from the EUT and its associated cabling do not exceed specified limits. The requirements vary depending on platform and application. The platforms
  • 189. 182 Emissions and Susceptibility—Radiated and Conducted required to meet these test parameters are: submarines; surface ships; space equip- ment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems. RE103—Antenna Spurious and Harmonic Outputs 10 kHz to 40 GHz The RE103 test is used to confirm that radiated spurious and harmonic emissions from transmitters do not exceed the specified limit requirements. RE103 has differ- ent starting frequencies depending on the actual operating frequency of the trans- mitters. The platforms required to meet this test are: submarines; surface ships; space equipment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems. 10.3.4 Radiated Susceptibility Requirements Mil-Std 461E RS101—Magnetic Field (30 kHz to 100 kHz) RS101 is primarily intended to ensure the performance of equipment potentially sensitive to low frequency magnetic fields. It is applicable to subsystems enclosures and electrical cable interfaces. It is not applicable to electromagnetic coupling via antennas. This test is required for the following platforms: submarines and Army and Navy aircraft. RS103—Electric Field (2 MHz to 40 GHz) The main purpose of RS103 is to ensure that the EUT will continue to operate with- out degradation in the presence of electromagnetic fields generated by antenna transmissions both on board and outside of the tested platform. According to RS103, the EUT should not exhibit any malfunction, degradation of performance, or deviation from the specified requirements. The requirements are applicable to equipment, subsystems enclosures, and all interconnecting cables. Most require- ments are referenced up to 18 GHz, with an optional 40-GHz range. The field strength can vary depending on the specific requirements. This is a widely used test and is required on the following platforms: submarines; surface ships; space equip- ment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and Air Force ground systems.Selected Bibliography Abrams, S., and C. R. Brown, “A Primer on Regulations and Standards,” Compliance Engineer- ing, 1998. Altay, B., and S. S. Seker, “Application Tables for MIL-STD 461D Emission Tests,” Proc. IEEE International Symposium on Electromagnetic Compatibility, August 18–22, 1997, pp. 500–503. Björklöf, D., “EMC Standards and Their Application,” Compliance Engineering, 1999. “DoD Interface Standard, Requirements for the Control of Electromagnetic Interference Emis- sions and Susceptibility,” MIL-STD-461E EMC, 1999. Dorey, P., “Gap Analysis of Military Standards for CE Marking,” Proc. EMCUK 2008, October 14–15, 2008, pp. 1–5.
  • 190. 10.3 Military Requirements 183 Klok, H. A., “Risk Analysis by the Use of Commercial Equipment in a Military Environment,” IEEE EMC Society Newsletter, Winter 2001. Smith, J., “EMI Testing for IEC 61000-4-3 Edition 3,” Microwave Journal, Vol. 50, No. 6, June 2007.
  • 191. CHAPTER 11Measurement Facilities Facilites used for measurement of susceptibility and emission, or interference include: full anechoic and semianechoic chambers, open area test sites (OATS), reverbation chambers, and transmission line structures; the two following are the most known: TEM and GTEM cells. OATS is the oldest and still the most accepted facility. Full and semianechoic chambers are mostly used for testing larger equip- ment and are generally the most expensive types of facilities. TEM and GTEM cells are mostly used for smaller equipment. The reverbation chamber can be almost any size.11.1 Full Anechoic and Semianechoic Chambers The anechoic chamber is a facility used for testing with an electromagnetic field absorbing wall, thus creating an electromagnetic-field-free environment. In acous- tics, anechoic rooms absorb sound; in RF, walls absorb electromagnetic radiation. The outer structure is the Faraday cage, which means that the interior of the room is quiet regarding RF radiation (i.e., there is no surrounding electromagnetic interfer- ence). Any radiation created inside the chamber cannot escape. For susceptibility testing, the floor must absorb the radiation (hence the name Full anechoic cham- ber), while for emission testing, the floor can be conductive (semianechoic chamber, see Figure 11.1). If the floor is removed, the chamber can be used for both types of measurements. Almost all chambers have ferrite absorber tiles, which can be used with pyrami- dal absorbers impregnated with carbon for the attenuation of radio waves. In early anechoic chambers only pyramidal absorbers were used for attenuation reflections, making the absorbers (approximately 1m long) effective only at frequencies from 100 MHz and above. Ferrite tiles are used for frequencies of 25 MHz and higher. For frequencies above 1 GHz, smaller absorbers (0.5m) are used in combination with ferrite tiles. In this way, the useful chamber range can rise up to 18 GHz. Fur- thermore, since the antennas in this frequency range have high directivity, the reflec- tions are localized, and only a part of the wall has to be covered with the absorbers. The commercial chambers should have an area with uniformity of the field less or equal to 6 dB and attenuation of 4 dB. In Table 11.1 some types of anechoic chambers are presented. Anechoic chambers can have various dimensions. For testing susceptibility and precompliance testing, the room must be large enough to allow for a distance of 3m (1m for military style applications) between the antenna and the device under test (DUT). There should also be additional 1m between the antenna, DUT, and the 185
  • 192. 186 Measurement Facilities Figure 11.1 Full anechoic and semianechoic chamber. Table 11.1 Commercial Anechoic Chamber Size Type l × w × h (m) Standard Testing Price (USD)* Smallest (26 MHz–1 GHz) 7 × 3 × 3 IEC 61000-4-3 RF susceptibility, emission $100,000 Small (26 MHz–18 GHz) 8×4×4 IEC 61000-4-3 RF susceptibility, emission $120,000 GR-1089 mm 3-m replica OATS 9 × 6 × 5.5 IEC 61000-4-3 RF susceptibility, emission, $300,000 (26 MHz–18 GHz) ANSI C63.4 EUT up to 2m GR-1089 EN 50147 5-m replica OATS 11 × 7 × 5.5 Experimental RF susceptibility, emission, $360,000 (26 MHz–18 GHz) (3m chamber) large EUT 10-m replica OATS-a 18 × 13 × 8 IEC 61000-4-3 RF susceptibility, emission, $1,100,000 (26 MHz–18 GHz) ANSI C63.4 large EUT GR-1089 EN 50147 * Turning tables, cameras, and raising the floor can increase the price for an additional $15,000 USD. room wall. The absorbing material must be placed on all six of the room walls. (For testing the emission, the floor can be movable.) Large chambers simulate 3-m or 10-m Open Area Test Sites (OATS), and must be high enough to place the antenna at 4-m height. Currently, 5m chambers (which are much cheaper), instead of the 10-m ones, are being experimented with. When building the chamber, cable filtering should be considered. If possible, optical cables should be used as well as other nonmetal interfaces. An access board with RF connectors should be as close as possible to the measurement equipment. The cables should be as short as possible. Thus, the necessary amplifiers for obtain- ing sufficient electromagnetic field levels will not have to be high in power, since they are very expensive. The cables should have as little attenuation as possible for a given frequency range. The door must be large enough for the biggest equipment, and should be sealed with gaskets and copper fingers to prevent leakage of electromagnetic fields in or out. In anechoic chambers, absorbers attenuate radio waves. It is desired that the incident wave continues traveling (i.e., to “see” the impedance, which is close to the one of free space). Such impedance has to be created, even though the metal wall impedance practically presents a short circuit. There are three methods for making
  • 193. 11.1 Full Anechoic and Semianechoic Chambers 187 the incident wave continue to travel: pyramidal absorbers, ferrite tiles, and Salisbury paper. First, only one frequency will be observed. At λ/4 from the wall, the impedance will be infinite (open end). The generator will “see” the short-circuited λ/4 trans- former as an open end. At this point, the reflected wave will be shifted by one period and added to the incident wave. If the Salisbury paper (with free space impedance of 377Ω) is placed at a distance of λ/4 from the metal wall, the metal wall will disap- pear for this narrow frequency range (Figure 11.2). For a plane wave, this effect cannot be distinguished from the wave propagating in free space. If the susceptibility testing had to be performed only at this frequency, the construction of the anechoic chamber would not be a problem. However, the chambers must be designed for a frequency range of 80 MHz to 1 GHz or more. A possible solution is shown in Figure 11.3, in which several Salisbury papers with different surface resistances are placed at λ/4 distance from the metal wall. Placing papers in such an order results in a reflection coefficient less than 0.1 at the frequency range of 2.5 to 1 around λ. Is it possible to cover the airplane with paper of a 377Ω impedance, thus mak- ing it invisible to the radar? The answer is no. The total impedance would still be 377Ω in parallel to whatever impedance is below the Salisbury paper. 11.1.1 Absorbers Pyramidal absorbers are just an expanded application of Salisbury papers. Many small reflections are created when the electromagnetic wave travels through the pyr- amid. Pyramids must be at least λ/2 long for the lowest frequency of interest (λ is even better). This is shown in Figure 11.4. Salisbury paper 377Ω Metal wall λ/4 Figure 11.2 Salisbury paper of impedance 377Ω placed at λ/4 from the metal results in the wall disappearing for the incident wave with a wavelength of λ.
  • 194. 188 Measurement Facilities R = 1565Ω R = 625Ω R = 250Ω 3 Salisbury paper Metal wall λ/4 λ/4 λ/4 Figure 11.3 Several Salisbury papers for a larger frequency range of lower reflection. µr = 1, εr = 2 − j1 λ/2 for lowest frequency Metal wall Figure 11.4 Pyramidal absorbers as a practical application of Salisbury papers. Pyramidal absorbers (Figure 11.5) in anechoic chambers are applied for suscep- tibility and emission testing as well as antenna calibration. Pyramidal absorbers are made of dense flexible foam and impregnated with carbon for obtaining the desired electrical characteristics. They are wideband and used in closed spaces. The pyrami- dal design enables multiple reflections of electromagnetic waves, while the carbon attenuates them through scattering and dissipation. The radio wave attenuation is around 45 dB.
  • 195. 11.1 Full Anechoic and Semianechoic Chambers 189 Base 60 cm Figure 11.5 Pyramidal absorber. Pyramidal absorbers are used in the frequency range of 80 MHz to 18 GHz; attenuation for a typical model is shown in Figure 11.6. 11.1.2 Ferrite Tiles Ferrite tiles (Figure 11.7) can be used in combination with pyramidal absorbers (or recently even alone) for covering walls of anechoic rooms for attenuating radio waves. Ferrite tiles are resistant to fire, moisture, and chemicals. Compared with pyramidal absorbers (> 1m), they are much smaller (6 cm). The ferrite tile imped- ance must be 377Ω (i.e., the ratio of permeability and permittivity). However, this will not stop the reflection of the wave. The ferrite must be of a complex impedance (i.e., with losses for absorbing the electromagnetic wave energy). The typical attenu- ation for a ferrite tile of 1-cm width at 100 MHz is 11 dB, amounting to 22 dB in total (tile attenuates both the incident and reflecting wave). Attenuation (dB) 40 35 30 25 20 15 10 5 0,01 0,10 1,00 10,00 100,00 f (GHz) Figure 11.6 Attenuation of a pyramidal absorber.
  • 196. 190 Measurement Facilities Metal wall Ferrite Dielectric Figure 11.7 Ferrite tile. Figure 11.8 shows the basic principles of an electromagnetic absorber. When the electromagnetic wave travels through free space and reaches a medium of differ- ent characteristics, the wave will partially reflect and partially absorb. The reflected wave is more important. The ferrite tile thickness is chosen in such a way that the relative phase of the reflected and transmitted wave cancel each other out and create x Reflected Er Et wave Hr Ht z y y Ei Transmitted Incident wave wave Hi Material 1 Material 2 z=0 Metal Figure 11.8 Incident, reflected, and transmitted waves.
  • 197. 11.2 Open Area Test Site (OATS) 191 a resonant state. It is a function that depends on the electric characteristics of ferrite materials such as relative permeability (µr) and permittivity (εr), which determine the reflection coefficient, impedance, and return loss according to the following expression and Table 11.2: ⎡⎛ j2 πd ⎞ ⎤ ( ) µr Zf = ⋅ tanh ⎢⎜ ⎟ µr εr ⎥ Ω (11.1) εr ⎣⎝ λ ⎠ ⎦ Figure 11.9 shows the attenuation of ferrite tiles depending on frequency. It is clear that ferrite tiles are best to use at frequencies from 10 MHz to several hundred megahertz.11.2 Open Area Test Site (OATS) OATS is the oldest and still the most accepted test facility for acceptance of results. Compared to anechoic chambers, it is much cheaper to build. OATS should be placed close to the production site, but in a quiet RF surrounding. These two differ- ent requirements are often opposite. Furthermore, when selecting a location, inter- national standards need to be checked. It is desirable to use numerical modeling methods before building. Table 11.2 Magnetic Characteristics Permeability µr 2,100 Curie temperature Tc > 95°C 6 Resistance ρ 5 · 10 Ωcm 3 Specific density 5.2 g/cm −6 Linear coefficient 9 · 10 /ºC 0 10 20 A (dB) 30 40 50 10 100 1000 f (MHz) Figure 11.9 Ferrite tile attenuation depending on frequency.
  • 198. 192 Measurement Facilities Figure 11.10 shows the schematics of OATS. Distance F depends on the test condition. The antenna should be able to move vertically in order to find the stron- gest signal (due to reflections from the ground). The ellipse dimension can be 3, 10, or 30m. The space above the ellipse should be free, without reflecting surfaces. In case of precipitation, a dielectric roof is allowed. The emission from the equipment under test (EUT) has to be measured with an appropriate receiver and antenna. Figure 11.11 shows the position of EUT and antenna. The antenna should be able to move vertically 1m to 4m. The turntable must be able to turn 180°. The dis- tance between EUT and the antenna can be 3m or 10m. Ferrites are placed on the cable going to the spectrum analyzer or test receiver. Perfect OATS should have an infinite ground without metal objects in the vicin- ity (fence, power cables, and so forth). To be able to use OATS in all weather condi- tions, it is desirable to build a protection roof resistant to weather conditions. The walls and roof should be made of dielectric materials, since they have less impact on higher frequencies. 2F Receiving antenna Movable Receiver 1.73 F EUT F Ellipse boundary Figure 11.10 Schematics of an open area test site. 3 or 10 m Ferrites on cable EUT 1–4m Turntable 80 cm Ground plane Figure 11.11 Placement of EUT and the antenna.
  • 199. 11.3 Reverberation Chamber 193 The reflection coefficient from the vertical incident wave to the thin wall is: πl( ε r − 1) r = (11.2) λf where l is the wall thickness (m), εr is the permittivity of the wall, and λf is the wave- length in free space. If the reflection coefficient is to be less than 0.1 (typical value) at 1 GHz (λf = 0.3m), then the acceptable width l for a given, nonferrite, nonconductible wall is: 03 ⋅ 01 . . 001 . l= ≈ (11.3) π( ε r − 1) ε r − 1 The basic modular design of OATS is shown in Figure 11.12. The structure is raised from the ground. When making a study, the attenuation of OATS can be affected by the follow- ing parameters: • Ground size; • Boundary conditions of the ground and surrounding terrain; • Influence of moisture in different types of soil; • Conductivity of soil as a function of temperature. The size of the EUT can be up to 2F/λ (Rayleigh criteria). Surface irregularities must be within ± 20 mm. The antenna should be placed at a 1–4-m height for obtaining the highest field level. It is also necessary to perform detailed ambient field measurements before building the OATS.11.3 Reverberation Chamber The reverberation chamber (Figure 11.13) is a relatively new type of facility for test- ing emission and susceptibility. It consists of a plain shielded chamber with low loss walls. It should not radiate outside, does not contain absorbers, and can be of any size. On resonant frequencies, the reverberation chambers are resonators with a large Q factor. Inside the chambers mode tuners are built in, which change bound- ary conditions inside the chamber, thus ensuring that the EUT is exposed to a full energy amount and that all of the EUT emissions can be measured. Statistically, uni- form and isotropic wave propagation with uniformly distributed polarization occurs. Testing in a reverberation chamber can be performed on frequencies above Ground Ground Figure 11.12 Raised ground—side look.
  • 200. 194 Measurement Facilities Figure 11.13 Reverberation chamber. the cutoff frequency (i.e., in the area where modes can exist inside the chamber). The Q factor can be calculated from 3 V Q= ⋅ (11.4) 2 Sδ where V is the chamber volume, S is the surface of inner walls, and δ is the wall thickness (skin effect): 1 δ= (11.5) πfσµ 3 Reverberation chambers are usually from 75 to 100 m , although they can be much smaller. They are used for testing at frequencies higher than 200 MHz (up to 18 GHz). Working with frequencies below 200 MHz requires very large rooms. For frequencies above 1 GHz, smaller chambers can be used (volume ~ 0.25 m3). The shape of the chamber is irrelevant—different designs prove quite good. The volume is the key factor. Around 50% of the volume is useful for testing, which is more than with other types of chambers. Mode tuners or propellers (Figure 11.14) are made of four equal boards turning around a vertical axis. The turning of the tuners is regulated by step motors and microcontrollers, which must be placed outside of the room. Step motors and microcontrollers should not conductively be coupled with the interior of the chamber. Reverberation chambers can only measure the isotropic radiation of EUT and not the electrical field at a certain distance, which is often required in international standards. The price of a reverberation chamber (6.55m × 5.85m × 3.50m) with the lowest usable frequency of 124 MHz is about $50,000.
  • 201. 11.4 TEM Cell 195 Figure 11.14 Mode tuners.11.4 TEM Cell The appearance of the TEM cell (or Crawford cell) in 1974 intensified testing in the fields of electromagnetic compatibility, biomedical effects, and electromagnetic dis- turbances. The first TEM cell, still in use, underwent many improvements. The TEM cell is a simple and economical surrogate for Open Area Test Systems. The test system should have an area with no outside interference, and at the same time a uni- form electromagnetic field inside. With increasing frequency, the cell dimensions get smaller, thus becoming too small for testing larger devices, except for possibly printed circuit boards. The goal of technical improvements to various cell types is increasing useful test area for higher frequencies using absorbers, obtaining a field as uniform as possible, and avoiding the appearance of higher-order modes of prop- agation. This is not a simple task and requires a lot of numerical modeling, use of various numerical methods, and testing of prototypes. In the meantime, TEM-cells have been acknowledged as standardized test systems for electromagnetic compati- bility and interference testing. Research is ongoing; new absorption materials are being developed and new geometrical structures tried. Transversal electromagnetic (TEM) transmission cells are devices used for establishing uniform electromagnetic fields in a shielded environment. They are structures with three closed transmission lines for preventing radio frequency radia- tion and electrical isolation. The TEM cell is made of a quadrature transmission line with pyramidal parts at the ends for adapting to standard coaxial connectors (Figure 11.15). A uniform TEM field is established inside the cell on any desired frequency below the cutoff frequency, at which higher-order modes start to appear. TEM cells are used for testing small equipment, calibrating of radio frequency probes, and bio- medical experiments. The wave propagating through the cell has a wave impedance equal to free-space impedance (377Ω), thus enabling a good approximation of a planar wave in a far field. The cells are wide bandwidth and have linear phase and amplitude frequency characteristics from the DC to the cell cutoff frequency. This feature enables testing with a continuous wave (CW) or over a selected frequency range, as well as with a
  • 202. 196 Measurement Facilities Septum Outer shield Coaxial termination Coaxial connector RF source EUT Figure 11.15 TEM cell. pulse or modulated signal. The cell has its limitations—the main being that the upper cutoff frequency is determined solely by the physical dimensions of the cell. This results in limitations on the DUT size. The expression for obtaining the electrical field in the cell, where V is the volt- age on the septum, b/2 is the distance from the septum to the cell wall, P is the power level introduced into the cell, and Z0 is the characteristic impedance of the cell, is defined as follows: V PZ 0 E= = (11.6) b2 b2 11.4.1 Characteristic Impedance Characteristic impedance of the symmetrical stripline, which has a metal shield at the sides (Figure 11.16), is given with the values of cross section dimensions and unknown edge capacitance over the unit length Cf´: 3766 . Z0 = (11.7) [ ] 4 w (b − t ) + C f′ ε −12 where ε0 = 8.852 · 10 , with an air dielectric. For a central conductor with small thickness, Cf 2b ⎡ ⎛ π a − w⎞⎤ t = ln 1 + coth ⎜ ⎟ + K (11.8) ε π(b − t ) ⎢⎣ ⎝ 2 b − t ⎠⎥ a − w ⎦ a g g w b/2 t b/2 Figure 11.16 TEM cell cross section.
  • 203. 11.4 TEM Cell 197 The upper expression is valid for (a w)/2b < 0.4. The Crawford cell (and other TEM cells) is designed to have a characteristic impedance of 52Ω. This value is chosen because when inserting a DUT, the charac- teristic impedance will drop slightly. 11.4.2 Higher-Order Modes The basic restriction of the TEM cell is the appearance of resonances, which destroy the uniform field distribution of the TEM mode of wave propagation. There are numerical methods for establishing the cutoff frequency of the higher-order modes as a function of septum (inner conductor) width. The determination of the resonant cell length is not simple, because the cell pyramidal parts are acting differently at each higher-order mode. The TEM cell is a resonant cavity with a high Q, where the higher-order modes tend to appear at exactly determined frequencies. There is a window between these resonances where the use of the TEM cell is still possible. To which degree these structures can be used with the presence of higher-order modes and whether it is possible to use them between these resonances will depend on the practical implementation for which the cell is desired. Generally, the cutoff fre- quency in a perpendicular waveguide for TE10 mode, which is usually the first higher-order mode that starts to propagate, is given with the following expression: c f c (TE10 ) = (11.9) 2a where c is the speed of light. The expression for the cutoff frequency for any higher-order mode TEmn is as follows: ( ) 12 c b 2 m2 + a 2 n 2 f c (TE m , n ) = (11.10) 2ba where a and b are the dimensions of the waveguide cross section and m and n are the number of half-periods of the electric field in the x- and y-axes. The TEM mode propagates through pyramidal parts of the cell without any sig- nificant change. Every higher-order mode is always reflected at the same place of the pyramidal part until it becomes too small to propagate. The energy of propagation of the higher-order mode suffers from repeated reflections inside the cell until it exhausts itself. The resonant conditions are fulfilled when the effective cell length for a particular mode is equal to the number of half-lengths (p = 1, 2, 3, ...), p being the number of half wavelengths. On resonant frequencies, fR(mnp), there is a resonant field of the TEmnp mode. If we use the expression: l ( mn ) = pλ g ( mn ) 2 ; p = 1, 2, 3 (11.11) and 1 1 1 2 = 2 + 2 (11.12) λ λg λ c ( mn )
  • 204. 198 Measurement Facilities where λ2c (cm) represents the value of wavelength at the cutoff frequency, the follow- ing expression, which predicts various resonant frequencies, can be obtained: 2 ⎛ pc ⎞ f 2 =f 2 +⎜ ⎟ (11.13) R ( mnp ) c ( mn ) ⎜ 2l ⎟ ⎝ ( mn ) ⎠ with f c (mn ) = c/ λ2c (mn ) . Spreading of the useful frequency range of the cell can be achieved by filling the cell with absorbers. It will lessen the quality factor (Q) inside the cell, which is fre- quency dependent. The absorbers improve the uniformity of the field between the septum and upper and lower walls, thus increasing the vertical component of the electric field on edges of the septum. Higher-order modes have a special influence on the TEM cells. Their appear- ance disrupts the uniformity of the electromagnetic fields inside the cell. The first higher-order mode depends solely on cell dimensions. After the first higher-order mode, the other higher-order modes starts to appear. The cell can be used even after higher-order modes start to appear. In the following text, a calculation of the higher-order modes according to experimental equations is shown. Higher-order modes have two characteristic frequencies: cutoff and resonant. For every mode, the expression for the cutoff frequency is as follows: x ⎛c⎞ 150 x f c ( m, n) = ⎜ ⎟ 2π ⎝ b⎠ [Hz] = π b [MHz] (11.14) 8 where c = 3 × 10 m/s, and where x depends on the mode and is given later. The appropriate resonant frequency is determined by: 2 ⎛ pc ⎞ f 2 R ( m, n, p) =f 2 c ( m, n) +⎜ ⎟ (11.15) ⎝ 2 L mn ⎠ where Lmn is the effective cell length for every mode, L mn = L c + X mn L E (11.16) and Lc is the length of the central section of the cell, and LE is the length of the two pyramidal ends. Xmn is an empirically defined multiplier. Table 11.3 gives the equa- tions for calculating cutoff frequencies of higher-order modes. 11.4.3 TEM Cell Construction The TEM cell (like most other cells) should be designed to have a characteristic impedance of 52Ω. When the TEM cell was designed at the Faculty of Electrical Engineering and Computing, Zagreb (FER), its future purpose was taken into con- sideration (biomedical experiments, probe calibration, and electromagnetic com- patibility). The aim was to achieve as much space as possible for testing at the frequency of 900 MHz, and sustain the characteristic impedance of the cell of 50Ω
  • 205. 11.4 TEM Cell 199 Table 11.3 Expressions for Calculating the Cutoff Frequencies of Higher-Order Modes Mode Expressions −1 TE01 π ⎛ b ⎞ ⎡ ⎛ 2a ⎞ ⎤ x tan x = ⎜ ⎟ ⎢ ln ⎜ ⎟ + RTE 01 ⎥ 2 ⎝ a ⎠ ⎣ ⎝ πg ⎠ ⎦ ∞ 1⎛ pπ b ⎞ ⎛ pπ g ⎞ RTE 01 = ∑ p ⎜ coth ⎝ a − 1⎟ cos 2 ⎜ ⎠ ⎝ a ⎠ ⎟ p =1 TE10 π ⎛ b⎞ x= ⎜ ⎟ 2 ⎝a⎠ TE11 ⎡ 2 ⎛ πb ⎞ 2 ⎤ x= ⎢y + ⎜ ⎟ ⎥ ⎢ ⎝ 2a ⎠ ⎥ ⎣ ⎦ −1 π b ⎡ ⎛ 8a ⎞ 2 ⎛ πg ⎞ ⎤ y tan y = ln ⎜ ⎟ − 2 cos ⎜ ⎟ + RTE 11 ⎥ ⎛ πg ⎞ a ⎢ ⎝ πg ⎠ ⎝ 2a ⎠ cos 2 ⎜ ⎟ ⎣ ⎦ ⎝ 2a ⎠ RTE 11 = 2 ∑ ∞ 1 ⎛ ⎜ coth (2 p + 1)πb − 1⎞ cos 2 ⎛ 2 p + 1 πg ⎞ ⎟ ⎜ ⎟ p =1 2p + 1⎜ ⎝ 2a ⎟ ⎠ ⎝ 2a ⎠ TE02 x=π TE12, TM12 ⎡ ⎛ b⎞ ⎤ 2 x = π ⎢1 + ⎜ ⎟ ⎥ ⎢ ⎝ a2 ⎠ ⎥ ⎣ ⎦ TE20 b x=π a TE21 ⎛ πg ⎞ 2 cos 2 ⎜ ⎟ cot y cot x ⎝ a ⎠ 2 a ⎡ ⎛ 2a ⎞ 2 ⎛ πg ⎞ ⎤ + = ⎢ ln ⎜ ⎟ − cos ⎜ ⎟ + RTE 21 ⎥ x y π b ⎣ ⎝ πg ⎠ ⎝ a ⎠ ⎦ 1 ⎡ ⎛ b⎞ ⎤2 2 y = ⎢x2 − ⎜ π ⎟ ⎥ ⎢ ⎝ a⎠ ⎥ ⎣ ⎦ ∞ 1⎛ pπ b ⎞ ⎛ pπ b ⎞ RTE 21 = ∑ ⎜ coth − 1⎟ cos 2 ⎜ ⎟ p=2 p⎝ a ⎠ ⎝ a ⎠ 1 TM11 ⎡ ⎛ πb ⎞ ⎤ 2 2 x = ⎢ y2 + ⎜ ⎟ ⎥ ⎢ ⎝ 2a ⎠ ⎥ ⎣ ⎦ −1 tan y 2 a ⎡ ⎛ 2 a ⎞ ⎤ 2 = ⎢⎜ ⎟ + 1 − RTM 11 ⎥ y π b ⎢ ⎝ πg ⎠ ⎣ ⎥ ⎦ 4a ∞ ⎡ (2 p + 1) + b − 1⎤ J ⎡(2 p + 1)πg ⎤ RTM 11 = ∑ ⎢ coth 2 a πg p = 1 ⎣ ⎥ 1⎢ 2a ⎥ ⎦ ⎣ ⎦ at the same time, which was difficult to achieve. The usual cell dimensions are a > b (i.e., the cell is wider than it is tall). The value a < b was chosen, which provided for more space in the vertical dimension. This resulted in the change of the characteris- tic impedance from 50Ω to 75Ω. Even though most network analyzers and signal generators operate at 50Ω, the adaptation can be achieved through a 50/75Ω trans- former. Figure 11.17 shows the blueprints for the TEM cell designed at FER.
  • 206. 200 Measurement Facilities 12.5 cm 25 cm 12.5 cm BNC 25 cm connector Door Guide Septum 30 cm holes 3.5 cm gap 25 cm 18 cm Dielectric supporters Figure 11.17 Scheme of the TEM cell. The size of the device that can be tested in the cell is 5 cm high, which is 1/3 of the total TEM cell height. The cell is made of aluminum, whereas the septum is made of 1.5-mm-thick copper; any metal available is acceptable. The connectors are BNC, and from the sides there are doors to insert the DUT. There are openings for the wires below the doors. The septum is supported with dielectric material (Tef- lon). The cell can withhold up to 50W without cooling. Since it is made of alumi- num, it is very light for carrying and handling. 11.4.4 Parameter Measurements The TEM cell was tested at the Department for Radiocommunications with Net- work Analyzer HP 8620B. VSWR (Figure 11.18) transmission characteristics (Fig- ure 11.19) were measured and the Smith chart (Figure 11.20) was obtained as a result. Transformers 50/75Ω were used for adaptation. At the frequency of 935 MHz, VSWR was measured to be only 1.06 and absorption 4.5 dB. Characteristic imped- ance was 80.8 − j0.92Ω, which should be higher than 75Ω because it will drop once the DUT is inserted in the cell. To ensure the field inside the cell, a probe is neces- sary. This will be discussed in the following chapters. Figure 11.21 shows the photo- graph of the TEM cell built at FER.
  • 207. 11.5 GTEM Cell 201 3.5 3 2.5 VSWR 2 1.5 1 100 200 300 400 500 600 700 800 900 Frequency (MHz) Figure 11.18 VSWR of the TEM cell. 8 7 Absorption (dB) 6 5 4 3 2 100 200 300 400 500 600 700 800 900 Frequency (MHz) Figure 11.19 Absorption of the TEM cell.11.5 GTEM Cell The GTEM cell is a transmission line based on the TEM cell approach. The letter G stands for gigahertz, since the GTEM cell operates from DC to 18 GHz. The slightly curved wave front (not a planar wave) travels from the source to the 50Ω quadratic shielded transmission line (radial type) to the hybrid termination without geometrical distortion of the TEM wave. This transmission line can be sym- metric or nonsymmetric (the latter being more frequent), in order to obtain a more useful testing area. Symmetrical transmission lines are also called coaxial. Since the waveguide incident angle is small (20°), the wave can be considered planar. The GTEM cell is an adaptable (pyramidal) part of a quadratic transmission line (TEM cell) with a characteristic impedance of 50Ω. The GTEM cell (Figure 11.22) starts with a precise apex, where the transition from the standard 50Ω N-type connector to a nonsymmetrical quadratic waveguide is done. The distributed load consists of absorption material used for the termination of the electromagnetic wave, and of distributed resistance used for terminating low frequency currents. On low frequencies, the cell has an impedance of 50Ω. At higher frequencies, the absorber attenuates the incident wave in much the same way as in the anechoic chamber. Thus, the matching from DC to several gigahertz is achieved. Wideband performance, which is enabled by termination load, lowers the influ- ence of higher-order modes. The absorbers significantly reduce the quality factor of
  • 208. 202 Measurement Facilities Figure 11.20 Smith chart of the TEM cell. Figure 11.21 TEM cell developed at FER.
  • 209. 11.5 GTEM Cell 203 Figure 11.22 GTEM cell. the cell, thus lessening the influence of resonances. The TEM mode generated with a continuous source or pulse generator simulates the planar wave for testing emission and susceptibility. 11.5.1 GTEM Cell Characteristics GTEM cell characteristics (Figure 11.23) are: • Characteristic impedance of 50Ω; • Septum at 3/4 of the cell height (for larger EUT); • Width/height ratio of 2/3; • 15° angle between the septum and the lower shield; • 5° angle between the septum and the upper shield. The N-type connector is placed at the end of the pyramidal part. The septum is supported with dielectric material. On the other side of the cell, there is a distributed termination. The DUT size can be 1/3 of the size between the septum and shield. 11.5.2 GTEM Cell Construction Figure 11.24 shows the schematic of the GTEM cell built at FER, Zagreb. The sep- tum, as well as the shielding, is made of copper. The N-type connector is placed at the beginning of the pyramidal end. 2w g g 2b 2a Figure 11.23 GTEM cell cross section.
  • 210. 204 Measurement Facilities Door Septum 20×20cm 38,5 cm N-type connector 115 cm 60 cm Figure 11.24 GTEM cell blueprint. Dielectric supporters of the septum are made of Teflon. On the other side of the cell there are pyramidal absorbers of 25 cm for matching electromagnetic waves and two parallel 100Ω resistors for current termination. The EUT size is 20 cm × 20 cm. The cell is designed in such a way as to enable replacement of pyramidal absorb- ers with ferrite (or some other more efficient absorbers) in the future. In this way, more testing area is obtained. The N-type connector can also be easily replaced in case of damage. The resistance array of 100Ω can also be replaced, or instead of two 100Ω resistors some other combination of resistors can be introduced (6 × 300Ω, for example). Figure 11.25 shows the GTEM cell cross section. The outer measures are slightly different from the inner due to the side connecting on edges. The sides are connected with silver. 11.5.3 GTEM Cell Parameter Measurement The GTEM cell must be tested for its voltage standing wave ratio (VSWR), trans- mission characteristics (reflection), and time-domain measurements. 10 mm 10 mm 600 mm inner 400 mm 620 mm Figure 11.25 Cross section of GTEM cell.
  • 211. 11.5 GTEM Cell 205 Figures 11.26 to 11.29 show the VSWR and reflection from 1 GHz to 20 GHz. The resistors have more influence on lower frequencies, but their influence weakens greatly after 100 MHz. After 500 MHz their influence is negligible and absorbers start dominating above that frequency. Figure 11.30 shows the time-domain response of the GTEM cell. The higher magnitude levels are due to reflections at the connector and dielectric supporters along the stripline. Measuring Electric Field Strength Inside the Cell In the FER project the measuring of the electric field was carried out with the radio frequency signal generator HP 8656A (0.1–1,040 MHz), amplifier MiniCircuits 28 dB (100–900 MHz), probe Holaday HI-4455, and readout device HI-4460. HI-4460 is a graphical device for reading the values of electromagnetic fields, and it has a screen made of crystals for displaying numerical and graphical values. The device can be connected to a computer through a RS232 interface. Probe HI-4455 is a battery-operated wide-bandwidth isotropic probe for measuring the electric field in the vicinity of the RF source. The application includes measuring microwave transceivers and antennas, and monitoring electromagnetic interference (EMI). The probe uses optical isolation for keeping field changes low during measurements and has a conical casing and sensor inside. The sensor is placed on one end of the sup- port, while the other part is connected to the electronics. With three orthogonally placed dipole antennas, the probe measures the field intensity in three directions, calculates the sum, and sends the results to the receiver over an optical cable. The frequency response is from 200 kHz to 40 GHz, whereas the dynamical range is from 1.5 to 300 V/m. 2.00 1.80 1.60 VSWR 1.40 1.20 1.00 0.20 0.40 0.60 0.80 1.00 f (GHz) Figure 11.26 VSWR up to 1 GHz of the GTEM cell. 3.50 3.00 2.50 VSWR 2.00 1.50 1.00 0.00 5.00 10.00 15.00 20.00 f (GHz) Figure 11.27 VSWR up to 20 GHz of the GTEM cell.
  • 212. 206 Measurement Facilities 0.20 0.40 0.60 0.80 1.00 0 −5 Reflection (dB) −10 −15 −20 −25 −30 f (GHz) Figure 11.28 Reflection up to 1 GHz of the GTEM cell. 0.00 5.00 10.00 15.00 20.00 0 −5 Reflection (dB) −10 −15 −20 −25 −30 −35 f (GHz) Figure 11.29 Reflection up to 20 GHz of the GTEM cell. 0.14 0.12 0.10 Magnitude 0.08 0.06 0.04 0.02 0.00 0.0 0.5 1.5 1.0 2.0 2.5 3.0 3.5 t (ns) Figure 11.30 Time-domain response of the GTEM cell. Figure 11.31 shows the field distribution inside the GTEM-cell at 1/2 of the sep- tum height—100 MHz. It is obvious that the measurements are in concordance with the modeled values (FEM method). The field was measured at different locations: the middle and at the edges of the cell. Figure 11.32 shows the frequency response of the GTEM cell in the middle of the area reserved for testing with an input power of 40 dBm. It is obvious that the results are within 3 dB.
  • 213. 11.5 GTEM Cell 207 5.2 5.1 5.0 4.9 4.8 Measured 4.7 Num. method 4.6 4.5 4.4 4.3 4.2 0.0 0.1 0.2 0.3 0.4 0.5 0.6 Figure 11.31 Field distribution inside the GTEM cell at 1/2 of the septum height (100 MHz). 1000 E (db/(V/m) 100 10 1 100 1000 f (MHz) Figure 11.32 Measured electric field strength inside the GTEM cell with an input power of 40 dBm. Higher-Order Modes Higher-order modes do not play such an important role with GTEM cells as with TEM cells, because the GTEM cell is not a high Q factor cell. Higher-order modes are attenuated and therefore hard to measure. There are analytical calculation equa- tions of higher-order modes, which are valid only for the first several modes since the absorbers have a high influence at higher frequencies. Higher-order modes appear due to dimensions change, nonuniform mediums of absorbers, and finite conductance of the walls. In the vicinity of the N connector higher-order modes can- not propagate, but when moving along the septum axis, the possibility of their appearance increases. The modes stimulated by the TEM mode are called essential. Other modes, stimulated by disturbances or small discontinuities, do not achieve large amplitudes and are called nonessential. The first several modes that can propagate in the GTEM cell are H01(TE01), H10, H11, and H20, and after them E11 and E22. Even though mode H01 is the first to start propagating, the first essential mode is H10. (Regarding resonances in the GTEM cell, it is the most important one.) In nonuniform waveguides, transversal fields are the functions of the z coordi- nate (i.e., the direction of wave propagation). They can be expressed with the fol- lowing vector functions:
  • 214. 208 Measurement Facilities ∞ r Htr ( x , y, z ) = ∑ I ( z )h ( x , y, z ) i i i =1 (11.17) r∞ Etr ( x , y, z ) = ∑ Vi ( z )e i ( x , y, z ) i =1 r r The vectors h i and e i for TE and TM modes can be expressed as: r (E) (E) r (E) r (E) e = −∇ tr T h = −n z × ∇ tr T (11.18) r (H ) r (H ) r (H ) (H ) e = −n 2 × ∇ tr T h = −∇ tr T (11.19) (E) (H) T and T fulfill differential equations and boundary conditions: (E) (E)2 (E) ∇ tr T 2 + kc T =0 (11.20) = 0 on L( z ) (E) T (H ) (H )2 ( HE ) ∇ tr T 2 + kc T =0 (H ) (11.21) ∂T = 0 on L( z ) ∂n where L is the boundary curve of the cross section. Putting (11.18) into the Maxwell equations, the following system of differential equations is obtained: dVk ∞ = − γ k ( z )Z Lk ( z )I k ( z ) + ∑ C ki ( z )Vi ( z ) dz i =1 (11.22) dI k γ k (z) ∞ =− V k ( z ) − ∑ C ki ( z ) I i ( z ) dz Z Lk ( z ) i =1 The nonuniform waveguide can be regarded as the system of coupled transmis- sion lines, where the coupling factors Cik are the functions of z. Field strengths can be expressed with voltages Vk and currents Ik. This equations system is known as telegraphic equations. The characteristic impedance values of ZL and spreading constant γ are given in Table 11.4. Separating one equation from the other (11.22), the Schroedinger equation for calculating higher-order modes is obtained: (E) d 2 Ik dz 2 − kc ( (E)2 ( z ) − k 2 )I kE ) ( =0 (11.23) (H ) d 2 Vk dz 2 − kc ( (H )2 ( z ) − k 2 )Vk( H ) =0 (11.24) for E (TM) modes (11.23) and H (TE) modes (11.24), where Vk and Ik are equivalent voltages and currents of the electromagnetic field inside the waveguide.
  • 215. 11.5 GTEM Cell 209 Table 11.4 Values of ZL and γ for TE and TM Modes TE Modes TM Modes (E ) (z) = ( E )2 (z) − k γ( ) (z) = ( ( z ) − k2 2 H H )2 γ kc kc 1 1 ( ) k2 − kc ) ( z ) (E 2 E ZL = ( ZL H ) = ωµ ( (z) ωε H )2 k2 − kc 1 ( ) ( z )Z )( z ) = jωε (H ) (H γ( ) ( z )ZL )( z ) = ( kc ) ( z ) − k2 (E 2 γ E E L jωε γ( ( z ) = jωε ) γ( ) ( z ) = 1 k( H )2 z − k2 E H ZL ( z ) (E ) ZL (H ) ( z ) jωε (c () ) The propagation constant in every cross section can be expressed as γ k (z) = kc ( z ) − k 2 2 (11.25) ⎧ real k 2 < kc 2 (E,H ) ⎪ γ is ⎨ 0 k 2 = kc2 (11.26) ⎪ imaginary k 2 < k 2 ⎩ c If γ is real, the mode will attenuate and (11.23) and (11.24) describe the wave that is attenuated. If γ is imaginary, the mode is above the cutoff frequency and Schroedinger equations describe the propagating wave. Local cutoff frequencies can be calculated according to the following expression: kc kc c fc = = (11.27) 2 π εµ 2π where kc is the wave number. For most TEM cells, the product of the wave number and the width of the cell are constant: kc ⋅ a = konst (11.28) By solving (11.23) and (11.24), the expression for resonant frequencies of the higher modes is obtained. For the GTEM cell, where kca = konst, the cross section of the cell changes linearly in the direction of wave propagation. The boundary curve can be described with a( z ) = mz + a( z = 0) (11.29) Using (11.25) the following is obtained: ⎛ ⎞ d 2 V ⎜ ( konst 2 ) 2 − − k2 ⎟ V = 0 (11.30) ⎜ ⎟ dz 2 ⎝ (mz + a(0)) 2 ⎠ where konst is different for every mode. If a replacement is introduced:
  • 216. 210 Measurement Facilities d2 d2 (3 4)b a = mz + b ⇒ 2 = m2 2 m= (11.31) dz da l Equation (11.31) takes a special form of the Bessel differential equation d 2 V ⎛ C1 ⎞ 2 2 − ⎜ 2 − C2 ⎟V = 0 2 (11.32) dz ⎝α ⎠ where ⎛ k 2 a⎞ k2 C1 = ⎜ c ⎟ i C 2 = 2 2 2 (11.33) ⎝ 2m ⎠ m By solving the Schroedinger equation, the following is obtained: ⎛ b ⎞⎞ 1 ⎛ kc a ⎞ V (H ) = A1 ( mz + b ) J v ⎜ k⎛ z + ⎜ ⎝ ⎟⎟ ; v = +⎜ ⎟ (11.34) ⎝ m⎠ ⎠ 4 ⎝ 2m ⎠ where m is the ratio of opening b and cell length l. Jv are the zero points of Bessel functions and constant A1 can be obtained by solving the telegraphic equations. Figure 11.33 shows the amplitudes of the first, second, and third resonance in the GTEM cell (made at FER) obtained by a specially developed software. For a GTEM cell, the following is valid: • Opening height in regard to width, b/a = 2/3; • Septum height of 3/4b; • Septum thickness in regard to opening height t/b = 0.004; • Septum width in regard to opening width w/a = 0.65; • Cell length l. The values of kca are given in Table 11.5. By solving (11.34) for mode H10, three resonant frequencies are obtained, stim- (H) ulated by the cutoff frequency of that particular mode at 249.614 MHz: f101 = (H) (H) 406.01 MHz; f102 = 558.28 MHz; f103 = 602.34 MHz. H01, which is the first Table 11.5 Values of kca Depending on the Propagation Mode MOD kc a H01 2.640 H10 3.138 H11 5.418 H20 6.281 E11 6.002 E21 8.699
  • 217. 11.5 GTEM Cell 211 0.20 f r1 0.15 f r2 f r3 0.10 0.05 0.00 −0.05 −0.10 −1.2 −1.0 −0.8 −0.6 −0.4 −0.2 0.0 z Figure 11.33 H01 amplitudes of the first, second, and third resonances in the GTEM cell. propagating mode (but actually the second mode), has the cutoff frequency of (H) 210.085 MHz, from which the following resonant frequencies appear: f011 = (H) (H) 362.03 MHz; f012 = 509.16 MHz; f013 = 650.21 MHz. For other modes, the reso- nant frequencies are even higher and the absorber influence is stronger above 450 MHz. 11.5.4 Current Distribution at Septum The resistance influence is more important at lower frequencies and can be neglected above 500 MHz. It is important to determine where to place the resistors; the VSWR must be kept in mind, and field must be uniform. This is why the resistors are placed closer to the septum edges, so that the currents in this area flow close to the edges and do not contribute to the nonuniformity of the field in the area desig- nated for testing. The testing area should be 25 to 45 cm away from the absorbers inside the cell. Placing the resistors close to the middle of the septum results in a worse VSWR at lower frequencies. The nonuniformity of the field increases as well. It is not important that the currents flow, but that they flow parallel with the septum edges (Figure 11.34). Figure 11.35 presents a photo of the GTEM cell built at FER. Testing area Good Not good Figure 11.34 Current flow in the septum.
  • 218. 212 Measurement Facilities Figure 11.35 GTEM cell built at FER.Selected Bibliography Crawford, M. L., “Generation of Standard EM Fields Using TEM Transmission Cells,” IEEE Transactions on Electromagnetic Compatibility, Vol. EMC-16, No. 4, November 1974. Crawford, M. L., J. L. Workman, and C. L. Thomas, “Expanding the Bandwidth of TEM Cells for EMC,” IEEE Transactions on Electromagnetic Compatibility, Vol. EMC-206, No. 3, August 1978. Hill, D. A., “Bandwidth Limitations of TEM Cells Due to Resonances,” Journal of Microwave Power, Vol. 18, June 1983, pp 181–195. Koch, M, and H. Garbe, “Analytische und Numerische Parameterstudien in TEM/Zellen rechteckformigen Querschnitts,” Proc. Elektromagnetische Vertraglichkeit, Deutschland: VDE- Verlag Gmbh, 1998, pp. 255–262. Koenigstein, D., Hansen, D., “A New Family of TEM-Cells with Enlarged Bandwidth and Opti- mized Working Volume,” Proc. 7th Zurich Symp. and Techn. Exh. on EMC, March 1987, pp. 172–132. Malaric, K., and J. Bartolic, “TEM Cell with 75Ω Impedance for EMC Measurements,” Proc. IEEE 1999 Int. Symposium on Electromagnetic Compatibility, Volume 1, August 2–6, 1999, Seattle, WA, pp. 234–238. Malaric, K., J. Bartolic, and B. Modlic, “Absorber and Resistor Contribution in the GTEM-Cell,” Proc. IEEE 2000 International Symposium on Electromagnetic Compatibility, August 21–25, 2000, Washington, D.C., pp. 891–896.
  • 219. 11.5 GTEM Cell 213 Malaric, K., J. Bartolic, and B. Modlic, “TEM-Cell with Increased Usable Test Area,” Proc. Inter- national Conference on Telecommunications—ICT ‘99, Vol. 2, June 15–18, 1999, Cheju, Korea, pp. 370–374. Morgan, D., A Handbook for EMC Testing and Measurement, London, U.K.: Peter Peregrinus Ltd., 1994. Nahman, N. S., et al., “Methodology for Standard Electromagnetic Field Measurements,” IEEE Transactions on Instrumentation and Measurement, Vol. IM-34, No. 4, December 1986. Weil, C. M., “The Characteristic Impedance of Rectangular Transmission Lines with Thin Center Conductor and Air Dielectric,” IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-26, No. 4, April 1978. Weil, C. M., and L. Gruner, “High-Order Mode Cutoff in Rectangular Striplines,” IEEE Trans- actions on Microwave Theory and Techniques, Vol. MTT-32, No. 6, June 1984. Wilson, P. F., et al., “Simple Approximate Expressions for Higher Order Mode Cutoff and Reso- nant Frequencies in TEM Cells,” IEEE Transactions on Electromagnetic Compatibility, Vol. EMC-28, No. 3, August 1986. Wilson, P., F. Gassmann, and H. Garbe, “Theoretical and Practical Investigation of the Field Dis- tribution Inside a Loaded/Unloaded GTEM Cell,” Proc. 10th International Zurich Symposium on EMC 1993, Zurich, Switzerland, 1993, pp. 595–598.
  • 220. CHAPTER 12Typical Test Equipment Interference into a piece of equipment or a system can be radiated or guided. While test facilities for radiated emission were covered in Chapter 11, in this chapter more attention will be paid to guided coupling. Coupling is performed either through capacitance or inductance. Typical test equipment includes: a line impedance stabi- lization network (LISN), coupling capacitors, coupling transformers, a parallel plate for a susceptibility test, coupling clamps and probes, injection clamps and probes, an EMI receiver, spectrum analyzers, and oscilloscopes.12.1 LISN—Line Impedance Stabilization Network A line impedance stabilization network (LISN) is a device for direct coupling with equipment under test (EUT). It is also called an artificial mains network (AMN) and is used to measure distortion signals on the mains cord of electrical EUTs and come in many types. Distortion signals are usually generated or picked up inside the EUT and the mains cord acts as an antenna. European and international EMC regula- tions define maximum permissible signal levels and frequency bands for such distor- tion signals (CISPR 16-1-2, MIL-STD-461D and E). For testing, LISN is placed between the power source and the equipment (device) to be tested during electromagnetic interference testing on power lines. Since the input impedance depends on frequency, LISN stabilizes the impedance at 50Ω. Furthermore, LISN filters the radio frequency noise from the mains supply. Finally, LISN transfers the conducted interference voltage produced by EUT to a spectrum analyzer or EMI receiver. LISN (Figure 12.1) is used for measuring the guided RF signal from the mains to the EUT. Every LISN has a low pass filter for rejecting noise on cables, as well as an effec- tive voltage transient limiter to protect sensitive analyzers or receivers from a strong energy shock. Large capacitors and inductors absorb unwanted noise energy and practically electrically isolate the EUT from the power mains. In this way, no unwanted guided noise from the power mains can enter the EUT, and vice versa. Thus, only the noise generated by the EUT will be measured by the spectrum ana- lyzer or EMI receiver. LISN is usually placed on a metallic board near the EUT (Figure 12.2). The EMI receiver is connected to the measuring port via a quality coaxial cable. Measuring unused LISN outputs are matched with 50Ω. LISN is widely used in commercial and military applications for electromagnetic compatibility measurements. How- ever, it is frequency limited. It can be used in the frequency range of 150 kHz to 30 MHz, although some models can be used at frequencies as high as 400 MHz. 215
  • 221. 216 Typical Test Equipment 50 µH To main s source To EUT 8 µF 0,25 µF To 50Ω load or to 50Ω input of meter 5Ω 1 kΩ Figure 12.1 LISN schematics. LISN for every line EUT Grounded board AC or DC port Coaxial Matching the unused cable 50Ω Input line with 50Ω impedance 50Ω Figure 12.2 Setup for measuring guided emission using LISN.12.2 Coupling Capacitor The coupling capacitor is used for testing spike voltages. Their typical value is 10 µF, which represents the known RF impedance on mains. Furthermore, it prevents undesirable frequencies from contaminating the power source (Figure 12.3). 10 µF capacitors should sustain voltages of up to 600-V DC and currents of up to 100A without significant losses. Transient generator Osciloscope EUT 10µF 10µF Figure 12.3 Testing spike voltage.
  • 222. 12.3 Coupling Transformer 217 Coupling capacitor dimensions are usually 0.85 × 0.85 × 0.7 cm. The attenua- tion of the RF signal on frequencies from 100 kHz to 1 GHz is approximately 60 dB.12.3 Coupling Transformer The coupling transformer is usually used to provide isolation of the power line from the main voltage. It should be connected parallel instead of with a serial connection, because the high current through the transformer can cause magnetic saturation of the transformer core. The values of R and C and the ratio of the transformer (Figure 12.4) depend on the data rate, power, and signal frequency. In industry, coupling transformers are used for ADSL, HDSL, VDSL, cables, and modems. Coupling transformers can prevent RF energy from entering the acoustic cables and are used in the frequency range from 30 Hz to 250 kHz.12.4 Parallel Plate for Susceptibility Test The parallel plate is used for radiated susceptibility tests from transient electromag- netic fields. The susceptibility of an EUT is the ability to withstand transient electro- magnetic fields. The parallel plate is shown in Figure 12.5. Its physical characteristics are width, w, and height, d. Characteristic impedance Z0 depends on the frequency, f, resistance, R, induc- tance, L, conductance, G, and capacitance, C, per unit length as shown: R + j2 πfL Z0 = (12.1) G + j2 πfC where C R N1 :N2 Input L1 L2 Output Figure 12.4 Coupling transformer. Conductor d Dielectric Conductor w Figure 12.5 Parallel plate.
  • 223. 218 Typical Test Equipment 2 R= (12.2) ωσ cond δ d L= µ (12.3) ω ω G = σ diel (12.4) d ω C=ε (12.5) d σcond and diel are the conductivities of the conductor and dielectric; µ and ε are permeability and permittivity of the dielectric. Skin depth, δ, is defined as: 1 δ= (12.6) πfµσ cond The parallel plate should be designed to have an impedance of 50Ω. The setup for testing susceptibility using the parallel plate is shown in Figure 12.6. Beside the plate, other necessary instruments include: a high voltage probe, a transient generator (ordinary monopulse), and a storage oscilloscope with a 200-MHz minimum single shot bandwidth and variable sampling rate up to 1 Gsa/s.12.5 Coupling Clamps and Probes Coupling clamps and probes are used in EMC testing, especially in emission testing. Emission testing is used to establish how much a certain device is emitting (or radi- ating). Immunity test are used to establish how immune the device is to interference Load High voltage probe Parallel plate line Transient generator Sensor Shielded enclosure Oscilloscope Figure 12.6 Susceptibility setup with parallel plate line.
  • 224. 12.5 Coupling Clamps and Probes 219 or outside radiation. Therefore there are instruments which measure the emission through capacitive or inductive coupling. They are usually not used for injecting the interference. The injection clamps or probes will be dealt with in the following section. 12.5.1 Capacitive Coupling Clamp The capacitive coupling clamp is an instrument used for measuring conducted elec- tromagnetic interference where there is no galvanic connection between the cou- pling clamp and the measured cables. The work principle of a coupling clamp is based on the capacity coupling that can exist between the measured cable and the clamp; it is used in testing cable resistance to fast transient sources like arcing on cable connectors when connecting the voltage network or ESD. If there is a source of continuous electromagnetic waves, the coupling clamp may be used for measur- ing the cables resistance. The capacitive coupling clamp conforms to the require- ments of ISO 61000-4-4. It guarantees that tests are carried out in strict compliance with the standard. A coupling clamp with its physical characteristics is shown in Figure 12.7. The clamp allows fast nanosecond pulse bursts (ISO 3a and 3b) to be injected in cable runs. The characteristic impedance of the unit is 50Ω. The coupling clamp is fitted with appropriate BNC connectors at both sides and is connected to the generator via a coaxial cable. The far side of the clamp has to be terminated with a 50Ω load resistor. It also provides a measurement output via a 40-dB attenuator. The coupling clamp can test cables of up to a 40-mm diameter and is actually a distributed capacitance. The effective coupling capacitance depends on the cross section and the material of the cable used, a typical value being around 100 pF. Inside the capacitive coupling clamp, where the cable is placed, an electric field exists; this results in capacity coupling. The time varying electrical field of an exter- nal system produces time varying charges in the disturbed system. For the capacitive coupling probe built at the Faculty of Electrical Engineering and Computing, Zagreb (Figure 12.8), the maximum coupling frequency is around 100 MHz, as it is with most models. 1000 mm High voltage mm 70 coaxial connector mm 70 Coupling plates 14 0m m 100 mm High voltage coaxial connector Insulating supports 1050 mm Figure 12.7 Capacitive coupling clamp dimensions.
  • 225. 220 Typical Test Equipment Figure 12.8 Capacitive coupling clamp built at FER, Zagreb. 12.5.2 Current Probe The current probe is a precision EMI measuring sensor, which clamps onto a wire, coaxial line, or cable carrying intentional or interference current. The diameter dimension is 5–10 cm. It is used to measure the current in a single wire, wire pair, coax, or bundle. Current probes are used on frequencies from 5 Hz to 1.2 GHz. Current probes are an excellent diagnostic tool, especially for locating and quantifying ground loops. When measuring the currents in cables or wires, the wires having the highest current often point to a solution, which can be: breaking ground loops or increasing their impedance by isolating or floating, applying ferrites, using bypass capacitors, applying a single-end grounded shield, or using filtered connectors. An important characteristic of current probes is transfer impedance. It is used for calibration. Figure 12.9 shows indirect measuring of the unknown current by measuring the voltage developed across a 50Ω load placed on the current probe’s coaxial cable (or input impedance of a spectrum analyzer). The computed unknown current, IdB µA, in units of dB µA equals the measured voltage, VdB µV, in units of dB µV minus the probe’s transfer impedance in units of dB (dB above an Ω per meter length): I dBµA = VdBµV − Z TdBΩ (12.7) The spectrum analyzer/EMI receiver can measure up to several mA of current. The principle is shown in Figure 12.10. There is a magnetic field around the wire through which the current flows. The ferrite core of the current probe concentrates this flow. At the current probe output, a voltage depending on permeability, ferrite core cross section, and the number of coils will appear: Vout = kµANfI in (12.8) Figure 12.9 Current probe.
  • 226. 12.6 Injection Clamps and Probes 221 Output to EMI Magnetic field receiver Vout Test coil around the wire H = 1/2πr I in I in Current through wire Figure 12.10 Cable current principle of work. where Vout is the output voltage, k is the constant, µ is the core permeability, A is the core cross section, N is the coil number, f is the frequency, and Iin is the wire current. Toroidal ferrite concentrates the field around the wire of EUT using the test coil. When constructing a current probe, it is necessary to install electrostatic shielding to prevent capacitive coupling between the coil and the EUT wire.12.6 Injection Clamps and Probes Beside emission tests, there are also immunity tests in which EUT is tested with out- side interference. The instruments for introducing interference into the EUT are called injection clamps or probes. They use capacitive or inductive coupling. The method is usually quite simple but includes large losses. It is used only when LISNs are not available. 12.6.1 Current Injection Probe The current injection probe is a method for injecting interference in immunity test- ing. It is simple, but relatively ineffective. It has high coupling losses (i.e., large power is necessary, and the results are hardly repeatable). It is recommended only if no other method is available or practical. As a transformer, the probe introduces only inductive coupling with no capaci- tive coupling (Figure 12.11). There is no isolation from other equipment, which is a serious drawback. The voltage on the EUT will depend on cable resonances at higher frequencies. Further- more, the parasitic capacitance between the probe and the cable will influence the local cable impedance. Thus, even though it is not necessary to ground the probe for the coupling, it is useful to ground its casing to the ground reference plane to lower this effect. The probe has losses of about 5 dB. It is used in the frequency range of up to 400 MHz. Usually a high power amplifier (200W) is necessary to perform the tests. 12.6.2 EM Clamp The EM clamp (Figure 12.12) is a tube made of slotted ferrite rings, which can be connected to a cable under test. It is not invasive and can be used on any cable type.
  • 227. 222 Typical Test Equipment Current injection probe Inductive coupling EUT Auxiliary equipment Ground reference plane Generator Figure 12.11 Injecting interference using a current injection probe. Figure 12.12 EM clamp. There are types with both inductive and capacitive coupling, which are used in the frequency range of 150 kHz to 1 GHz. However, this method is not as good as LISN. The losses are not high and it is not necessary to use a high power amplifier as is the case with the current injection probe. It is desirable to ground the clamp for better repeatability of test results. The clamp (ferrites) can also be used as a coupling clamp for emission tests, in which the clamp absorbs interfering signals from the cable. The setback is that for every testing frequency it is necessary to move the clamp along the cable, which can take a lot of time. Tests are almost impossible to perform automatically and have to be done manually. Table 12.1 shows the necessary power (in watts) for achieving 3 V/m and 10 V/m for different methods of injecting interference into the EUT. The best are the LISN and EM clamps; the current injection probe is recommended only if there are no other methods available.
  • 228. 12.6 Injection Clamps and Probes 223 Table 12.1 Necessary Power (W) Current LISN EM Clamp Injection Probe 3V 10V 3V 10V 3V 10V 10 kHz — — — — 585 6,500 150 kHz 0.59 6.5 1.46 16.25 29.32 325.78 27 MHz 0.59 6.5 0.94 10.4 4.21 46.8 80 MHz 0.59 6.5 0.59 6.5 4.62 51.35 230 MHz 0.59 6.5 0.59 6.5 5.85 65 12.6.3 Electrostatic Discharge (ESD) Generator Electrostatics discharge from the human body to a device or discharge between two devices can lead to interference or destruction of sensible electronic devices. The generated voltages can be up to several kV. Electrostatic discharge is characterized by a fast rise time (1 nanosecond), and intense discharge from humans, clothing, furniture, and other charged dielectric sources. The discharge resistance of humans may vary from several hundred ohms to 10 kohms. There are portable generators for testing ESD (IEC 1000-4-2, EN 61000-4-2), which can generate 50,000 pulses of 16.5 kV in the air and 10 kV in contact. They simulate ESD from a human or furniture. An ESD gun or generator is a hand-gun-shaped instrument containing a capacitor (typically about 150 pF—simu- lating a human), which can be charged up from 1 kV to 15 kV (sometimes more). One or more discharge resistors (approximately 1,500 ohms) and pulse-shaping networks acheive the undesired output waveform. The waveform (Figure 12.13) is characterized by a subnanosecond rise time and a current value of a few amperes. Figure 12.14 shows an equivalent circuit of the Imax 90% I at 30 ns I at 60 ns 10% 0.7–1 ns 30 ns 60 ns Figure 12.13 Waveform of an ESD generator. R C Figure 12.14 ESD generator circuit (R = 330Ω, C = 150 pF).
  • 229. 224 Typical Test Equipment generator. The resistance-capacitance network would produce a zero rise time. For generating the waveform, the actual ESD gun has a series inductance (due to the return current ground cable), which is responsible for the finite rise time in the cur- rent waveform. For the ESD experiment (Figure 12.15), the discharge could not be performed directly on the PCB trace because the gun-radiated field would have coupled to the traces, which could not be modeled. The model considers an impulsive transverse electromagnetic mode (TEM) wave reaching the victim trace where the suppressor is mounted. Although this con- dition can be realized by connecting a coaxial cable to the PCB and performing the discharge at the beginning of the cable, it must be done inside a shielded room to avoid propagation of the gun-radiated field. The PCB traces are then connected to a 1-GHz oscilloscope to capture the structure’s response to the ESD. The correspond- ing response time of the oscilloscope should be 0.35 nanosecond—sufficiently lower than the rise times.12.7 EMI Receiver The EMI receiver measures conducted emissions using an LISN or current probe, and radiated emissions using antennas according to international standards (CISPR 16-1). They measure RF signals with high accuracy. At the input, they have tunable preselectors for overload protection, bandwidth selection, tuning capability, and several detector functions. Tunable preselectors are passive filters that split pulsed signals into different frequency bands. At low frequencies these filters increase the size and weight of the EMI receiver in comparison to the spectrum analyzer. Mea- surements are performed on one frequency at a certain time, but some models can be programmed to sweep the frequency spectrum and record the results automatically. For covering the frequency range of 20 Hz to 40 GHz, at least three receivers are required. The most common digital interfaces are: RS232, Centronics, Ethernet, IEC-Bus (IEC625-2/IEEE 488-2), PS2-Keyboard, PS2-Mouse, USB, Userport, and VGA connectors. Impulse generators are used for broadband calibration of EMI receivers. However, most EMI receivers have internal built-in calibrating impulse generators. The main specifications of the EMI receiver include: interference, selectivity and shape factor, adjacent-channel interference, spurious response, intermodulation, cross-modulation, quasi-peak, peak and average value, frequency range at −6 dB, PC Oscilloscope Shielded room PCB ESD generator Figure 12.15 ESD measurement setup.
  • 230. 12.8 Spectrum Analyzer 225 time constants of charging and decharging, selectivity, intermodulation product limitation, noise limitation, and accuracy. The main advantages of an EMI receiver versus a spectrum analyzer are better sensitivity, durable circuitry, higher accuracy at measuring frequency, and ampli- tude, as well as higher dynamics.12.8 Spectrum Analyzer Spectrum analyzers are less expensive than EMI receivers and usually do not have an RF preselector. They have a higher noise figure (less sensitivity), less dynamic range for broadband emissions (such as an impulse generator), fewer detector func- tions, and less shielding in the case housing. Low-noise RF preamplifiers, preselec- tors, and CISPR quasi-peak adapters are optional. The spectrum analyzer usually has a tracking generator, several half-decade bandwidth selections, scan rate/band- width interlocks, and data storage for statistical manipulation. Similar to EMI receivers, spectrum analyzers measure the frequency and ampli- tude of electromagnetic signals in the frequency domain, but are used for different purposes. EMI receivers are used for distortion-free measurement of noncontinuous RF signals, while spectrum analyzers are used for high frequency, fast sweep, and continuous signals. EMI receivers respond properly to pulsed signals, which would overload a spectrum analyzer’s input circuitry (especially those with a low pulse repetition rate). The advantage of spectrum analyzers is that they can be very small, which is useful for design and diagnostic purposes. Cheap models have fewer functions and are used only for precompliance test- ing, while more sophisticated models can be used for the final compliance testing. With a current probe, the spectrum analyzer can be used to measure conducted emissions.12.9 Oscilloscopes An oscilloscope is a test instrument that displays waveforms of electronic and elec- trical circuits. In the past, oscilloscopes consisted of a cathode-ray tube and compo- nents that directed the electron beam based on the voltage of the input signal (vertical) and the scan produced by a time base (horizontal). Today most oscillo- scopes are digital. A digital oscilloscope converts the analog input signal into digital form (a series of binary numbers), which is then displayed or stored in memory. The ability to store waveforms is especially important for viewing one-time events. The oscilloscope is usually used with its probe. Probes usually have built-in 10:1 attenuators. More sophisticated oscilloscopes have broadband amplifiers for mea- suring voltage in submillivolt levels.Selected Bibliography C.I.S.P.R., “Specification for Radio Disturbance and Immunity Measuring Apparatus and Meth- ods,” International Electrotechnical Commission, Geneva, Switzerland, 1999.
  • 231. 226 Typical Test Equipment Cormier, B., and W. Boxleitner, “Electrical Fast Transient (EFT) Testing—An Overview,” Proc. IEEE International Symposium on Electromagnetic Compatibility, August 12–16, 1991, pp. 291–296. De Leo, R., F. Moglie, and V. M. Primiani, “Analyzing ESD Transient Suppressors in Printed Cir- cuits,” Compliance Engineering, 2001. Dipak, L., D. L. Sengupta, and V. V. Liepa, Applied Electromagnetics and Electromagnetic Com- patibility, New York: Wiley-Interscience, 2006. Morgan, D., A Handbook for EMC Testing and Measurement, London, U.K.: Peter Peregrinus Ltd., 1994. Kaires, R.G., “Stopping Electromagnetic Interference at the Printed Circuit Board,” Conformity, November 2003, pp. 12–21. Radman, S., I. Bacic, and K. Malaric, “Capacitive Coupling Clamp,” International Conference on Software, Telecommunications and Computer Networks, SOFTCOM, Split, CD-ROM, 2008. Reinhold, L., and P. Bretchko, RF Circuit Design: Theory and Applications, Upper Saddle River, NJ: Prentice-Hall, 2000. Sakulhirirak, D., V. Tarateeraseth, and W. Khanngern, “The Analysis and Design of Line Imped- ance Stabilization Network for an In-House Laboratory,” Proc. 2006 4th Asia-Pacific Confer- ence on Environmental Electromagnetics, August 2006, pp. 232–234. Sklar, B., Digital Communications: Fundamentals and Applications, Upper Saddle River, NJ: Prentice-Hall, 2001. Smith, D., “Current Probes, More Useful Than You Think,” Proc. 1998 IEEE International Sym- posium on EMC, 1998, pp. 284–289.
  • 232. CHAPTER 13Control of Measurement Uncertainty13.1 Evaluation of Standard Uncertainty Measurement uncertainty is a parameter associated with the result of a measure- ment that characterizes the dispersion of the values that could reasonably be attrib- uted to the measurand. Uncertainty expresses doubt about the result of a measurement. The real value of a measured quantity can never be known exactly—it can only be estimated. The true value lies inside the uncertainty interval with a certain degree of probability (level of confidence). For most cases the suffi- cient level of confidence is 95%, obtained with coverage factor k = 2 (for k = 1 the level of confidence is 68%—one standard deviation). The evaluation of standard uncertainty is defined either by statistical analysis of a series of observations (i.e., repeatable measurements) (type A), or by systematic components of uncertainty (type B). For combining uncertainty components of the measurement, a probability density function (pdf) must be chosen for each uncertainty component. If the uncer- tainty component has random errors (electrical noise, connector repeatability), then the pdf usually has a normal (Gaussian) distribution. For systematic errors (from the manufacturer’s data sheet) rectangular (uniform) distribution is used, while uncertainties in measurements at microwave frequencies (phase influence) are best described with a U- shaped distribution. 13.1.1 Type A Evaluation of Standard Uncertainty When evaluation of uncertainty is done by statistical analysis of a series of observa- tions, then it is called a type A evaluation. In this type of evaluation, standard uncer- tainty of a measurand is calculated from a series of repeated observations. Even if for some measurements the random component of uncertainty is not relevant to other contributions of uncertainty, it is desirable to establish the scale of random effects on the measurement process. The average or mean value of the measure- ments should be calculated. If there are n independent repeated values for a quantity Q, then the mean value q is obtained by 1 n q + q + q Kq q= ∑qj = 1 2 n 3 n n j =1 (13.1) 227
  • 233. 228 Control of Measurement Uncertainty The measured results will be spread over a certain range, which depends on var- ious factors, such as: measurement method, measuring device used, and even the person making measurements. The resulting dissipation is defined as standard devi- ation σ 1 n ( ) 2 σ= ∑ qj − q n j =1 (13.2) The above expression gives the standard deviation σ for the particular set of n measurements. If the process is repeated at some later time, with a different number n of measurements, different values of q and σ will be obtained. For a very large n, the mean value will approach the central limit of the distribution of all possible val- ues. The probability density function will have a normal distribution. From the results of a single set of measurements and their standard deviation, σ, an estimate for all possible values of the measurand s(qj) can be made with 1 n ( ) ( ) 2 s qj = ∑ qj − q n − 1 j =1 (13.3) The mean value q is obtained from a finite number n of measurement results; the mean value is not the exact mean that would have been obtained if an infinite number of measurements could have been taken. Therefore, even the mean value has its uncertainty, which is called the standard deviation of the mean. Its value can be obtained from the estimated standard deviation using s( q ) = s qj( ) (13.4) n 13.1.2 Type B Evaluation of Standard Uncertainty The type B evaluation is defined as all uncertainty other than repeatable measure- ment uncertainty. It is associated with systematic errors. The evaluation of these contributions depends on previous experience, the measurement process, manufac- turer specifications, calibration data, and the environment. When all possible systematic components of uncertainty are identified, proba- bility distributions should be assigned to them. Although probability distributions can be of any form, the most common ones for the type B evaluation of standard uncertainty are the rectangular (uniform) and U-shaped distributions.13.2 Distributions The three most common distributions for the evaluation of uncertainties are normal (Gaussian), rectangular, and U-shaped. While normal distribution is used for the type A evaluation, rectangular and U-shaped distributions are used for the type B evaluation.
  • 234. 13.2 Distributions 229 13.2.1 Normal (Gaussian) Distribution Normal distribution is shown in Figure 13.1. The distribution size is described with the standard deviation. The shaded area represents 1 standard deviations from the center of distribution. This is approximately 68% of the area under the curve. This means that for coverage factor k = 1 there is a 68% probability that the measure- ment value is in this range. Table 13.1 shows the coverage probability versus cover- age factor. In some situations it is necessary to use a higher coverage factor for a higher probability. Usually p = 95% (k = 1.96 or 2.00) is enough. The values xi of the input quantities Xi all have their uncertainties, u(xi), which are called standard uncertainties. Standard uncertainty for a normal distribution is equal to the standard deviation of the mean [i.e., s(q )]: u( x i ) = s( q ) (13.5) 13.2.2 Rectangular Distribution A rectangular distribution (Figure 13.2) is used for a measuring instrument with an accuracy of ±x or ±dB without any statistical information. The result may lie any- where between −x to +x with equal probability. Outside of this range the probabil- ity of xi is zero. This distribution is used for type B evaluations. For some instruments the resolution will be a = 0.5 of the least significant digit. When there is no previous knowledge about the measurement quantity, rectangular distribution must be used. Standard uncertainty for a rectangular distribution is calculated from Table 13.1 Coverage Probability Depending on the Coverage Factor Coverage Coverage Probability p (%) Factor k 68% 1.00 90% 1.64 95% 1.96 95.45% 2.00 99% 2.58 99.73% 3.00 68% Figure 13.1 Normal (Gaussian) probability distribution.
  • 235. 230 Control of Measurement Uncertainty a a Probability p xi − a xi xi + a Figure 13.2 Rectangular distribution. ai u( x i ) = (13.6) 3 13.2.3 U-Shaped Distribution Since measurements at microwave frequencies often involve vector quantities (both magnitude and phase), it is sometimes necessary to use a U-shaped distribution (Fig- ure 13.3). This distribution is most commonly used for the RF mismatch uncer- tainty where the phase information for a given vector is unknown. Mismatch uncertainty occurs when there is no perfect matching of impedance between the source and load (termination). If the phase is unknown, the cosine function will determine the probability distribution. With this function, xi will probably be closer to one of the edges of the distribution rather than in the center. Standard uncertainty for a rectangular distribution is calculated from ai u( x i ) = (13.7) 2 13.2.4 Combined Standard Uncertainty When different input uncertainties are combined, the normal distribution will be used. Since normal distribution is described with standard deviation, all input uncertainties have to be evaluated and combined with their sensitivity coefficients xi − a xi xi + a Figure 13.3 U-shaped distribution.
  • 236. 13.3 Sources of Error 231 to form the normal distribution. The standard uncertainties, xi, and their sensitivity coefficients, ci, will give a single value to be associated with y of the measurand Y. The combined standard uncertainty will then be: m m u c ( y) = ∑c 2 i u2 ( x i ) ≡ ∑ u ( y) 2 i (13.8) i =1 i =1 When one contribution dominates, the resulting contribution will be very simi- lar to the dominating one. With the excepting of a few cases, the resulting contribu- tion will usually be normal, no matter what the contribution distribution is. 13.2.5 Expanded Uncertainty When purchasing measurement equipment, a calibration certificate usually quotes an expanded uncertainty, U, with a high coverage probability. Using coverage fac- tor k, the standard uncertainty can be calculated as follows: U u( x i ) = (13.9) k13.3 Sources of Error There are two types of measurement errors: random and systematic. Random errors are evaluated in type A evaluations and are normally distributed as shown above. Systematic errors are evaluated in type B evaluations and can shift the mean value (probable value) and add an uncertainty. 13.3.1 Stability All instrument performance changes with time; the value of resistors changes and microwave attenuators drift, which is evaluated by calibration. The drift is most likely not linear. The data over time should be displayed graphically and the most probable value selected. 13.3.2 Environment Temperature and humidity can affect the performance of attenuators, power sen- sors, and other equipment. Therefore, measurements should be performed in labo- ratory conditions with defined temperature and humidity. 13.3.3 Calibration Data Usually, calibration points are limited. Sometimes quantity values other than the calibration point have to be measured, at which systematic errors can occur. If pos- sible, calibration should be performed with some other calibration instrument, or the value prediction can be made from other similar models’ data.
  • 237. 232 Control of Measurement Uncertainty 13.3.4 Resolution Another systematic error of the measuring device is the digital rounding error. A quantization error of ±0.5 is present, since the measured value is converted from analog to digital. Noise in the system can cause fluctuations of the last digit as well. 13.3.5 Device Positioning The position between the measuring instrument and the device under test can also lead to systematic error. There could be leakage currents to Earth as well as electro- magnetic leakage fields. Mutual heating can be avoided by placing the instruments farther apart. 13.3.6 RF Mismatch Error Characteristic impedance mismatch of the measurement transmission line is one of the most common systematic errors in power and attenuation measurements because the phases of voltage reflection are usually unknown, making it hard to make the corrections.13.4 Definitions The terms described in this chapter are given in the “ISO Guide to the Expression System of Uncertainty Measurement,” “IEEE Std 100-1988,” and in “International Vocabulary of Basic and General Terms in Metrology.” For definitions, please see the Glossary.Selected Bibliography Agilent Technologies, “Application Note 64-1B, Fundamentals of RF and Microwave Power Measurements Classic Application Note on Power Measurements,” 2000. Bronaugh, E., and J. Osburn, “A Process for the Analysis of the Physics of Measurement and Determination of Measurement Uncertainty in EMC Test Procedures,” Proc. IEEE 1996 Interna- tional Symposium on Electromagnetic Compatibility, August 19–23, 1996, p. 245. “CISPR 16-4-2 (Ed.1.0). Specification for Radio Disturbance and Immunity Measuring Appara- tus and Methods—Part 4-2: Uncertainties, Statistics and Limit Modeling—Uncertainty in EMC Measurements,” IEC Standard, November 2003, p. 43. Heise, E. R., and R. E. W. Heise, “A Method to Calculate Uncertainty of Radiated Measure- ments,” Proc. IEEE 1997 Symposium on Electromagnetic Compatibility, August 18–22, 1997, p. 359. “International Vocabulary of Metrology—Basic and General Concepts and Associated Terms (VIM),” ISO/IEC Guide 99, 2007. Kurosawa, T., et al., “Study on Measurement Uncertainty in Immunity Testing: IEC61000-4-6,” Proc. Electromagnetic Compatibility and 19th International Zurich Symposium on Electromag- netic Compatibility, May 19–23, 2008, pp. 598–601. Ridler, N., et al., “Measurement Uncertainty, Traceability, and the GUM,” IEEE Microwave Magazine, Vol. 8, No. 4, August 2007, pp. 44–53.
  • 238. 13.4 Definitions 233 Taylor, B. N., and C. E. Kuyatt, Guidelines for Evaluating and Expressing the Uncertainty of NIST Measurement Results, NIST Tech. Rep. TN1297, Gaithersburg, MD, 1994.
  • 239. Appendix ACommunication Frequency AllocationsA.1 Frequency Allocation in the United States 218–219-MHz Radio Service 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz 218–2,190 — — — 700-MHz Guard Service (Digital TV) 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz — 746–747 — — 762–764 776–777 792–794 3650–3700-MHz Radio Service 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz — — 3.650–3.700 — Access Broadband over Power Line (BPL) 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz 30–80 — — — Advanced Wireless Services Including 3G 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz — 1,710–1,755 — — 1,915–1,920 1,995–2,000 2,020–2,025 2,110–2,180 235
  • 240. 236 Appendix A Amateur Radio 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz 50–54 420–450 3.3–3.5 47.0–47.2 144–148 902–928 5.650–5.925 76–81 219–220 1,240–1,300 10.0–10.5 122.25–123.0 222–225 2,300–2,310 24.0–24.25 134–141 2,390–2,450 241–250 275–300 Auditory Assistance Devices 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz 72–73 — — — 74.6–74.8 75.2–76 216.75–217 Automatic Vehicle Identification Systems 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz — 2,900–3,000 3.0–3.26 — 3.267–3.332 3.339–3.3458 3.358–3.6 Auxiliary Broadcasting 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz 54–72 450–454 6.425–6.525 76–81 76–108 455–456 6.875–7.125 152.855–154 470–608 12.7–13.25 157.45–161.575 614–806 17.7–18.3 161.625–161.775 944–960 19.3–19.7 162.0125–173.2 2,025–2,110 174–216 2,450–2,483.5 Aviation/Aeronautical 30 MHz–300 MHz 300 MHz–3,000 MHz3 GHz–30 GHz 30 GHz–300 GHz 72–73 328.6–335.4 3.5–3.65 32.3–33.4 74.6–75.2 849–851 4.2–4.4 108–137 894–896 5.00–5.25 156.2475–157.0375 960–1,215 5.35–5.46 1,300–1,350 9.0–9.2 1,435–1,525 13.25–13.4 1,535–1,660.5 15.4–15.7 2,310–2,320 24.75–25.05 2,345–2,395 2,700–3,000
  • 241. Appendix A 237 Basic Exchange Telephone Radio Service 30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz 152–159 450–460 — —