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  • 1. Peter Devine Peter Devine Radar level measurement Radar level measurement Radar level measurement The user's guide The user's guide
  • 2. Radar level measurement - The users guide Peter Devine written by Peter Devine additional information Karl Grießbaum type setting and layout Liz Moakes final drawings and diagrams Evi Brucker © VEGA Controls / P Devine / 2000 All rights reseved. No part of this book may reproduced in any way, or by any means, without prior permissio in writing from the publisher: VEGA Controls Ltd, Kendal House, Victoria Way, Burgess Hill, West Sussex, RH 15 9NF England. British Library Cataloguing in Publication Data Devine, Peter Radar level measurement - The user´s guide 1. Radar 2. Title 621.3´848 ISBN 0-9538920-0-X Cover by LinkDesign, Schramberg. Printed in Great Britain at VIP print, Heathfield, Sussex.
  • 3. Contents Foreword Acknowledgement Introduction ix xi xiii Part I 1. History of radar 2. Physics of radar 3. Types of radar 1. CW-radar 2. FM - CW 3. Pulse radar 1 13 33 33 36 39 Part II 4. Radar level measurement 1. FM - CW 2. PULSE radar 3. Choice of frequency 4. Accuracy 5. Power 47 48 54 62 68 74 5. Radar antennas 1. Horn antennas 2. Dielectric rod antennas 3. Measuring tube antennas 4. Parabolic dish antennas 5. Planar array antennas Antenna energy patterns 77 81 92 101 106 108 110 6. Installation A. Mechanical installation 1. Horn antenna (liquids) 2. Rod antenna (liquids) 3. General consideration (liquids) 4. Stand pipes & measuring tubes 5. Platic tank tops and windows 6. Horn antenna (solids) B. Radar level installation cont. 1. safe area applications 2. Hazardous area applications 115 115 115 117 120 127 134 139 141 141 144
  • 4. Foreword To suggest that any one type of level transmitter technology could be regarded as 'universal' would be unrealistic and potentially irresponsible due to the variation and complexity of available applications when liquids, powders and solids are all considered. However, the rate at which radar based level transmitters have established themselves over the last couple of years would tend to suggest that this technology is closer to that definition that any principle has ever been. I have personally been involved in the development, applications, sales and marketing of level transmitters, controllers and indicators of most types over the last twenty years. In that time nothing has, in my opinion, come close to matching the significance of radar in terms of its overall suitability, for not just conventional but extreme process conditions applications for the vast majority of substances in vessels of virtually any size or complexity. This unique principle combined with current reflections processing software, materials of construction, simplicity of installation and transmitter digital communications allows this to be considered as a day to day 'first consideration' for level, whereas only a very short time ago it was regarded as expensive and specialised - this is no longer the case. The purpose of this publication is quite specific, and that is to explain some of the principles involved, and to show that by applying some simple guidelines, what is obviously a sophisti-cated technology can be simple and reliably used in an enormously wide range of industrial and process applications. We make no apology for including a chapter on Vega specific products, and hope this guide stimulates a radar user, or some greater depth of knowledge if you have some experience, we look forward to hearing from you. Mel Henry Managing Director Vega Controls Ltd. ix
  • 5. Acknowledgements In writing and compiling this book I had the invaluable assistance of several colleagues from VEGA in Schiltach both in the developing department and within the product management. Particular thanks must go to Karl Griessbaum for his lucid explanations of the 'secrets' of pulse radar; his insight into the workings of FM - CW radar and the drawings to accompany the explanations. Thanks also to Juergen Skowaisa and Juergen Motzer for their technical contributions to the book. The publication of 'radar level measurement - the user´s guide' is a reflection of the wealth of product knowledge of radar level application experience in the VEGA group of companies and our agents and distributors world wide. This experience has accelerated since the advent of the VEGAPULS 50 series two wire, loop powered radar. I would like to thank all those who contributed to the section on radar applications. This in-cludes Doug Anderson, Dave Blenkiron, Chris Brennan, Graeme Cross and John Hulme in the UK, Paal Kvam of Hyptech in Norway, Dough Groh and his colleagues at Ohmart VEGA in the USA, and Juergen Skowaisa and Roger Ramsden from VEGA Germany. Thank also to the VEGA marketing department in Germany and the UK for their assistance in producing and collating pictures and photographs. Thank to all the other unnamed contributors. Finally, the most important contributors to this book are all VEGA radar users world wide without whom our high level of expertise in process radar measurement applications would not be possible. Peter Devine Technical manager Vega Controls Ltd. xi
  • 6. Introduction The technical benefits of radar as a level measurement technique are clear. Radar provides a non-contact sensor that is virtually unaffected by changes in process temperature, pressure or the gas and vapour composition within a vessel. In addition, the measurement accuracy is unaffected by changes in density, conductivity and dielectric constant of the product being measured or by air movement above the product. These benefits have become more significant to the process industry since the advent of low costs, high performance, two wire loop powered radar level transmitters. This breakthrough, in the summer of 1997, produced an unprecedented boom in the use of non-contact microwave radar transmitters for liquid and solids process level application. 'Radar level measurement - the user´s guide' is offered as a reference book for all those interested in the technology, the application, and the practical installation of radar level sen-sors. We cover many practical process level applications rather than the closed niche market of custody transfer measurement. Radar history, physics and techniques are presented as well as descriptions of types of ra-dar antenna and mechanical and electrical installations. Now radar is an affordable option for process level measurement. We compare it closely with all of the other process level techniques and give many examples of the myriad applications of radar across all industries. Radar level measurement has come of age. We hope that this book will be invaluable in helping you to see the potential of this latest and almost universal level measurement technology. More than anything, we hope that you enjoy delving into the pages of this book. Peter Devine Technical manager Vega Controls Ltd xiii
  • 7. 1. History of radar James Clerk Maxwell predicted the existence of radio waves in his theory of electromagnetism as long ago as 1864. He showed mathematically that all electromagnetic waves travel at the same velocity in free space, independent of their wavelength. This velocity is of the order of 300,000 kilometres per second, the speed of light. Heinrich Rudolf Hertz, verified Maxwell’s theory by experiments carried out in 1886-87 at Karlsruhe Polytechnic. He used a spark gap transmitter producing bursts of high frequency electromagnetic waves at about 455 MHz, or a wavelength of 0.66 metres. Hertz confirmed that these electromagnetic radio waves had the same velocity as light and could be reflected by metallic and dielectric bodies. In addition to their reflective properties, Hertz demonstrated that radio waves exhibit refraction, diffraction, polarization and interference in the same way as light. These early experiments in reflecting radio waves off metal plates were the first manifestations of radar as we know it today. The first practical form of radar was produced by a German engineer, Christian Hülsmeyer. Patented in various countries in 1904 as the ‘Telemobiloscope’, Hülsmeyer’s apparatus was described as ‘A Hertzian wave projecting and receiving apparatus adapted to indicate or give warning of the presence of a metallic body, such as a ship or a train, in the line of projection of such waves’. An addition to the patent in the same year described ‘Improvements in Hertzian wave projecting and receiving James Clerk Maxwell predicted the existence of radio waves in his theory of electromagnetism (Pic. 1.1 - J.C.M.F) Heinrich Hertz Hertz confirmed by experiment that electromagnetic radio waves have the same velocity as light and can be reflected by metallic and dielectric bodies (Pic. 1.2 - I.N.T) 1
  • 8. Prior to World War II, radar was being developed independently in a number of different countries, including Britain, Germany, the United States, Italy, France and the Soviet Union. In 1934, following a series of experiments at the Naval Research Laboratory in the United States, a patent was granted to Taylor, Young and Hyland for a ‘System for detecting objects by radio’. In February 1935, British scientist, Robert Watson-Watt presented a paper on ‘The detection and location of aircraft by radio methods’ to the Tizard Committee for the Scientific Survey of Air Defence. Christian Hülsmeyer produced the first practical radar patented in 1904 (Pic. 1.3 - D.M.M) apparatus for locating the position of distant metal objects’. A successful demonstration of the telemobiloscope was made at the International Shipping Congress in Rotterdam in 1904, and also to the German navy. However, the telemobiloscope was considered to be limited and was not a commercial success. Guglielmo Marconi, is famous for pioneering trans-Atlantic radio communications. In 1922 Marconi had also recognised the potential of using short wave radio for the detection of metallic objects. Marconi envisaged the use of radio for ship to ship detection at night or in fog. However, he did not appear to receive the support or have the resources to carry these ideas further at the time. 2 Guglielmo Marconi recognised the potential of using short wave radio for the detection of metallic objects in 1922 (Pic. 1.4 - GEC Marconi)
  • 9. 1. History of radar Sir Robert Watson - Watt was a senior figure in the development of British radar in the 1930’s & 40’s (Pic. 1.5 - I.W.M) Subsequently, a practical demonstration was carried out using a BBC radio transmitter at Daventry. About five and a half miles (9 km) away, a separate radio receiver connected to an oscilloscope was used to detect the presence of a Handley Page Heyford aircraft as it flew between the transmitter and receiver. Both the American system and Watson-Watt’s Daventry experiment were types of continuous wave (CW) radar. Called CW wave-interference radar or bistatic CW radar, a continuous single frequency was transmitted from one point and detected by a receiver at a separate location. The receiver also detects doppler shifted echoes from the target object. The interference between the frequency of the direct signal and reflected signals at a slightly different frequency indicated the presence of the target object. If you are unfortunate enough to live on an airport flight path, you may have witnessed this effect on your television screen. As an aircraft approaches, the picture on the screen may flicker with regular horizontal bands scrolling vertically on the screen. These diminish when the aircraft is directly overhead and then continue as the aircraft moves away. Although it proved a point at Daventry, CW wave-interference radar was not a practical device. It could detect the presence but not the position of the target. After Daventry, the British effort continued at Orford Ness and then nearby Bawdsey Manor on the Suffolk coast. It was clear that pulse radar would be needed to provide the required distance and direction information essential for a defensive radio detection system. The British, under the direction of Watson-Watt developed a defensive system of CH (Chain Home) radar stations which eventually covered all of the coastal approaches to Britain. The standard chain home radars had a relatively low frequency of between 22 & 30 MHz (wavelength 10 to 13.5 metres). They had a power of 200 kilowatts and a range of up to 190 kilometres. However, the long range CH radar transmitters were blind to low flying aircraft and therefore they were supplemented by CHL (Chain Home Low) radar transmitters which had a shorter range and covered the lower altitudes that were overlooked by the main CH 3
  • 10. British Chain Home Radar aerials Radar was instrumental in the defence of Britain during the second world war (Pic. 1.6 - I.W.M) transmitters. They operated on a frequency of 200 MHz (wavelength 1.5 metres). It is well documented that the CH and CHL network of radar stations were a crucial factor during the Battle of Britain in the summer of 1940. It enabled the fighters of the Royal Air Force to be deployed when and where they were needed and rested when the threat receded. The limited resources in men and machines were not wasted in long standing patrols. German radar research was also conducted in secret in the late 1930’s. Whereas the development effort in Britain was focused on air defence, in Germany separate radar developments were carried out for the Navy, Army and Luftwaffe. Companies involved in German naval research produced a range of ship 4 mounted sea search radar transmitters called Seetakt. These were delivered as early as 1938 with a frequency of 366 MHz (wavelength 82 cm) and were installed on many vessels including the famous battleships, Bismarck and Graf Spee. German Naval developments also produced the Freya range of search radars operating on 125 MHz (wavelength 2.4 metres). These were found to be effective for tracking aircraft at long range, and were subsequently supplied to the Luftwaffe for early warning. However, they could not provide altitude information. Other German radars in wide use were the parabolic antenna Würzburg and Würzburg Riese (Giant Würzburg) transmitters. The standard Würzburgs were generally used for directing searchlights and flak batteries and the Würzburg Riese for tracking individual intruders and directing night fighters to intercept them. In a similar fashion to the British Chain Home system, the Germans built a defensive network of ‘Himmelbett’ radar stations. The literal translation of Himmelbett is four poster bed. The four ‘posts’ of the bed consisted of a Freya early warning radar, a Würzburg radar for tracking the intruding aircraft, a Würzburg radar to guide the night fighter to the intruder and a Seeburg plotting table (Seeburgtisch) to monitor the interception. This defensive radar system became known by the British as the ‘Kammhuber Line’ after the German general in charge of night fighters.
  • 11. 1. History of radar Above - The famous aerial reconnaissance photograph of a German Würzburg radar antenna at Bruneval in northern France. This image alerted the British to the presence and advanced state of German defensive radar which led to a commando action in which components from the radar were taken back to Britain for analysis (Pic. 1.7 - I.W.M) Right - The German Würzberg radar was used for directing searchlights and flak batteries and for tracking individual targets and directing interceptors to them (Pic. 1.8 - P.D) 5
  • 12. Both Britain and Germany developed airborne radar for fighter interception by night. British airborne radar trials started in 1937 with the production AI Mark 1 taking to the air in May 1939. The first practical British Airborne Interception radar was the AI Mark IV which was first tested in August 1940. In Germany the Lichtenstein airborne radar was available in mid 1941. The characteristic external radar aerial array of the Lichtenstein caused significant aerodynamic drag. This could reduce the aircraft speed by as much as 40 kilometres per hour. By 1943 the range had been extended to 6000 metres. It became clear to radar researchers that a shorter ‘centimetric’ wavelength would be more useful for a number of applications. This would enable a more focused airborne radar that would not suffer from the ground returns that restricted capabilities of the first airborne radars. The higher frequency could be used for a ground mapping radar unit to locate towns and other geographic features. The problem was how to find a method of generating sufficient power at the desired wavelength of 10 centimetres. British Airborne Radar - AI Mark IV developed for fighter interception by night in 1940 German Airborne Radar ‘Lichtenstein’ available in mid 1941 - the external aerial radar caused significant aerodynamic drag (Pic. 1.9 - I.W.M) (Pic. 1.10 - I.W.M) 6
  • 13. 1. History of radar In late February 1940, an historic breakthrough was made by John Randall and Harry Boot, researchers at the University of Birmingham, when they tested their world changing invention the Cavity Magnetron. The heart of this cavity magnetron was a simple solid copper block with six cavities machined into it. In the centre was the cathode. When a strong magnetic field and high voltage was applied between the copper block and the cathode, the stream of electrons resonated in unison within the cavities instead of passing directly to the copper block anode. The frequency of oscillation was calculated to be about 3 GHz (10 centimetre wavelength). The theoretical calculations of the prototype cavity magnetron were correct. The actual wavelength was found to be 9.87 centimetres and the all important power of the prototype was 400 Watts. Cavity Magnetron the world changing invention by John Randall and Harry Boot invented in 1940 (Pic. 1.11 - GEC) Production of cavity magnetrons followed very quickly and the power output was significantly increased. Britain developed microwave airborne interception AI radar sets for night fighters which had a vastly improved long and near range. The British microwave airborne interception radar was the AI Mark VII which was introduced in mid 1942. The improved AI Mark VIII was mass produced and in wide use by early 1943. The Cavity Magnetron was used in centrimetric ‘microwave’ airborne radar and duced a quantum leap in performance. The radar dish was protected inside a plastic nose assembly pro- (Pic. 1.12 & 1.13 - H.R.A) 7
  • 14. Britain also used the cavity magnetron in the development of a ground mapping radar called H2S. This device enabled aircraft to be accurately navigated to their destinations without the aid of ground based beacons or beams. Britain shared this secret microwave technology with the United States where additional development took place at the Radiation Laboratory within the Massachusetts Institute of Technology. From the work carried out at MIT, further airborne interception radars and gun laying radars were mass produced and delivered to the allied forces. The American SCR-720 (known as AI Mark X in Britain) was first delivered to the USAAF by late 1942. This radar unit became a standard device long after the war had finished. War time secrecy meant that radio detection devices were given coded names. In Britain, the early chain home radar was called RDF after the existing Radio Direction Finding systems in the hope that it would mislead their real function. In the same way in Germany, radar was disguised as ‘Dezimeter Telegraphie’ or ‘De-Te’, translated as decimetric telegraphy It was the Americans who introduced the now universally used palindrome, RADAR or RAdio Detection And Ranging. The history of the development of radar during the course of the Second World War is a huge subject in itself. Many devices were developed. Measures and counter measures were taken in the radar war. Since 1945, radar has been used for an increasing number of peaceful applications. The giant Würzburg parabolic radar transmitters of the Second World War became post war radio telescopes. The basic designs were developed and enlarged and can be seen at the well known Jodrell Bank Observatory near Manchester which has a dish diameter of 75 metres. Viewed from Earth, the planet Venus Modern radar systems are exemplified by this ‘AWAC’ airborne early warning aircraft. Multiple targets can be detected at extreme range (Pic. 1.14 - P.D) 8
  • 15. 1. History of radar is one of the brightest celestial bodies. However, the mysteries of our close neighbour in the Solar System were only uncovered with the assistance of radar. The surface of Venus is shrouded in dense clouds of vapour including carbon dioxide gas at pressures of 90 bar and an average temperature of 750 K. Earth bound pulse radar measurements over an extended period of time were used to calculate the radius of the orbit of Venus. Doppler shift measurements from the surface were used to calculate the rate of rotation of the shrouded planet. The Venus ‘day’ was found to be 243 Earth days. During the 1970’s, radar mapping of the planet’s surface by space probe uncovered surface features such as craters. Jodrell Bank - the observatory near Manchester which has a 75 metre dish diameter (Pic. 1.15 - P.D) Detection by radar is not always desirable. Huge sums of money have been spent reducing the radar signature of the F117 stealth fighter (Pic. 1.16 - P.D) 9
  • 16. Radar technology is part of our everyday lives. The cavity magnetron is used in microwave ovens. Continuous wave (CW) radars are used in automatic door detection and vehicle speed measurement. Other well known civilian radar applications include air traffic control, shipping and weather radar. Radar altimeters developed in the 1930’s use a form of radar called FM - CW or Frequency Modulated Continuous Wave radar. In the 1970’s, the same FM - CW measurement technique was used in the production of the first radar level tank gauge. Initially these radar level transmitters were used to measure petroleum products in supertankers. Further developments of FM - CW level transmitters led to their use on shore based storage tanks in the mid 1980’s. Originally these were expensive, high accuracy systems for fiscal measurement of petroleum products. Later, lower accuracy FM - CW radar transmitters became available for the process industry. In the late 1980’s, pulse radar level transmitters were developed for process measurement applications. The availability of suitable crystals and solid state components such as GaAs FET oscillators enabled cost effective radar level transmitters to enter the market. In 1997 a significant improvement in the specification of radar level transmitters was achieved. VEGA produced the world’s first two wire, loop powered, intrinsically safe radar level transmitter. For the first time low cost, high specification radar level transmitters became available. It is likely that these advances will continue into the new millennium and that radar level transmitters will become a commodity item in the same way as differential pressure transmitters. In the field of radar level measurement, technological advances have resulted in two wire, intrinsically safe transmitters (Pic. 1.17 - Vega) 10
  • 17. 1. History of radar Comparing the old with the new A raw oscilloscope echo trace had to be interpreted by skilled operators using the British war time Chain Home Low radar (Pic. 1.18 & 1.19 - I.W.M) Comprehensive information is available on the PC echo trace of the latest two wire loop powered radar level transmitters (Pic. 1.20 - Vega Pic. 1.21 - Vega) 11
  • 18. Inhalt Foreword Acknowledgement Introduction ix xi xiii Part I 1. History of radar 2. Physics of radar 3. Types of radar 1. CW-radar 2. FM - CW 3. Pulse radar 1 13 33 33 36 39 Part II 4. Radar level measurement 1. FM - CW 2. PULSE radar 3. Choice of frequency 4. Accuracy 5. Power 47 48 54 62 68 74 5. Radar antennas 1. Horn antennas 2. Dielectric rod antennas 3. Measuring tube antennas 4. Parabolic dish antennas 5. Planar array antennas Antenna energy patterns 77 81 92 101 106 108 110 6. Installation A. Mechanical installation 1. Horn antenna (liquids) 2. Rod antenna (liquids) 3. General consideration (liquids) 4. Stand pipes & measuring tubes 5. Platic tank tops and windows 6. Horn antenna (solids) B. Radar level installation cont. 1. safe area applications 2. Hazardous area applications 115 115 115 117 120 127 134 139 141 141 144
  • 19. 2. Physics of radar Electromagnetic waves Th e velocity of light in free space is 299,792,458 metres per second, but who is timing? For the purposes of the calculations in this book, we will call it 300,000 kilometres per second or 3 x 108 metres per second. Maxwell’s theories of electromagnetism were confirmed by the experiments of Heinrich Hertz. These show that all forms of electromagnetic radiation travel at the speed of light in free space. This applies equally to long wave radio transmissions, microwaves, infrared, visible and ultraviolet light plus X-rays and Gamma rays. Maxwell showed that the velocity of light in a vacuum in free space is given by the expression : Examples :- 1 co = µo εo o x εo) [Eq. 2.1] velocity of electromagntic wave in a vacuum in metres / second the permeability of free space (4 π x 10 -7 henry / metre) the permittivity of free space (8.854 x 10 -12 farad / metre) c = f xλ [Eq. 2.2] c velocity of electromagnetic waves in metres / second f λ frequency of wave in second -1 wavelength in metres The original cavity magnetron had a wavelength of 9.87 centimetres. This corresponds to a frequency of 3037.4 MHz (3.0374 GHz). The frequency of a pulse radar level transmitter may be 26 GHz or 26 x 108 metres per second. The wavelength is 1.15 centimetres. The electromagnetic waves have an electrical vector E and a magnetic vector B that are perpendicular to each other and perpendicular to the direction of the wave. This will be discussed and illustrated further in the section on polarization. The electrical vector has the major influence on radar applications. λ direction of wave amplitude co (µ The velocity of an electromagnetic wave is the product of the frequency and the wavelength. Fig 2.1 13
  • 20. The Electromagnetic spectrum 10 8 10 7 10 6 10 5 10 4 10 3 10 2 electric waves 10 1 10 2 10 1 10 0 10 -1 10 -2 10 -3 10 -4 radio waves 10 3 10 4 10 5 10 6 10 7 10 8 infra 10 9 10 10 10 11 10 12 3m 0.3 m 3 cm 3 mm 100 MHz 1 GHz 10 GHz 100 GHz The microwave frequencies of the electromagnetic spectrum. Radar level transmitters range between 5.8 GHz (5.2cm) and 26 GHz (11.5mm) 14
  • 21. 2. Physics of radar 10 -5 10 -6 red 10 13 10 -7 10 -8 ultra violet 10 14 10 15 10 16 10 -9 10 -10 10 -11 10 -12 10 -13 10 -14 10 -15 10 -16 X rays 10 17 10 18 m gamma rays 10 19 10 20 10 21 10 22 10 23 10 24 Hz Fig 2.2 Electromagnetic spectrum. All electromagnetic waves travel at the speed of light in free space. This spectrum shows the range of frequencies and wavelengths from electric waves to gamma rays 15
  • 22. Permittivity In electrostatics, the force between two charges depends upon the magnitude and separation of the charges and the composition of the medium between the charges. Permittivity ε is the property of the medium that effects the magnitude of the force. The higher the value of the permittivity, the lower the force between the charges. The value of the permittivity of free space (in a vacuum) εo, is calculated indirectly and empirically to be: 8.854 x 10-12 farad / metre. Relative permittivity or dielectric constant εr The ratio of the permittivity of a medium to the permittivity of free space is a dimensionless property called ‘relative permittivity’ or ‘dielectric constant’. For example, at 20° C the relative permittivity of air is close to that of a vaccum and is only about 1.0005 whereas the relative permittivity of water at 20° C is about 80. (Dielectric constant is also widely known as DK.) The value of the dielectric constant of the product being measured is very important in the application of radar to level measurement. In non-conductive products, some of the microwave energy will pass through the product and the rest will be reflected off the surface. This feature of microwaves can be used to advantage or, in some circumstances, it can create a measurement problem. 16 Permeability µ and relative permeability µr The magnetic vector, B, of an electromagnetic wave also has an influence on the velocity of electromagnetic waves. However, this influence is negligible when considering the velocity in gases and vapours which are non-magnetic. The relative permeability of the product being measured has no significant effect on the reflected signal when compared with the effects of the relative permittivity or dielectric constant. For the non-magnetic gases above the product being measured, the value of the relative permeability, µr = 1. Frequency, velocity and wavelength As we have already stated, the frequency (f), velocity (c) and wavelength (λ) of the electromagnetic waves are related by the equation c = f x λ. The frequency remains uninfluenced by changes in the propagation medium. However, the velocity and wavelength can change depending on the electrical properties of the medium in which they are travelling. The speed of propagation can be calculated using equation 2.3. c = co (µ x ε ) r c co µr εr r [Eq. 2.3] velocity of electromagnetic wave in the medium in metres/second velocity of electromagnetic waves in free space the relative permeability (µ medium / µo) the relative permittivity
  • 23. 2. Physics of radar Changes in the wavelength and velocity of microwaves are apparent in certain radar level applications. Changes in temperature, pressure and gas composition have a small effect on the running time of microwaves because the dielectric constant of the propagation medium is altered to a greater or lesser extent. This is discussed in detail later. Radar level transmitters can be used to measure conductive liquids through low dielectric ‘windows’ such as glass, polypropylene and PTFE. The optimum thickness of the low dielectric window is a half wavelength or multiple of half wavelength. For example, polypropylene has a dielectric constant εr of 2.3 and the half wavelength at a frequency of 5.8 GHz is 17 mm compared with a half wavelength of about 26 mm in a vacuum. It follows that the speed of Empty vessel: large echo from metal bottom microwaves in polypropylene is about two thirds of the speed in air. As with low dielectric windows, non-conductive, low dielectric constant liquids may absorb more power than they reflect from the surface. The velocity of the microwaves within the liquid is slower than in the vapour space above. For example, if there is about 0.5 metres of solvent in the bottom of a metallic vessel, a radar level transmitter may see a larger echo from the vessel bottom than from the product. This large echo will appear to be further away than it really is because the running time within the solvent is slower. For this reason, special considerations must be made within the echo processing software to ensure that the radar follows the solvent level and does not follow the vessel bottom as it apparently moves away! As the vessel fills with solvent two echoes are received. The echo from the vessel bottom appears further away because the running time of the microwaves in solvent is slower solvent echo Fig 2.3 - Effect of dielectric constant on the running time of a microwave radar 17
  • 24. The same effect can be experienced when looking at interface detection using guided microwave level transmitters to detect oil and water or solvent and aqueous based liquids. Fig 2.4 Oil/water interface detection using a guided microwave level transmitter. Note that the water echo has a reduced amplitude and appears to be further away. The running time of microwaves in oil is slower than in air reference echo (water without oil) oil echo water echo Effects on the propagation speed of microwaves Microwave radar level transmitters can be applied almost universally because, as a measurement technique, they are virtually unaffected by process temperature, temperature gradient, vacuum and normal pressure variations, gas or vapour composition and movement of the propagation medium. However, changes in these process conditions do cause slight variations in the propagation speed because the dielectric constant of the propagation medium is altered. Calculating the propagation speed of microwaves The temperature, pressure and the gas composition of the vapour space all have an effect on the dielectric constant of the propagation medium through which the microwaves must travel. This in turn affects the propagation speed or running time of the instrument. 18 The dielectric constant or relative permittivity can be calculated as follows : εr = 1 + (εrN - 1) x θN x P θ x PN [Eq. 2.4] εr εrN calculated dielectric constant (relative permittivity) dielectric constant of gas/vapour under normal conditions (temperature 273 K, pressure 1 bar absolute) θN PN θ P temperature under normal conditions, 273 Kelvin pressure under normal conditions, 1 bar absolute process temperature in Kelvin process pressure in bar absolute
  • 25. 2. Physics of radar From equation 2.4 and equation 2.3, we can calculate the percentage error caused by variations in the dielectric constant of different gases and vapours and the relative effects of changes in process temperature and pressure. Gases and vapours By definition, the dielectric constant in a vacuum is equal to 1.0. The dielectric constants of the gases and vapours that may be present above the product differ but they have only a very small effect on the accuracy of radar. Radar level transmitters are usually calibrated in air. For this reason, the following tables show 1. Dielectric constant of different gases at normal temperature and pressure (273K, 1 Bar A) 2. Percent error in the running time in the gases compared with air Table 2.1 The dielectric constants under normal conditions, εrN and the error caused by the dielectric constant of typical process gases under normal conditions % Error from air (at normal temperature and pressure) Vacuum Air Argon Ammonia / NH 3 Hydrogen Bromide HBr Hydrogen Chloride HCl Carbon Monoxide / CO Carbon Dioxide / C0 2 Ethane / C 2 H6 Ethylene / C 2H4 Helium ε rN (dielectric constant at normal conditions) 1.0000 1.000633 1.000551 1.006976 1.002994 1.004078 1.000692 1.000985 1.001503 1.001449 1.000072 Hydrogen / H 2 Methane / CH 4 Nitrogen / N 2 Oxygen / O 2 1.000275 1.000878 1.000576 1.000530 + 0.0179 - 0.0122 + 0.00285 + 0.0052 Gas / Vapour + 0.0316 0.0 + 0.0041 + 0.3154 - 0.1178 - 0.1717 - 0.00295 - 0.0176 - 0.0434 - 0.0407 + 0.0280 19
  • 26. Temperature High temperature or large temperature gradients have very little effect on the transit time of microwaves within an air or vapour space. At a temperature of 2000° C the variation is only 0.026% from the measurement value at 0° C. Radar level transmitters with air or nitrogen gas cooling are used on molten iron and steel applications. 0.03 0.025 % error 0.02 0.015 0.01 0.005 0.0 0 250 500 750 1000 1250 1500 1750 2000 Temperature in ° C Fig 2.5 Temperature effect on radar measurement of air at a constant pressure of 1 BarA 20
  • 27. 2. Physics of radar Pressure Pressure does have a small but more significant influence on the velocity of electromagnetic waves. At a pressure of 30 Bar, the error is only 0.84%. However this becomes more significant and at a pressure of 100 Bar there is a velocity change of 2.8%. If the pressure is varying constantly between atmospheric pressure and 100 Bar, the velocity variations can be compensated using a pressure transmitter. 10 % error 8 6 4 2 0 0 50 100 150 200 250 300 350 400 Pressure in Bar (absolute) Fig 2.6 The influence of pressure on radar measurement in air at a constant temperature of 273 K 21
  • 28. Waveguides, stilling tubes & bypass tubes In the preceding equations, we have assumed that the microwaves are travelling in ‘free space’ in a vacuum. However, in practice the proximity of metallic vessel walls and other structures will have an influence on the propagation velocity of the microwaves. This is particularly true when microwave radar level transmitters are fitted inside bypass tubes or stilling tubes or when a horn antenna is fitted with a waveguide extension. When microwaves are propagating within a metallic tube the running time appears to slow down because the microwaves travel further bouncing off the inside wall of the tube and currents are set up on the inside surface of the tube. This effect is discussed in more detail in the chapters on antennas and mechanical installations. The waveguide effect can be compensated during calibration and the use of stilling tubes and bypass tubes can be beneficial in some level applications. Electromagnetic waves exhibit the same properties as light. · · · Reflection Polarization Diffraction · · Refraction Interference Reflection of electromagnetic waves Conductive products Using a spark gap transmitter, Heinrich Hertz demonstrated that electromagnetic waves could be reflected off metallic objects and objects with a relatively high dielectric constant. In the same way, radar can easily measure conductive aqueous liquids such as acids and caustic and other conductive products ranging from molten metal to saturated spent grain in the brewing process. When microwaves from a radar hit a conductive surface the electrical field E is short circuited. The resultant current in the conductive product causes the microwaves to be re-transmitted or reflected from the surface. 22 Radar level transmitters have no problem in measuring conductive liquids and solids because the microwaves with frequencies between 5.8 GHz and 26GHz are readily reflected off a conductive surface producing relatively large echoes. Non-conductive products If a liquid or solid is non-conductive, the value of the dielectric constant (relative permittivity εr) becomes more important. The theoretical amount of reflection at a dielectric layer can be calculated using equation 2.5
  • 29. 2. Physics of radar W1 Transmitted power: W2 Reflected power: Dielectric constant: εr Then the percentage of reflected power at the dielectric layer, Π = 1- 4 x εr (1 + ε ) 2 r W2 Π = [Eq. 2.5] W1 Typical examples are as follows: Acetone Solvent with a dielectric constant, εr = 20 Toluene Solvent with a low dielectric consta t n, εr = 2.4 Π = 1- (2.4) 4x (1 + Π = 1- 2 (2.4)) 4.46% power is reflected 4x (1 + ( 20 ) (20) 2 ) 40 % power is reflected Π x 100% power reflected 100 80 60 40 20 0 0 10 20 30 40 50 60 70 80 Dielectric constant, εr Fig 2.7 Reflected radar power depends upon the dielectric constant of the product being measured 23
  • 30. In radar level measurement the reflected energy from a product surface becomes more critical at a dielectric constant (εr) of less than 5. The following graph shows this important region. Π x 100% power reflected 20 15 10 5 0 1.0 1.5 2.0 3.0 2.5 3.5 Dielectric constant, εr 4.0 4.5 5.0 Fig 2.8 Reflected radar power depends upon the dielectric constant of the product being measured. This graph shows the critical region where care must be taken over choice of radar antenna 0 Loss L, dB - 10 - 20 - 40 - 60 3.0 3.5 2.5 Dielectric constant, εr Fig 2.9 Reflection loss in dB: loss L = 10 log Π 1.0 1.5 2.0 4.0 4.5 5.0 Most electrically conductive products or products with a dielectric constant of more than 1.5 can be measured using microwave radar level transmitters. Stilling tubes can be used to concentrate the microwaves for lower dielectric constant products. 24
  • 31. 2. Physics of radar Polarization Electromagnetic waves have an electrical vector E and magnetic vector B that are in phase but perpendicular to each other. The direction of propagation of the waves is perpendicular to the electrical and magnetic vectors as shown in the diagram below. Polarization defines the orientation of the electromagnetic waves and refers to the direction of the electrical vector E. Most process radar level transmitters exhibit linear polarization as in the dia- gram. The direction of the linear polarization is set by the orientation of the signal coupler from the microwave module. The properties of the polarization of microwaves can be important in the application of radar to level measurement. In television and microwave communications, linear polarization is also referred to as horizontal or vertical polarization depending on the relative orientation of the aerials or antennas. E direction of wave B Fig 2.10 Diagram showing linear polarization and the relative orientation of the electric vector E, the magnetic vector B and the direction of propagation of the microwaves 25
  • 32. Another form of polarization is elliptical polarization. A specific form of elliptical polarization is circular polarization where the electrical vector E and magnetic vector B rotate through 360° within the space of a single wavelength, when a linear or circular polarized signal is reflected the direction of polarization is reversed. With circular polarization it is possible to use the reversal of polarization to distinguish between a direct echo and an echo that has made two reflections. Circular polarization can also be used in search radars to separate the reflections from aircraft or ships from interference echoes from rain. The almost spherical shape of the rain drops causes a definite reversal of polarization which can be easily rejected by the receiving antenna. However, the scattered reflections from the ship or aircraft provide roughly equal amounts of reversed and un-reversed energy that enables detection. λ Fig 2.11 Circular polarization involves rotation of the electrical and magnetic vectors through 360° within a wavelength 26
  • 33. 2. Physics of radar The linear polarization that is common with process radar level transmitters can be used to minimise the effects of false echo returns from the internal structure of a process vessel. These false echoes could be reflected from probes, welds, agitators and baffles. In some applications, the effect of false echoes within a vessel can be significantly reduced by rotating the radar in the connection flange or boss. The principle is illustrated below and detailed in the section on mechanical installations in Chapter 6. Polarization can be used to reduce the amplitude of false echoes E Direction of wave B Large echo Fig 2.12 If a metallic or high dielectric object is orientated in the same plane as the electrical vector of the polarized microwaves, the radar level transmitter will receive a large amplitude echo E Direction of wave Small echo B Fig 2.13 If the same object is orientated at right angles to the plane of the electrical vector, the received echo will have a smaller amplitude 27
  • 34. Diffraction Beam angle is often discussed in relation to radar transmitters. This can give the impression that the radar antenna can direct a finely focused beam towards the target. Unfortunately this is not the case. In practice, although they are designed to produce a directed beam, a radar antenna radiates some energy in all directions. As well as the main lobe which accounts for most of the radiated power, there are also weaker side lobes of energy. This phenomenon is caused, in part, by diffraction. In addition to this, destructive interference causes the null points or notches that form the characteristic side lobes. Chapter 5 provides a detailed explanation of beam angles, side lobes and types of antennas. side lobes main lobe antenna Fig 2.14 The lobe structure of antenna beams is caused by diffraction and destructive interference Refraction In the same way as light is refracted at an air/glass or air/water interface, microwaves are refracted when they encounter a change in dielectric. This could be a low dielectric window (PTFE/glass/polypropylene) or a nonconductive low dielectric liquid such as a solvent. reflected energy The angle of refraction depends on the angle of the incident wave and also on the ratio of the dielectric constants at the interface. It is possible to utilise the refractive properties of electromagnetic waves to construct a dielectric lens that will focus microwaves. a a Fig 2.15 Refraction & reflection microwave interface dielectric window / product refracted energy 28 B
  • 35. 2. Physics of radar Interference - Phase Problematic interference effects are caused primarily by the inadvertent mixing of signals that are out of phase. The microwave signals have a sinusoidal waveform. Phase angle 45° Fig 2.16 In this illustration both of the sine waves have an identical frequency and amplitude but the second wave has a 45° phase lag Interference can be ‘constructive’ where in-phase signals produce a signal with a higher amplitude or it can be destructive where signals that are 180° out of phase effectively cancel each other out. signals in-phase constructive interference 180° out of phase destructive interference Fig 2.17 Illustration of constructive and destructive interference 29
  • 36. Interference Microwaves can manifest interference effects in exactly the same way as light. Potentially this can cause measurement problems. The causes of interference should be understood and avoided by design and installation considerations. The wrong choice of antenna, installation of an antenna up a nozzle, positioning transmitters too close to vessel walls or other obstructions can all lead to interference of the signal. The chapter on mechanical installation should help a radar level user to avoid this potential problem. However, we use destructive interference to our advantage when we apply pulse radar level measurement through a low dielectric ‘window’ to measure conductive or high dielectric liquids. A + = C B’ B B” Fig 2.18 Interference caused by positioning an antenna too close to the vessel wall. If a radar level transmitter is installed too close to the vessel wall it is possible that interference will occur. With indirect reflection A B’ B’’ C, the phase may be altered by 180° when compared with the direct reflection A B C. For this reason the microwaves may partially cancel out due to destructive interference 30
  • 37. 2. Physics of radar The thickness of the dielectric window must be a half wavelength of the window material. When the half wavelength is used, there is destructive interference between the reflection off the top surface of the window and the reflection off the internal second surface of the window. There is a 180° phase shift between these reflections and they cancel each emitted wave plastic vessel ceiling other out. This type of installation is explained more fully in Chapter 6 on the mechanical installations of radar level transmitters together with a table showing the optimum thickness of most important plastics and glasses which are suitable for penetration with radar sensors. reflection with phase shift from top surface reflection without phase shift from internal surface D emitted wave reflection with phase shift off top surface of window reflection without phase shift off internal face of window Fig 2.19 Destructive interference is a benefit when using pulse radar to measure through a low dielectric window. The reflection from the top surface and the reflection from the internal second surface cancel each other if the thickness is a half wavelength 31
  • 38. Contents Foreword Acknowledgement Introduction ix xi xiii Part I 1. History of radar 2. Physics of radar 3. Types of radar 1. CW-radar 2. FM - CW 3. Pulse radar 1 13 33 33 36 39 Part II 4. Radar level measurement 1. FM - CW 2. PULSE radar 3. Choice of frequency 4. Accuracy 5. Power 47 48 54 62 68 74 5. Radar antennas 1. Horn antennas 2. Dielectric rod antennas 3. Measuring tube antennas 4. Parabolic dish antennas 5. Planar array antennas Antenna energy patterns 77 81 92 101 106 108 110 6. Installation A. Mechanical installation 1. Horn antenna (liquids) 2. Rod antenna (liquids) 3. General consideration (liquids) 4. Stand pipes & measuring tubes 5. Platic tank tops and windows 6. Horn antenna (solids) B. Radar level installation cont. 1. safe area applications 2. Hazardous area applications 115 115 115 117 120 127 134 139 141 141 144
  • 39. 3. Types of radar 1a. CW, continuous wave radar In continuous wave or CW Radar, a continuous unmodulated frequency is transmitted and echoes are received from the target object. If the target object is stationary, the frequency of the return echoes will be the same as the transmitted frequency. The range of the object cannot be measured. However, the frequency of the return signal from a moving object is changed depending on the speed and direction of the object. This is the well known ‘doppler effect’. The doppler effect is apparent when the siren note of an emergency vehicle changes as it speeds past a pedestrian. The pitch of the siren note is higher as it approaches the listener and lower as it recedes. The doppler effect is also used by astronomers to monitor the expansion of the Universe. By measuring the ‘red shift’ of the spectrum of distant stars and galaxies the rate of expansion can be measured and the age of distant objects can be estimated. In the same way, when an object that has been illuminated by a CW Radar approaches the transmitter, the frequency of the return signal will be higher than the transmitted frequency. The echo frequency will be lower if the object is moving away. yv elocit rece requ ived f tv targe f + f dp ency t itted m trans ave yf w uenc t, lengt hλ freq Fig 3.1 CW radar uses doppler shift to derive speed measurement 33
  • 40. In Fig 3.1, the aircraft is travelling towards the CW radar. Therefore the received frequency is higher than the transmitted frequency and the sign of fdp is positive. If the aircraft was travelling away from the radar at the v = λ x fdp 2 = same speed, the received frequency would be ft - fdp. The velocity of the target in the direction of the radar is calculated by equation 3.1 c x fdp c v ft 2 x ft fdp [Eq. 3.1] ft+fdp is the velocity of microwaves is the target velocity is the frequency of the transmitted signal is the doppler beat frequency which is proportional to velocity is received frequency. The sign of fdp depends upon whether the target is closing or receding 1b. CW wave-interference radar or bistatic CW radar We have already mentioned that CW radar was used in early radar detection experiments such as the famous Daventry experiment carried out by Robert Watson - Watt and his colleagues. In this case, the transmitter and receiver were separated by a considerable distance. A moving object was detected by the receiver because there was interference between the fre- quency received directly from the transmitter and the doppler shifted frequency reflected off the target object. Although the presence of the object is detected, the position and speed cannot be calculated. In essence, this is what happens when a low flying aircraft interferes with the picture on a television screen. See Fig 3.2. 1c. Multiple frequency CW radar Standard continuous wave radar is used for speed measurement and, as already explained, the distance to a stationary object can not be calculated. However, there will be a phase shift between the transmitted signal and the return signal. If the starting position of the object is known, CW radar could be used to detect a change in position of up to half wavelength (λ/2) of the transmitted wave by measuring the phase shift of the echo signal. Although further movement could be detected, the range 34 would be ambiguous. With microwave frequencies this means that the useful measuring range would be very limited. If the phase shifts of two slightly different CW frequencies are measured the unambiguous range is equal to the half wavelength (λ/2) of the difference frequency. This provides a usable distance measurement device. However, this technique is limited to measurement of a single target. Applications include surveying and automobile obstacle detection.
  • 41. transmitter transmitted signal direct television interference reflected signal (doppler shift) Fig 3.2 The effect of low flying aircraft on television reception is similar to the method of detection by CW wave-interference radar transmitted signal indirect target 3. Types of radar 35
  • 42. 2. FM-CW, frequency modulated continuous wave radar If the distance to the target is R, and c is the speed of light, then the time taken for the return journey is:- 2xR c ∆t = [Eq. 3.2] We can see from Fig. 3.3 that if we know the linear rate of change of the transmitted signal and measure the difference between the transmitted and received frequency fd, then we can calculate the time ∆t and hence derive the distance R. frequency Single frequency CW radar cannot be used for distance measurement because there is no time reference mark to gauge the delay in the return echo from the target. A time reference mark can be achieved by modulating the frequency in a known manner. If we consider the frequency of the transmitted signal ramping up in a linear fashion, the difference between the transmitting frequency and the frequency of the returned signal will be proportional to the distance to the target. cy en u eq r df itte m ns tra re ∆t fd c e eiv df re e qu nc y ∆t = 2xR c time Fig 3.3 The principle of FM - CW radar 36
  • 43. 3. Types of radar In practice, the FM - CW signal has to be cyclic between two different frequencies. Radio altimeters modulate between 4.2 GHz and 4.4 GHz. Radar level transmitters typically modulate between about 9 GHz and 10 GHz or 24 GHz and 26 GHz. The cyclic modulation of FM - CW radar transmitter takes different forms. These are sinusoidal, saw tooth or triangular wave forms. FM - CW wave forms transmitted frequency received frequency frequency Fig 3.4 Sine wave Commonly used on aircraft radio altimeters between 4.2 and 4.4 GHz 4.4GHz time 4.2GHz frequency Fig 3.5 Triangular wave Used on FM - CW radar transmitters time frequency Fig 3.6 Saw tooth wave 10 GHz 9 GHz time Most commonly used on most FM - CW process radar level transmitters 37
  • 44. If we look at a triangular wave form we can see that there is an interruption in the output of the difference frequency , fd. In practice, the received signal is heterodyned with part of the transmitted frequency to produce the difference frequency which has a posi- tive value independent of whether the modulation is increasing or decreasing. The diagram below makes the assumption that the target distance is not changing. If the target is moving, there will be a doppler shift in the difference frequency. frequency time difference frequency fd time Fig 3.7 & 3.8 The change in direction between the ramping up and down of the frequency creates a short break in the measured value of the difference frequency. This has to be filtered out. The transmitted frequency is represented by the red line and the received frequency is represented by the dark blue line. The difference frequency is shown in light blue on the bottom graph 38
  • 45. 3. Types of radar 3. Pulse radar a. Basic pulse radar Pulse radar is and has been used widely for distance measurement since the very beginnings of radar technology. The basic form of pulse radar is a pure time of flight measurement. Short pulses, typically of millisecond or nansecond duration, are transmitted and the transit time to and from the target is measured. The pulses of a pulse radar are not discrete monopulses with a single peak of electromagnetic energy, but are in fact a short wave packet. The number of waves and length of the pulse depends upon the pulse duration and the carrier frequency that is used. These regularly repeating pulses have a relatively long time delay between them to allow the return echo to be received before the next pulse is transmitted. t τ 3rd pulse 2nd pulse 1st pulse Transmitted pulses Fig 3.9 Basic pulse radar The inter pulse period (the time between successive pulses) t is the inverse of the pulse repetition frequency fr or PRF. The pulse duration or pulse width, τ, is a fraction of the inter pulse period. The inter pulse period t effectively defines the maximum range of the radar. Example The pulse repetition frequency (PRF) is defined as fr = 1 t If the pulse period t is 500 microseconds, then the pulse repetition frequency is two thousand pulses per second. In 500 microseconds, the radar pulses will travel 150 kilometres. Considering the return journey of an echo reflected off a target, this gives a maximum theoretical range of 75 kilometres. If the time taken for the return journey is T, and c is the speed of light, then the distance to the target is R= Txc 2 [Eq. 3.3] 39
  • 46. b. Pulse doppler radar The pulses transmitted by a standard pulse radar can be considered as a very short burst of continuous wave radar. There is a single frequency with no modulation on the signal for the duration of the pulse. If the frequency of the waves of the transmitted pulse is ft and the target is moving towards the radar with velocity v, then, as with the CW radar already described, the frequency of the return pulse will be ft + fdp , where fdp is the doppler beat frequency. Similarly, the received frequency will be ft - fdp if the target is moving away from the radar. Therefore, a pulse doppler radar can be used to measure speed, distance and direction. The ability of the pulse doppler radar to measure speed allows the system to ignore stationary targets. This is also commonly called ‘moving target indication’ or MTI radar. In general, an MTI radar has accurate range measurement but imprecise speed measurement, whereas a pulse doppler radar has accurate speed measurement and imprecise distance measurement. 40 The velocity of the target in the direction of the radar is calculated in equation 3.4: c = c x fdp λ x fdp = 2 x ft 2 [Eq. 3.4] This is the same calculation as for CW radar. The distance to the target is calculated by the transit time of the pulse, equation 3.3. R = Txc 2 [Eq. 3.3] As well as being used to monitor civil and military aircraft movements, pulse doppler radar is used in weather forecasting. A doppler shift is measured within storm clouds which can be distinguished from general ground clutter. It is also used to measure the extreme wind velocities within a tornado or ‘twister’.
  • 47. R Fig 3.10 Pulse doppler radar provides target speed, distance and direction f t + f dp ft Pulse doppler radar 3. Types of radar 41
  • 48. c. Pulse compression and ‘Chirp’ radar frequency With pulse radar, a shorter pulse duration enables better target resolution and therefore higher accuracy. However, a shorter pulse needs a significantly higher peak power if the range performance has to be maintained. If there is a limit to the maximum power available, a short pulse will inevitably result in a reduced range. With limited peak power, a longer pulse duration, τ , will provide more radiated energy and therefore range but (with a standard pulse radar) at the expense of resolution and accuracy. Pulse compression within a ‘Chirp’ radar is a method of achieving the accuracy benefits of a short pulse radar together with the power benefits of using a longer pulse. Essentially, Chirp radar is a cross between a pulse radar and an FM - CW radar. f1 f2 time t2 t1 amplitude τ time Fig 3.11 Chirp radar wave form. Chirp is a cross between pulse and FM - CW radar 42
  • 49. 3. Types of radar Each pulse of a Chirp radar has linear frequency modulation and a constant amplitude. The echo pulse is processed through a filter that compresses the echo by creating a time lag that is inversely proportional to the frequency. Therefore, the low frequency that arrives first is slowed down the most and the subsequent higher frequencies catch up producing a sharper echo signal and improved echo resolution. Time lag Filter Frequency Long frequency modulated echo pulse Compressed signal Fig 3.12 Pulse compression of chirp radar echo signal Pulse compression of chirp radar echo signal Another method of echo compression uses binary phase modulation where the transmitted signal is specially encoded with segments of the pulse either in phase or 180° out of phase. The return echoes are decoded by a filter that produces a higher amplitude and compressed signal. The name ‘Chirp’ radar comes from the short rapid change in frequency of the pulse which is analogous to the chirping of a bird song. The above methods of radar detection are used widely in long range distance or speed measurement. In the next chapter we look at which of these methods can be applied to the unique problems involved in measuring liquid or solid levels within process vessels and silos. 43
  • 50. Part II Radar level measurement Radar antennas Radar level installations 45
  • 51. Contents Foreword Acknowledgement Introduction ix xi xiii Part I 1. History of radar 2. Physics of radar 3. Types of radar 1. CW-radar 2. FM - CW 3. Pulse radar 1 13 33 33 36 39 Part II 4. Radar level measurement 1. FM - CW 2. PULSE radar 3. Choice of frequency 4. Accuracy 5. Power 47 48 54 62 68 74 5. Radar antennas 1. Horn antennas 2. Dielectric rod antennas 3. Measuring tube antennas 4. Parabolic dish antennas 5. Planar array antennas Antenna energy patterns 77 81 92 101 106 108 110 6. Installation A. Mechanical installation 1. Horn antenna (liquids) 2. Rod antenna (liquids) 3. General consideration (liquids) 4. Stand pipes & measuring tubes 5. Platic tank tops and windows 6. Horn antenna (solids) B. Radar level installation cont. 1. safe area applications 2. Hazardous area applications 115 115 115 117 120 127 134 139 141 141 144
  • 52. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 47 4. Radar level measurement The benefits of radar as a level measurement technique are clear. Radar provides a non-contact sensor that is virtually unaffected by changes in process temperature, pressure or the gas and vapour composition within a vessel. In addition, the measurement accuracy is unaffected by changes in density, conductivity and dielectric constant of the product being measured or by air movement above the product. The practical use of microwave radar for tank gauging and process vessel level measurement introduces an interesting set of technical challenges that have to be mastered. If we consider that the speed of light is approximately 300,000 kilometres per second. Then the time taken for a radar signal to travel one metre and back takes 6.7 nanoseconds or 0.000 000 006 7 seconds. How is it possible to measure this transit time and produce accurate vessel contents information? Currently there are two measurement techniques in common use for process vessel contents measurement. They are frequency modulated continuous wave (FM - CW) radar and PULSE radar In this chapter we explain FM - CW and PULSE radar level measurement and compare the two techniques. We discuss accuracy and frequency considerations and explore the technical advances that have taken place in recent years and in particular two wire, loop powered transmitters. 47
  • 53. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 48 FM-CW, frequency modulated continuous wave The FM - CW radar measurement technique has been in use since the 1930's in military and civil aircraft radio altimeters. In the early 1970's this method was developed for marine use measuring levels of crude oil in supertankers. Subsequently, the same technique was used for custody transfer level measurement of large land based storage vessels. More recently, FM CW transmitters have been adapted for process vessel applications. FM - CW, or frequency modulated continuous wave, radar is an indirect method of distance measurement. The transmitted frequency is modulated between two known values, f1 and f2, and the difference between the transmitted signal and the return echo signal, fd, is measured. This difference frequency is directly proportional to the transit time and hence the distance. (Examples of FM - CW radar level transmitters modulation frequencies are 8.5 to 9.9 GHz, 9.7 to 10.3 GHz and 24 to 26 GHz). The theory of FM - CW radar is simple. However, there are many practical problems that need to be addressed in process level applications. An FM - CW radar level transmitter requires a voltage controlled oscillator, VCO, to ramp the signal between the two transmitted frequencies, f1 and f2. It is critical that the frequency sweep is controlled and must be as linear as possible. A linear frequency modulation is achieved either by accurate frequency measurement circuitry with closed loop regulation of the output or by careful linearisation of the VCO output including temperature compensation. f2 frequency Transmitted signal fd ∆t Received signal f1 t1 time Fig 4.1 The FM - CW radar technique is an indirect method of level measurement. fd is proportional to ∆t which is proportional to distance 48
  • 54. f(t) Directional Coupler Signal sampling and Fast Fourier transforms (FFT) Frequency Measurement Intermediate frequency Amplifier Filter Mixer f (t + Dt) Directional Coupler f (t + Dt) Fig 4.2 Typical block diagram of FM - CW radar. A very accurate linear sweep is required Signal Microprocessor Front end control function Linear ramp generator Voltage Control V(t) Linear sweep control loop Voltage Controlled Oscillator VCO f(t) 4. Radar level measurement 49
  • 55. radar_applied_to_level_rb.qxd 15.01.2007 FM - CW block diagram (Fig 4.2) The essential component of a frequency modulated continuous wave radar is the linear sweep control circuitry. A linear ramp generator feeds a voltage controller which in turn ramps up the frequency of the Voltage Controlled Oscillator. A very accurate linear sweep is required. The output frequency is measured as part of the closed loop control. The frequency modulated signal is directed to the radar antenna and 18:46 Seite 50 hence towards the product in the vessel. The received echo frequencies are mixed with a part of the transmission frequency signal. These difference frequencies are filtered and amplified before Fast Fourier Transform (FFT) analysis is carried out. The FFT analysis produces a frequency spectrum on which the echo processing and echo decisions are made. Pic 2 Typical glass lined agitated process vessel. A radar must be able to cope with various false echos from agitatior blades and baffles Simple storage applications usually have a large surface area with very little agitation, no significant false echoes from the internal structure of the tank and relatively slow product movement. These are the ideal conditions for which FM - CW radar was originally developed. However, in process vessels there is more going on and the problems become more challenging. 50 Low amplitude signals and false echoes are common in chemical reactors where there is agitation and low dielectric liquids. Solids applications can be troublesome because of the internal structure of the silos and undulating product surfaces which creates multiple echoes. An FM - CW radar level sensor transmits and receives signals simultaneously.
  • 56. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 51 4. Radar level measurement fd1, -f d2 , -fd3, -fd4, -fd5 f2 f1 t1 Transmitted signal Real echo signal False echo signals Fig 4.3a FM - CW radar level transmitters in an active process vessel In an active process vessel, the various echoes are received as frequency differences compared with the frequency of the transmitting signal. These frequency difference signals are received by the antenna at the same time. The amplitude of the real echo signals are small compared with the transmitted signal. A false echo from the end of the antenna may have a significantly higher amplitude than the real level echo. The system needs to separate and identify these simultaneous signals before processing the echoes and making an echo decision. The separation of the various received echo frequencies is achieved using Fast Fourier Transform (FFT) analysis. This is a mathematical proce- dure which converts the jumbled array of difference frequencies in the time domain into a frequency spectrum in the frequency domain. The relative amplitude of each frequency component in the frequency spectrum is proportional to the size of the echo and the difference frequency itself is proportional to the distance from the transmitter. The Fast Fourier Transform requires substantial processing power and is a relatively long procedure. It is only when the FFT calculations are complete that echo analysis can be carried out and an echo decision can be made between the real level echo and a number of possible false echoes. 51
  • 57. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 52 Mixture of frequencies received by FM - CW radar Signal amplitude fd1, fd2, fd3, fd4, fd5 etc combined Fig 4.3b combined echo frequencies are received simultaneously Signal amplitude Combination of mixed difference frequencies received by FM - CW radar Individual difference frequencies fd1, f d2 , fd3, are shown Fig 4.3c The individual frequencies must be separated from the simultaneously received jumble of frequencies 52
  • 58. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 53 4. Radar level measurement amplitude Frequency spectrum echoes Each echo is within an envelope curve of frequencies frequency Fig 4.4 FM - CW frequency spectrum after Fast Fourier transform. The Fast Fourier transform algorithm converts the signals from the time domain into the frequency domain. The result is a frequency spectrum of the difference frequencies. The relative amplitude of each frequency component in the spectrum is proportional to the size of the echo and the difference frequency itself is proportional to the distance from the transmitter. The echoes are not single frequencies but a span of frequencies within an envelope curve Complex process vessels and solids applications can prove too difficult for some FM - CW radar transmitters. Even a simple horizontal cylindrical tank can pose a serious problem. This is because a horizontal tank produces many large multiple echoes that are caused by the parabolic effect of the cylindrical tank roof. Sometimes the amplitudes of the multiple echoes are higher than the real echo. The processors that carry out the FFT analysis are swamped by different amplitude signals across the dynamic range all at the same time. As a result, the FM - CW radar cannot identify the correct echo. As we shall see, this problem does not affect the alternative pulse radar technique. 53
  • 59. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 54 PULSE radar level transmitters Pulse radar level transmitters provide distance measurement based on the direct measurement of the running time of microwave pulses transmitted to and reflected from the surface of the product being measured. Pulse radar operates in the time domain and therefore it does not require the Fast Fourier transform (FFT) analysis that characterizes FM CW radar. As already discussed, the running time for a distance of a few metres is measured in nanoseconds. For this reason, a special time transformation pro- cedure is required to enable these short time periods to be measured accurately. The requirement is for a ‘slow motion’ picture of the transit time of the microwave pulses with an expanded time axis. By slow motion we mean milliseconds instead of nanoseconds. Pulse radar has a regular and periodically repeating signal with a high pulse repetition frequency (PRF). Using a method of sequential sampling, the extremely fast and regular transit times can be readily transformed into an expanded time signal. Fig 4.5 Pulse radar operates purely within the time domain. Millions of pulses are transmitted every second and a special sampling technique is used to produce a ‘time expanded’ output signal 54
  • 60. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 55 4. Radar level measurement To illustrate this principle, consider the sine wave signal in Fig 4.6. It is a regular repeating signal with a period of T1. If the amplitude (voltage value) of the output of the sine wave is sampled into a memory at a time period T2 which is slightly longer than T1, then a time expanded version of the original sine wave is produced as an output. The time scale of the expanded output depends on the difference between the two time periods T1 and T2. T1 Periodic Signal (sine wave) Sampling signal T2 Expanded time signal Fig 4.6 The principle of sequential sampling with a sine wave as an example. The sampling period, T2, is very slightly longer than the signal period, T1. The output is a time expanded image of the original signal A common example of this principle is the use of a stroboscope to slow down the fast periodic movements of rotating or reciprocating machinery. Fig 4.7 shows how the principle of Periodic Signal (radar echoes) sequential sampling is applied to pulse radar level measurement. The example shown is a VEGAPULS transmitter with a microwave frequency of 5.8 GHz. T1 Emission pulse Echo pulse T2 Sampling signal Fig 4.7 Sequential sampling of a pulse radar echo curve. Millions of pulses per second produce a periodically repeating signal. A sampling signal with a slightly longer periodic time produces a time expanded image of the entire echo curve 55
  • 61. radar_applied_to_level_rb.qxd 15.01.2007 18:46 This periodically repeating signal consists of the regular emission pulse and one or more received echo pulses. These are the level surface and any false echoes or multiple echoes. The transmitted pulses and therefore the received pulses have a sine wave form depending upon the pulse duration. A 5.8 GHz pulse of 0.8 nanosecond duration is shown in Fig 4.8. The period of the pulse repetition is shown as T1 in Fig 4.7. Period T1 is Seite 56 the same for the emission pulse repetition as for any echo pulse repetition as shown. However, the sampling signal repeats at period of T2 which is slightly longer in duration than T1. This is the same time expansion procedure by sequential sampling that has already been described for a sine wave. The factor of the time expansion is determined by T1 / (T2-T1). Fig 4.8 Emission pulse (packet). The wave form of the 5.8 GHz pulse with a pulse duration of 0.8 nanoseconds Example The 5.8 GHz, VEGAPULS radar level transmitter has the following pulse repetition rates. Transmit pulse 3.58 MHz Reference pulse 3.58 MHz - 43.7 Hz Therefore the time expansion factor is 81920 giving a time expanded pulse repetition period of 22.88 milliseconds. There is a practical problem in sampling the emission / echo pulse signals of a short (0.8 nanosecond) pulse at 5.8 GHz. An electronic switch would need to open and close within a few picoseconds if a sufficiently short value of the 5.8 GHz sine wave is to be sampled. These would have to be very special and expensive components. 56 T1 = 279.32961 nanoseconds T2 = 279.33302 nanoseconds The solution is to combine sequential sampling with a ‘cross correlation’ procedure. Instead of very rapid switch sampling, a sample signal of exactly the same profile is generated but with a slightly longer time period between the pulses. Fig 4.9 compares sequential sampling by rapid switching with sequential sampling by cross correlation with a sample pulse.
  • 62. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 57 4. Radar level measurement Emission / Echo pulse Sample signal Sampling with picosecond switching Sampling by cross correlation with a sample pulse Fig 4.9 Comparison of switch sampling with ‘cross correlation’ sampling. The pulse radar uses cross correlation with a sample pulse. This means that rapid ‘picosecond’ switching is not required Instead of taking a short voltage sample, cross correlation involves multiplying a point on the emission or echo signal by the corresponding point on the sample pulse. The multiplication leads to a point on the resultant signal. All of these multiplication results, one after the other, lead to the formation of the complete multiplication signal. Fig 4.10 shows a short sequence of multiplications between the received signal (E) and the sampling pulse signal (M). The resultant E x M curves are shown on page 58. Then the E x M curve is integrated and represented on the expanded curve as a dot. The sign and amplitude of the signal on the time expanded curve depends on the sum of the area of the E x M curve above and below the zero line. The final integrated value corresponds directly to the time position of the received pulse E relative to the sample pulse M. The received signal E and sample signal M in Fig 4.10 are equivalent to the periodic signal (sine wave) and sample signal in Fig 4.6. The result of the integration of E x M in Fig 4.10 is directly analogous to the expanded time signal in Fig 4.6. 57
  • 63. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 58 E M ExM max Integral ExM 0 min Fig 4.10 Cross correlation of the received signal E and the sampling M. The product E x M is then integrated to produce the expanded time curve. The technique builds a complete picture of the echo curve The pulse radar sampling procedure is mathematically complicated but a technically simple transformation to achieve. Generating a reference signal with a slightly different periodic time, multiplying it by the echo signal and integration of the resultant product are all operations that can be handled easily within analogue circuits. Simple, but good quality components such as diode mixers for multiplication and capacitors for integration are used. 58 This method transforms the high frequency received signal into an accurate picture with a considerably expanded time axis. The raw value output from the microwave module is an intermediate frequency that is similar to an ultrasonic signal. For example the 5.8 GHz microwave pulse becomes an intermediate frequency of 70 kHz. The pulse repetition frequency (PRF) of 3.58 GHz becomes about 44 Hz.
  • 64. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 59 4. Radar level measurement amplitude Pulse echoes in a process vessel are separated in time t1 t 2 t3 t4 t5 time transmit pulse Fig 4.12 With a pulse radar, all echoes (real and false) are separated in time. This allows multiple echoes caused by reflections from a parabolic tank roof to be easily separated and analysed Pulse radar operates entirely within the time domain and does not need the fast and expensive processors that enable the FM - CW radar to function. There are no Fast Fourier Transform (FFT) algorithms to calculate. All of the pulse radar processing is dedicated to echo analysis only. Part of the pulse radar transmission pulse is used as a reference pulse that provides automatic temperature compensation within the microwave module circuits. The echoes derived from a pulse radar are discrete and separated in time. This means that pulse radar is better equipped to handle multiple echoes and false echoes that are common in process vessels and solids silos. Pulse radar takes literally millions of ‘shots’ every second. The return echoes from the product surface are sampled using the method described above. This technique provides the pulse radar with excellent averaging which is particularly important in difficult applications where small amounts of energy are being received from low dielectric and agitated product surfaces. The averaging of the pulse technique reduces the noise curve to allow smaller echoes to be detected. If the pulse radar is manufactured with well designed circuits containing good quality electronic components they can detect echoes over a wide dynamic range of about 80 dB. This can make the difference between reliable and unreliable measurement. 59
  • 65. 15.01.2007 18:46 Seite 60 Fig 4.11 Block diagram of PULSE radar microwave module radar_applied_to_level_rb.qxd 60
  • 66. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 61 4. Radar level measurement Pulse block diagram - (Fig 4.11) The raw pulse output signal (intermediate frequency) from the pulse radar microwave module is similar, in frequency and repetition rate, to an ultrasonic signal. This pulse radar signal is derived in hardware. Unlike FM - CW radar, PULSE does not use FFT analysis. Therefore, pulse radar does not need expensive and power consuming processors. The pulse radar microwave module generates two sets of identical pulses with very slightly different periodic times. A fixed oscillator and pulse former generates pulses with a frequency of 3.58 MHz. A second variable oscillator and pulse former is tuned to a frequency of 3.58 MHz minus 43.7 Hz and hence a slightly longer periodic time. GaAs FET oscillators are used to produce the microwave carrier frequency of the two sets of pulses. The first set of pulses are directed to the antenna and the product being measured. The second set of pulses are the sample pulses as discussed in the preceding text. The echoes that return to the antenna are amplified and mixed with the sample pulses to produce the raw, time expanded, intermediate frequency. Part of the measurement pulse signal is used as a reference pulse that provides automatic temperature compensation of the microwave module electronics. Pic 3 Two wire pulse radar level transmitter mounted in a process reactor vessel 61
  • 67. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 62 Choice of frequency Process radar level transmitters operate at microwave frequencies between 5.8 GHz and about 26 GHz. Manufacturers have chosen frequencies for different reasons ranging from licensing considerations, availability of microwave components and perceived technical advantages. There are arguments extolling the virtues of high frequency radar, low frequency radar and every frequency radar in between. In reality, no single frequency is ideally suited for every radar level measurement application. If we compare 5.8 GHz radar with 26 GHz radar, we can see the relevant benefits of high frequency and low frequency radar. 2.6 GHz 5.8 GHz Fig 4.14 Comparison of 5.8 GHz and 26 GHz radar antenna sizes. These instruments have almost identical beam angles. However this is not the full picture when it comes to choosing radar frequencies 62
  • 68. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 63 4. Radar level measurement Antenna size - beam angle The higher the frequency of a radar level transmitter, the more focused the beam angle for the equivalent size antenna. With horn antennas, this allows smaller nozzles to be used with a more focused beam angle. For example, a 1½" (40 mm) horn antenna radar at 26 GHz has approximately the same beam angle as a 6" (150 mm) horn antenna at 5.8 GHz. However, this is not the complete picture. Antenna gain is dependent on the square of the diameter of the antenna as well as being inversely proportional to the square of the wavelength. Antenna gain is proportional to:2 diameter wavelength 2 Antenna gain also depends on the aperture efficiency of the antenna. Therefore the beam angle of a small antenna at a high frequency is not necessarily as efficient as the equivalent beam angle of a larger, lower frequency radar. A 4" horn antenna radar at 6 GHz gives excellent beam focusing. A full explanation of antenna gain and beam angles at different frequencies is given in Chapter 5 on radar antennas. Focusing at different frequencies 5 GHz 10 GHz 15 GHz 20 GHz 25 GHz Fig 4.13 For a given size of antenna, a higher frequency gives a more focused beam 63
  • 69. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 64 Antenna focusing and false echoes A 26 GHz beam angle is more focused but, in some ways, it has to be. The wavelength of a 26 GHz radar is only 1.15 centimetres compared with a wavelength of 5.2 centimetres for a 5.8 GHz radar. The short wavelength of the 26 GHz radar means that it will reflect off many small objects that may be effectively ignored by the 5.8 GHz radar. Without the focusing of the beam, the high frequency radar would have to cope with more false echoes than an equivalent lower frequency radar. Fig 4.15 a Low frequency radar has a wider beam angle and therefore, if the installation is not optimum, it will see more false echoes. Low frequencies such as 5.8 GHz or 6.3 GHz tend to be more forgiving when it come to false echoes from the internal structure of a vessel or silo Fig 4.15 b High frequency radar has a much narrower beam angle for a given antenna size. The narrower beam angle is important because the short wavelength of the higher frequencies, such as 26 GHz, reflect more readily from the internal structures such as welds, baffles, and agitators. The sharper focusing avoids this problem 64
  • 70. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 65 4. Radar level measurement Agitated liquids and solid measurement High frequency radar transmitters are susceptible to signal scatter from agitated surfaces. This is due to the signal wavelength in comparison to the size of the surface disturbance. The high frequency radar will receive considerably less signal than an equivalent 5.8 GHz radar when the liq- uid surface is agitated. The lower frequency transmitters are less affected by agitated surfaces. It is important that, whatever the frequency, the radar electronics and echo processing software can cope with very small amplitude echo signals. As discussed, pulse radar has an advantage in this area no matter what the frequency. Fig 4.16 High frequency radar transmitters are susceptible to signal scatter from agitated surfaces. This is due to the signal wavelength in comparison to the size of the surface disturbance. It is important that radar electronics and echo processing software can cope with very small amplitude echo signals. By comparison, 5.8 GHz radar is not as adversely affected by agitated liquid surfaces. Lower frequency radar is generally better suited to solid level applications Condensation and build up Steam and dust High frequency radar level transmitters are more susceptible to condensation and product build up on the antenna. There is more signal attenuation at the higher frequencies, such as 26 GHz. Also, the same level of coating or condensation on a smaller antenna naturally has a greater effect on the performance. A 6" horn antenna with 5.8 GHz frequency is virtually unaffected by condensation. Also, it is more forgiving of product build up. Lower frequencies such as 5.8 GHz and 6.3 GHz are not adversely affected by high levels of dust or steam. These frequencies have been very successful in applications ranging from cement, flyash and blast furnace levels to steam boiler level measurement. In steamy and dusty environments, higher frequency radar will suffer from increased signal attenuation. 65
  • 71. radar_applied_to_level_rb.qxd 15.01.2007 Foam The effect of foam on radar signals is a grey area. It depends a great deal on the type of foam including the foam density, dielectric constant and conductivity. However, low frequencies such as 5.8 GHz and 6.3 GHz cope with low density foam better than higher frequencies such as 26 GHz. For example, a 26 GHz radar signal will be totally attenuated by a very thin detergent foam on a water surface. A 5.8 GHz radar signal will see through this type of foam and continue to see the liquid surface as the foam thickness increases to 150 mm or even 250 mm. 18:46 Seite 66 However, the thickness of foam will cause a small measurement error because the microwaves slow down slightly as they pass through the foam. When foam is present, it is important to provide the radar manufacturer with as much information as possible on the application. Minimum distance Higher frequency radar sensors have a reduced minimum distance when compared with the lower frequencies. This can be an additional benefit when measuring in small vessels and stilling tubes. Summary of the effects of radar frequency Better focusing at higher emitting frequency means: higher antenna gain (directivity) less false echoes reduced antenna size focusing . . . 5 GHz 10 GHz 15 GHz frequency range Fig 4.17 Focusing and radar frequency 66 20 GHz 25 GHz
  • 72. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 67 4. Radar level measurement reduced signal caused by damping Reduced signal strength caused by damping at higher emitting frequency caused by: . . . 5 GHz 10 GHz condensation build - up steam and dust 15 GHz 20 GHz 25 GHz frequency range Fig 4.18 Signal damping and radar frequency Higher damping caused by agitated product surface reflection from medium . . . 5 GHz 10 GHz wave movement material cones with solids signal scattered 15 GHz 20 GHz 25 GHz frequency range Fig 4.19 Signal strength from agitated and undulating surfaces and radar frequency 67
  • 73. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 68 Accuracy Pulse radar bandwidth There is no inherent difference in accuracy between the FM - CW and PULSE radar level measurement techniques. In this book, we are concerned specifically with process level measurement where ‘process accurate’ and cost effective solutions are required. The achievable accuracy of a process radar depends heavily on the type of application, the antenna design, the quality of the electronics and echo processing software employed. The niche market for custody transfer level measurement applications is outside the scope of this book. These custody transfer radar ‘systems’ are used in bulk petrochemical storage tanks. Large parabolic or planar array antennas are used to create a finely focused signal. A lot of processing power and on site calibration time is used to achieve the high accuracy. Temperature and pressure compensation are also used. The carrier frequency of a pulse radar varies from 5.8 GHz to about 26 GHz. The pulse duration is important when it comes to resolving two adjacent echoes. For example, a one nanosecond pulse has a length of about 300 mm. Therefore, it would be difficult for the radar to distinguish between two echoes that are less than 300 mm apart. Clearly a shorter pulse duration provides better range resolution. An effect of a shorter pulse duration is a wider bandwidth or spectrum of frequencies. For example, if the carrier frequency of a pulse is 5.8 GHz and the duration is only 1 nanosecond, then there is a spectrum of frequencies above and below the nominal carrier frequency. The amplitude of the pulse spectrum of frequencies changes according to a Range resolution and bandwidth In process level applications, both FM - CW and PULSE radar work with an ‘envelope curve’. The length of this envelope curve depends on the bandwidth of the radar transmitter. A wider bandwidth leads to a shorter envelope curve and therefore improved range resolution. Range resolution is one of a number of factors that influence the accuracy of process radar level transmitters. 68 sin x x curve. The shape of this curve is shown in Fig 4.21. The null to null bandwidth BWnn of a pulse radar is equal to 2 τ where τ is the pulse duration. It is clear from the curve that the amplitude of frequencies reduces significantly away from the main pulse frequency.
  • 74. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 69 4. Radar level measurement pulse frequency 5.8 GHz 4.8 GHz shorter pulse better range resolution Fig 4.20 Pulse radar range resolution. The guaranteed range resolution is the length of the pulse. A shorter pulse has a wider bandwidth and better range resolution 6.8 GHz bandwidth BW nn, 2 equal to τ Fig 4.21 The null to null bandwidth BWnn of a radar pulse is equal to 2 / τ where τ is the pulse duration. Example a 5.8 GHz radar with a pulse duration of one nanosecond has a null to null bandwidth of 2 GHz Pulse radar envelope curve Fig 4.22 shows how a pulse radar echo curve is used in process level measurement. A higher frequency pulse with a shorter pulse duration will allow better range resolution and also better accuracy because the leading edge of the envelope curve is steeper. Fig 4.22 Envelope curve with pulse radar High frequency, short duration pulse Lower frequency pulse with longer duration Fig 4.23 A shorter pulse duration gives better range resolution. The combination of shorter pulse duaration and higher frequency allows better accuracy because the leading edge of the envelope curve is steeper 69
  • 75. radar_applied_to_level_rb.qxd 15.01.2007 18:46 FM-CW radar bandwidth The bandwidth of an FM - CW radar is the difference between the start and finish frequency of the linear frequency modulation sweep. Unlike pulse radar, the amplitude of the FM - CW signal is constant across the range of frequencies. Seite 70 A wider bandwidth produces narrower difference frequency ranges for each echo on the frequency spectrum. This leads to better range resolution in the same way as with shorter duration pulses with pulse radar. This is explained in the following diagrams and equations. frequency fd = ∆F x 2R Ts x c fd ∆F Ts R fd c ∆F [Eq. 4.1] bandwidth sweep time distance difference frequency speed of light time Ts Fast Fourier Transform The FAST FOURIER TRANSFORM produces a frequency spectrum of all echoes such as that at fd. There is an ambiguity ∆fd for each echo fd. amplitude fd ∆fd = 2 Ts [Eq. 4.2] ∆fd 70 frequency
  • 76. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 71 4. Radar level measurement The ambiguity of the distance R, is ∆ R ∆fd fd ∆R R = 2 Ts ∆F x 2 R Ts x c ∆R R = c ∆F x R ∆R R ∆R = = c ∆F ∆R R amplitude distance Fig 4.24 to 4.26 - FM - CW range resolution [Eq. 4.3] From equation 4.3, it can be seen that with an FM - CW radar the range resolution ∆R is equal to:- c ∆F Therefore, the wider the bandwidth, the better the range resolution. Examples: A linear sweep of 2 GHz has a range resolution of 150 mm whereas a 1 GHz bandwidth has a range resolution of 300 mm. In process radar applications, each echo on the frequency spectrum is processed with an envelope curve. The above equations (Equations 4.1 to 4.3) show that the Fast Fourier Transforms (FFTs) in process radar applications do not produce a single discrete difference frequency for each echo in the vessel. Instead they produce a difference frequency range ∆fd for each echo within an envelope curve. This translates into range ambiguity. 71
  • 77. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 72 FM - CW frequency spectrum - bandwidth and range resolution Frequency spectrum - narrow bandwidth of linear sweep amplitude envelope curves around echoes frequency Frequency spectrum - wide bandwidth of linear sweep amplitude envelope curves around echoes frequency Fig 4.28 Illustration of envelope curve around the frequency spectram of FM - CW radars. The same four echoes are shown for radar transmitters with different bandwidths. An improvement in the range resolution is achieved with a wider bandwidth of the linear sweep Other influences on accuracy As we have demonstrated, FM - CW and PULSE process radar transmitters use an envelope curve for measurement. A wider bandwidth produces better range resolution. The correspondingly short echo will have a steep slope and therefore a more accurate measurement can be made. Other influences on accuracy include signal to noise ratio and interference. 72 A high signal to noise ratio allows more accurate measurement while interference effects can cause a disturbance of the real echo curve leading to inaccuracies in the measurement. Choice of antenna and mechanical installation are important factors in ensuring that the optimum accuracy is achieved.
  • 78. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 73 4. Radar level measurement High accuracy radar FM - CW radar High accuracy of the order of + 1 mm is generally meaningless in an active process vessel or a solids silo. For example, a typical chemical reactor will have agitators, baffles and other internal structures plus constantly changing product characteristics. Although custody transfer level measurement applications are not in the scope of this book, this section discusses how a higher accuracy can be achieved. The fundamental requirement for an accurate FM - CW radar is an accurate linear sweep of the frequency modulation. As with the pulse radar, it is possible to look inside the envelope curve of the frequency spectrum if the application has a simple single echo that is characteristic of a liquids storage tank. This is achieved by measuring the phase angle of the difference frequency. However, this is only practical with custody transfer applications where fast and expensive processors are used with temperature and pressure compensation. Pulse radar For most process applications, measurement relative to the pulse envelope curve is sufficient. However, if the liquid level surface is flat calm and the echo has a reasonable amplitude, it is possible to look inside the envelope curve wave packet at the phase of an individual wave. However, the envelope curve of a high frequency radar with a short pulse duration is sufficiently steep to produce a very accurate and cost effective level transmitter for storage vessel applications. frequency error f2 f2 t1 Fig 4.30 It is essential that the linear sweep of the FM - CW radar is accurately controlled Fig 4.29 Higher accuracy of pulse radar level transmitters can be achieved by looking at the phase of an individual wave within the envelope curve. This is only practical in slow moving storage tanks 73
  • 79. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 74 Power Two wire, loop powered radar Microwave power Radar is a subtle form of level measurement. The peak microwave power of most process radar level transmitters is less than 1 milliWatt. This level of power is sufficient for tanks and silos of 40 metres or more. The average power depends on the sweep time and sweep repetition rate of FM - CW radar and on the pulse duration and pulse repetition frequency of pulse radar transmitters. An increase in the microwave power will produce higher amplitude echoes. However, it will produce higher amplitude false echoes and ringing noise as well as a higher amplitude echoes off the product surface. The average microwave power of a Pulse radar can be as little as 1 microWatt. Pulse radar The low energy requirements of pulse radar enabled the first ever two wire, loop powered, intrinsically safe radar level transmitter to be introduced to the process industry in mid-1997. The VEGAPULS 50 series of pulse radar transmitters have proved to be very capable in difficult process conditions. The performance of the two wire, 4 to 20 mA, sensors is equal to the four wire units that preceded them. The pulse microwave module only needs a 3.3 volt power supply with a maximum power consumption of 50 milliWatts. This drops down to 5 milliWatts when it is in stand-by mode. The difference between the two wire pulse and the four wire pulse is that the two wire radar sends out bursts of pulses and updates the output about once every second. The four wire sends out pulses continuously and updates seven times a second. With high quality electronics, the complete 24 VDC, 4 to 20 mA transmitter is capable of operating at only 14 VDC. This allows it to directly replace existing two wire sensors. Processing power FM - CW radars need a high level of processing power in order to function. This processing power is used to calculate the FFT algorithms that produce the frequency spectrum of echoes. The requirement for processing power has restricted the ability of FM - CW radar manufacturers to make a reliable two wire, intrinsically safe radar transmitter. Pulse radar transmitters work in the time domain without FFT analysis and therefore they do not need powerful processors for this function. Safety The low power output from microwave radar transmitters means that they are an extremely safe method of level measurement. 74 Pulsed FM - CW The low power requirements of pulse radar have allowed two wire radar to become sucessful. FM - CW radar requires processing power and time for the FFT's to be calculated. Power saving has been used to produce a ‘pulsed’ FM - CW radar. However, this device is limited to simple storage applications because the update time is too long and the processing too limited for arduous process applications.
  • 80. radar_applied_to_level_rb.qxd 15.01.2007 18:46 Seite 75 4. Radar level measurement Summary of radar level techniques FM - CW (frequency modulated continuous wave) radar · Indirect method of level measurement · Requires Fast Fourier Transform (FFT) analysis to convert signals into a frequency spectrum · FFT analysis requires processing power and therefore practical FM - CW process radars have to be four wire and not two wire loop powered · FM - CW radars are challenged by large numbers of multiple echoes (caused by the parabolic effects of horizontal cylindrical or dished topped vessels) PULSE radar · Direct, time of flight level measurement · Uses a special sampling technique to produce a time expanded intermediate frequency signal · The intermediate frequency is produced in hardware and does not require FFT analysis · Low processing power requirement mean that practical and very capable two wire, loop powered, intrinsically safe pulse radar can be used in some of the most challenging process level applications 75
  • 81. Contents Foreword Acknowledgement Introduction ix xi xiii Part I 1. History of radar 2. Physics of radar 3. Types of radar 1. CW-radar 2. FM - CW 3. Pulse radar 1 13 33 33 36 39 Part II 4. Radar level measurement 1. FM - CW 2. PULSE radar 3. Choice of frequency 4. Accuracy 5. Power 47 48 54 62 68 74 5. Radar antennas 1. Horn antennas 2. Dielectric rod antennas 3. Measuring tube antennas 4. Parabolic dish antennas 5. Planar array antennas Antenna energy patterns 77 81 92 101 106 108 110 6. Installation A. Mechanical installation 1. Horn antenna (liquids) 2. Rod antenna (liquids) 3. General consideration (liquids) 4. Stand pipes & measuring tubes 5. Platic tank tops and windows 6. Horn antenna (solids) B. Radar level installation cont. 1. safe area applications 2. Hazardous area applications 115 115 115 117 120 127 134 139 141 141 144
  • 82. 5. Radar antennas The function of an antenna in a radar level transmitter is to direct the maximum amount of microwave energy towards the level being measured and to capture the maximum amount of energy from the return echoes for analysis within the electronics. Antennas for level measurement come in five basic forms: · Horn (cone) antenna · Dielectric rod antenna · Measuring tube antenna (stand pipes/ bypass tubes etc.) · Parabolic reflector antenna · Planar array antenna Horn antennas and dielectric rod antennas are already commonly used within process level measurement. We will be discussing how these designs have been developed for increasingly arduous process conditions and how antenna efficiencies have been improved. The horn antenna and versions of the dielectric rod antenna are also used in measuring tube applications within the process industry. Parabolic antennas and planar array antennas have been applied to fiscal measurement radar systems rather than for level measurement within process vessels. We will discuss the design of these antennas although at present their use in process vessels is limited. Antenna basics An important aspect of an antenna is directivity. Directivity is the ability of the antenna to direct the maximum amount of radiated microwave energy towards the liquid or solid we wish to measure. No matter how well the antenna is designed, there will be some microwave energy being radiated in every direction. The goal is to maximise the directivity. Fig 5.1 shows the pattern of radiated energy from a typical horn antenna. This is a 250 mm (10") horn antenna operating at a frequency of 5.8 GHz. The measurements are made some distance from the antenna in what is called the far field zone. It is clear that most of the energy is contained within the main lobe, but also there is a reasonable amount of energy contained within the various side lobes. Technical information and sales literature on radar level transmitters quote beam angles for different antennas. Clearly there is not a tight beam. The convention is to measure the angle at which the microwave energy has reduced to 50 percent of the value at the central axis of the beam. This is quoted in decibels:the - 3dB point. 77
  • 83. Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 20,4 dB 120 60 30 150 180 0 10 20 150 30 0 30 120 main lobe direction : 0,0 deg. angular width (3dB) : 14,9 deg. side lobe suppression : 21,6 dB 60 90 Extent of measured microwave energy showing main lobe and side lobes The - 3 dB point is the beam angle i.e. the energy has reduced to 50% Side lobe energy Fig 5.1 Typical radiation pattern from a radar level transmitter Radiation patterns of different antennas and radar frequencies are compared at the end of this chapter. 78
  • 84. 5. Radar antennas A measure of how well the antenna is directing the microwave energy is called the ‘antenna gain’. Antenna gain is a ratio between the power per unit of solid angle radiated by the antenna in a specific direction to the power per unit of solid angle if the total power was radiated isotropically, that is to say, equally in all directions. isotropic power directional power Isotropic equivalent with total power radiating equally in all directions Directional power from antenna Fig 5.2 Illustration of antenna gain Antenna gain ‘G’ can be calculated as follows: G = ηx ( πxD λ 2 ) = ηx 4π x A λ2 [Eq. 5.1] Where The aperture efficiencies of radar level antennas are typically between η = 0.6 and η = 0.8. D = antenna diameter.* It is clear from equation 5.1 that the directivity improves in proportion A = antenna area.* to the antenna area. At a given freλ = microwave wavelength * quency, a larger antenna has a narrower beam angle η = aperture efficiency * must be same units 79
  • 85. Also, we can see that the antenna gain and hence directivity is inversely proportional to the square of the wavelength. For a given size of antenna the beam angle will become narrower at higher frequencies (shorter wavelengths). For example the beam angle of a 5.8 GHz radar with a 200 mm (8") horn antenna is almost equivalent to a 26 GHz radar with a 50 mm (2") horn antenna. This Beam angle φ = means that a 26 GHz antenna is lighter and easier to install for the same beam angle. However, as discussed in Chapter 4, this is not the whole story when choosing the right transmitter for an application. For a standard horn antenna the beam angle φ, that is the angle to the minus 3 dB position, can be calculated using equation 5.2. 70° x λ D [Eq. 5.2] The following graph shows horn antenna diameter versus beam angle for the most common radar frequencies, 5.8 GHz, 10 GHz and 26 GHz. Antenna beam angles (diameter / frequency) beam angle in degrees (-3dB) 80 5.8 GHz 10 GHz 60 26 GHz 40 20 0 50 75 100 125 150 175 200 225 250 antenna diameter, mm Fig 5.3 Graph showing relation between horn antenna diameter and beam angle for 5.8 GHz, 10Ghz and 26GHz radar 80
  • 86. 5. Radar antennas 1. Horn antennas The metallic horn antenna or cone antenna is well proven for process level applications. The horn is mechanically robust and in general it is virtually unaffected by condensation and product build up, especially at the lower radar frequencies such as 5.8 GHz. There are variations in the internal design of horn antennas. The microwaves that are generated within the microwave module are transmitted down a high frequency cable for encoupling into a waveguide. The metal waveguide then directs the microwaves towards the horn of the antenna. A low dielectric material such as PTFE, ceramic or glass is often used within the waveguide. At the transition from the waveguide to the horn of the antenna the low dielectric material is machined to a pointed cone. The angle of this cone depends on the dielectric constant of the material. For example, ceramic has a sharper angle than PTFE. The microwaves are emitted from this pointed cone in a controlled way and are then focused towards the target by the metal horn. After reflection from the product surface, the returning echoes are collected within the horn antenna for processing within the electronics. Fig 5.4 The transition of microwaves from the low dielectric waveguide into the metallic horn where they are focused towards the product being measured 81
  • 87. Horn antenna design 1 Fig 5.5 1. HF Cable 1 2 3 4 5 6 7 8 2. Signal coupling 3. Waveguide (air filled) Transition rectangular to circular cross section 4. PTFE transition 5. Glass waveguide 9 6. Metallic grid 7. Seal between glass and PTFE 8. PTFE cone 9. Metal horn antenna In this first design of horn antenna the HF cable signal coupling is into an air filled waveguide with a rectangular cross section. The microwaves are directed towards the antenna. There is a transition from rectangular to circular cross section. At this point the waveguide changes to PTFE with a ¼ wavelength step design. The waveguide is then glass filled until it reaches the inside of the antenna horn where it changes to a PTFE cone for the impedance matching into the vapour space in the horn This PFTE cone in combination with the metallic horn focuses the microwaves towards the target. 82 An antenna of this design is capable of withstanding process temperatures up to 250° C and up to 300 Bar. A potential problem with the design is the sealing between the PTFE and glass on the process side. The thermal expansion of glass and PTFE are different and it is possible for condensation to get between the glass and PTFE and to affect the transmission and receipt of the microwave signals. The explosion proof design requires metallic grid around the glass of the waveguide at the joint between the housing casting and the flange casting.
  • 88. 5. Radar antennas Horn antenna design 2 Fig 5.6 1. HF cable 2. Signal coupling 1 3. Waveguide (PTFE filled) 2 3 4 5 4. Process seals Viton or Kalrez 5. PTFE cone 6. Metallic horn antenna 6 With this antenna design, the HF cable is encoupled into the PTFE material inside the waveguide. The metal waveguide is welded to the flange and there are two process seals between the metal waveguide and the PTFE. These seals protect the signal coupler from the process. This seal material can be Viton for stainless steel horn antennas or Kalrez for Hastelloy C horn antennas. There is a continuous transition for the microwaves within a single piece of PTFE which is machined into a cone form for the transition into the horn antenna. The PTFE cone and the metallic conical horn focus the microwaves and collect the return signals in the usual manner. An antenna of this design is capable of withstanding a process temperature of 200° C + and a process pressure of 40 Bar. This antenna design can also be used on very high temperature, ambient pressure applications with air or nitrogen gas cooling of the antenna. 83
  • 89. Horn antenna design 2a Fig 5.7 Very high temperature, ambient pressure applications. Air/nitrogen cooling through flange 1. HF cable 1 2. Signal coupling 2 3. Waveguide (PTFE filled) 3 4 4. Tappings for air/nitrogen keeps antenna area cool Air / N2 5. Metallic horn antenna 5 This adaptation of the previous antenna allows the antenna to be cooled with air or nitrogen gas. This is achieved by drilling two holes, 180° apart, laterally from the flange edge into the horn antenna next to the PTFE cone. The flow of air or nitrogen prevents hot gases from affecting the PTFE and the viton seal and it effectively cools the entire flange and horn area. This technique has been used successfully with very high temperatures, including 1500° C + in the steel industry with applications such as blast 84 furnace burden level and molten iron ladle levels. The microwaves are unaffected by the air movement within the horn area. In addition to cooling, this air purging technique is also used for solids applications where very high levels of conductive dust, such as carbon, heavily coat the inside of the horn and cause signal attenuation. Water purging has also been used where heavy product build up is expected.
  • 90. 5. Radar antennas Horn antenna design 3 Fig 5.8 Special enamel coated antenna 1 2 1. Signal coupling 2. PTFE waveguide 3 4 5 3. PTFE flange face 4. PTFE seal 6 5. Lapped flange 7 6. Steel internals of horn antenna 7. Enamelled coating This antenna is also a development of the antenna design in Fig 5.6. The waveguide, PTFE transition cone and process flange are standard. The face of the flange is all PTFE. The difference is in the application of a special enamel (glass) coated horn that provides excellent process materials compatibility without resorting to more expensive metals such as Tantalum. The external dimensions of the antenna represent a simple cylinder. The internal dimensions of the antenna are identical to a standard horn antenna (150 mm (6")) is illustrated. At the bottom of the antenna there is a gradual lip between the external cylinder and the internal horn. The top of the cylinder has a flange for sealing between the PTFE transition cone and the process flange and also between the glassed antenna and the vessel nozzle. External studs hold the enamel antenna to the process flange and PTFE seals are used to provide internal sealing. The antenna is manufactured from carbon steel with blue enamel coating which is identical to the enamel found in glass lined vessels. It provides the efficiency benefits of a horn antenna with first class materials compatibility. 85
  • 91. Horn antenna design 4 Fig 5.9 High temperature / high pressure antenna with ceramic waveguide 1 2 3 4 1. Connection to HF cable from microwave module 2. Coaxial tube to signal coupling 3. Signal coupling in ceramic waveguide 4.Vacon/ceramic brazing seal 5 5. Graphite seal 6 6. Ceramic waveguide cone The above antenna has been designed with both high temperature and high pressure in mind. The mechanical strength and sealing ability of PTFE degrades at elevated temperature and is therefore limited to about 200° C. This special design of radar has a chemically and thermally stable ceramic (Al2O3) waveguide within a stainless steel or Hastelloy C horn antenna and flange. The ceramic waveguide is fused to a ‘vacon’ steel bush using a special brazing technique. ‘Vacon’ is used because it has a coefficient of thermal expansion that is similar to ceramic, whereas normal 86 stainless steel expands more than twice as much as ceramic. A double graphite seal is fitted on the process side of the ‘vacon’ bush. The entire waveguide assembly is laser welded to ensure that the transmitter is gas tight and that differential thermal expansion is negligible. In order to withstand constant process temperatures of 400° C, the electronics housing of the radar is mechanically isolated from the high process temperature by a temperature extension tube. The microwave module is connected via the HF cable and an air coaxial tube to the signal coupler in the ceramic waveguide.
  • 92. 5. Radar antennas Fig 5.10 Close up of ceramic waveguide assembly 1 2 3 1. HF cable (coaxial) 2. Signal coupling 4 5 3. Ceramic waveguide 6 4. Brazing of ceramic to vacon 5. Vacon bush 6. Graphite seal 7. Metallic horn antenna 7 Fig 5.11 This antenna design is capable of with standing 160 Bar at 400° C with dual graphite seals. Graphite seals have proved to be superior to tantalum seals Ceramic signal coupling Vacon/ceramic brazing Graphite / Tantalum seal 87
  • 93. Adapting horn antenna radars a. Measurement through a PTFE window Another possible variation of a horn antenna radar is measurement through a low dielectric window. We have discussed Hastelloy, Tantalum and the special enamel coated horn antenna. However, if a liquid is being measured and it is conductive or has a dielectric constant of more that εr = 10, then it is possible to measure through a low dielectric window or lens. Some antennas are manufactured with a PTFE window as part of the construction. Antenna housing Horn antenna Process flange PTFE window Fig 5.12 Horn antenna radar is constructed with a metal housing around the antenna and a PTFE process ‘window’ Fig 5.13 Variations of this design include the use of cone shaped windows. The cone can point towards the horn or towards the process 88
  • 94. 5. Radar antennas b. Horn antenna waveguide extension In the first section of Chapter 6, Radar level installations, we discuss how horn antenna radars should be installed. It is recommended that the end of the antenna is a minimum of 10 mm inside the vessel. A 150 mm (6") horn antenna is 205 mm (8") long. If the nozzle is longer than 200 mm, we should consider a waveguide extension piece between the radar flange and the horn antenna. Waveguide extensions should only be used with highly reflective products. c. Horn antenna bent waveguide extensions As well as simple waveguide extensions it is possible to bend waveguide extensions in order to avoid obstructions or to utilise side entry flanges. A simple 90° bend or an ‘S’ shaped extension tube are possible. The waveguide extensions should be free from any internal welds and the minimum radius of curvature should be 200 mm. Fig 5.14 Extended waveguide horn antenna to enable measurement in long nozzles or through a concrete tank or sump roof Waveguide extension with ‘S’ bend Fig 5.15 Waveguide extensions with bends. The direction of the polarization is important Waveguide extension with 90° bend 89
  • 95. High frequency radar antennas The majority of antennas in this chapter are designed for microwave frequencies of between 5.8 GHz and 10 GHz. Later in this chapter, we discuss the use of radar in measuring tubes where there is a minimum critical diameter for each frequency. A measuring tube is a waveguide. The minimum theoretical tube diameter for a 5.8 GHz radar is 31 mm. At a higher frequency the minimum diameter of a waveguide is smaller. At this minimum diameter, the microwaves are established within the waveguide with a single mode and hence a single velocity. As the waveguide diameter increases in size, more modes become established for the given frequency. Measurement problems will be encountered if there are multiple modes within an antenna waveguide. This is because with different modes the microwaves travel at different velocities in the waveguide and therefore a single target will reflect more than one return echo. Measurement will become inaccurate or impossible. For this reason, the encoupling of a high frequency radar must be made into a small waveguide. The small waveguide assemblies of high frequency radar are susceptible to contamination by condensation and build up when compared with lower frequencies such as 5.8 GHz. 90 A special patented high frequency antenna design from VEGA minimises the potential problems associated with small waveguide assemblies. The encoupling is made within a small PTFE waveguide to establish a single mode. As the microwaves travel towards the horn antenna, there is a carefully designed transition that increases the diameter of the PTFE waveguide while maintaining the single mode. The increased diameter of the PTFE waveguide reduces the adverse effects of condensation and build up where the tapered cone of the waveguide enters the metallic horn of the antenna. Compare this design with horn antenna design 2, Fig 5.6. The 5.8 GHz radar does not need a transition in the waveguide diameter and the angle of the metallic horn is not as sharp as for the high frequency radar. Viton or Kalrez process seals are fitted between the PTFE and stainless steel body of the waveguide. Extended versions of the high frequency antenna design involve lengthening the HF cable within a stainless steel extension tube and welding the waveguide assembly to the end of the extension tube.
  • 96. 5. Radar antennas Fig 5.16 High frequency (26GHz) horn antenna design 1. HF cable from microwave module 2. Signal coupling into smaller diameter PTFE waveguide assembly 1 2 3 4 5 6 3. Carefully designed transition from small diameter to larger diameter without affecting the waveguide mode 4. Viton or Kalrez process seals between PTFE and stainless steel of the waveguide 5. Cone shape of PTFE waveguide for the transition into the metallic horn of the antenna 6. Metallic horn antenna of high frequency radar. It has a sharper angle than the lower frequency radars 91
  • 97. 2. Dielectric rod antennas The dielectric rod antenna is an extremely useful option when applying radar level technology to modern process vessels. Dielectric rods can be used in vessel nozzles as small as 40 mm (1½") and they are manufactured from PP, PTFE or ceramic wetted parts. This means that, normally, radar level transmitters can be retro-fitted into existing tank nozzles and they have low cost materials compatibility with most aggressive liquids including acids, alkalis and solvents. The design of dielectric rod antennas has been refined in recent years. Essentially the microwaves are fed from the microwave module through an HF cable to a signal coupler in the waveguide. As with the horn antenna the waveguide can be air filled or filled with a low dielectric material such as PTFE . The waveguide feeds the microwaves to the antenna. The microwaves pass down the parallel section of the rod until they reach the tapered section of the rod. The tapered section of the rod acts like a lens and it focuses the microwaves towards the product being measured. The size and shape of the dielectric rod depends on the frequency of the microwaves being transmitted. 92 The reflected echoes are captured in a similar fashion for processing by the radar electronics. Rod antennas should only be used on liquids and slurries and not on powders and granular products. There are some important considerations when applying rod antenna radars. First of all, the tapered section of the rod must be entirely within the vessel. If the tapered section is in a nozzle, it will cause ‘ringing’ noise that will effectively blind the radar. This is explained more fully in Chapter 6. Also, it can be seen from Fig 5.17 that the microwaves rely on the rod antenna being clean. If a rod antenna is coated in viscous, conductive and adhesive products, the antenna efficiency will deteriorate very quickly. With the horn antenna product build up is not a particular problem. However, product build up works against the reliable functioning of a rod antenna radar.
  • 98. 5. Radar antennas Fig 5.17 Dielectric rod antenna The microwaves travel down the inactive parallel section of the rod towards the tapered section . The tapered section of the rod focuses the microwaves toward the liquid being measured . It is very important that all of the tapered section of the rod must be inside the vessel It is not good practice to allow a rod antenna to be immersed in the product If a rod antenna is coated in viscous, conductive and adhesive product, the antenna efficiency will deteriorate 93
  • 99. Rod antenna design 1 Fig 5.18 Rod antenna for short process nozzles 1 2 3 1. HF cable 2. Process connection PVDF boss 4 3. Signal coupling within PTFE/PP filled waveguide 4. Inactive section with metallic waveguide, PTFE/PP inner and outer parts 5. Solid PTFE/PP active tapered section of antenna focuses the microwaves towards the product surface 5 This rod antenna is a simple and low cost design that provides a radar level transmitter with good materials compatibility. It is ideal for vented and low pressure vessels such as acid and alkali tanks. It is designed for use in short 1½" BSP / NPT process nozzles. The nozzle height should not exceed 60 mm (2½"). The process connection is a 1½" PVDF boss and the antenna is polypropylene (PP) or PTFE. 94 The HF cable from the microwave module is coupled into PTFE/PP inside a metallic tube that acts as a waveguide. This metallic tube is totally enclosed within the PTFE/PP parallel section of the antenna. The microwaves pass down the metallic waveguide directly to the tapered section of the antenna where they are focused towards the product being measured.
  • 100. 5. Radar antennas Rod antenna design 2 Fig 5.19 Rod antenna with solid PTFE extendible rod 1. HF cable 1 2 3 4 5 2. Signal coupling 3. Air waveguide 4. PTFE cone 5. Process connection 6 7 With this design of rod antenna the signal coupling is into an air filled waveguide. The microwaves are directed towards the antenna. There is a transition to PTFE via a cone shaped element. The microwaves continue through the PTFE waveguide to the solid PTFE dielectric rod. The tapered section of the rod focuses the microwaves towards the product being measured. 6. Solid PTFE parallel section length can be extended 7. Solid PTFE tapered section If this type of antenna is to be used in a long nozzle, the parallel section of the solid rod is extended to ensure that the tapered section is entirely within the vessel. An extended, solid PTFE rod antenna can suffer from ‘ringing’ noise caused by microwave leakage from the parallel section resonating within the nozzle. See Fig 5.20. 95
  • 101. Fig 5.20 Extended rod antenna in solid PTFE. This design can suffer from ‘ringing’ noise caused by leakage of microwave energy from the parallel section of the solid PTFE rod resonating in the vessel nozzle In theory, the microwaves should travel within the parallel section for the entire length until it reaches the tapered section. However, in practice, some of the microwave energy escapes from the parallel sides. Some solid PTFE rod antennas are supplied with screw - on extendible antennas. In addition to the ‘ringing’ noise problem described, this design can suffer from condensation forming between the rod sections causing signal attenuation. 96 Also the PTFE expands at elevated temperatures and under certain process conditions it is possible for the rod sections to detach. The potential problems of solid PTFE rod antennas have been solved by the latest designs. It is important to have a completely inactive parallel section within a vessel nozzle. This is achieved by special screening or signal coupling beyond the nozzle.
  • 102. 5. Radar antennas Rod antenna design 3 Fig 5.21 Extended rod antenna with inactive section and signal coupling below nozzle level 1. HF cable 1 2. Rod extension casting (metal within PTFE) 2 3. Signal coupling at the bottom of the rod extension 3 4. Inactive section 4 5. Solid PTFE tapered ‘active’ section of rod antenna 5 This antenna is designed for use in nozzles of either 100 mm length or 250 mm length. All wetted parts of the antenna are PTFE. The parallel section that is designed to be within the nozzle has a PTFE coating on a cast metal tube. Below this parallel section is the active, solid PTFE, tapered antenna. The HF cable from the microwave module is fed through the metal casting and the signal coupling is made just above the tapered rod. The parallel and tapered sections are sealed together and are designed to withstand a process temperature of 150° C . This antenna design is used with 1½" BSP (M) stainless steel bosses or with PTFE faced flanged transmitters. The flanged version is designed for maximum chemical resistance to acids, alkalis and solvents. The flange face is PTFE with a tight seal between the flange PTFE and the top of the PTFE covered inactive section. 97
  • 103. Extended rod antenna for 250 mm nozzle Extended rod antenna for 100 mm nozzle Fig 5.22 Extended rod antenna with inactive section and signal coupling below nozzle level. All wetted parts are PTFE on the flanged version of this antenna For less arduous applications a stainless steel extension tube is used instead of the PTFE covered tube. The tapered section of the antenna is made of polyphenylene sulphide (PPS). Fig 5.23 Extended rod antenna with stainless steel inactive section and PPS rod antenna. This is for less chemically arduous process conditions 98
  • 104. 5. Radar antennas Rod antenna design 4 Fig 5.24 Extended rod antenna with metallic grid waveguide extension within carbon impregnated PTFE inactive rod. Tapered active section of virgin PTFE 1. HF cable 1 2. Signal coupling 2 3 4 5 3. PTFE waveguide 6 7 4. Screwed connection 5. Carbon impregnated PTFE antenna parallel section and flange face 6. Internal metal grid acts as extended waveguide and prevents microwave leakage from the parallel section of the antenna 7. PTFE waveguide 8 This design of dielectric rod antenna is for use with flanged process connections. The HF cable is connected into a PTFE filled waveguide which directs the microwave energy towards the rod antenna. There is a PTFE male screwed fitting at the end of the waveguide within the process flange. The fabricated, one piece, rod antenna screws on to this connection. The antenna flange facing and the parallel section of the antenna have carbon impregnated PTFE wetted parts. Inside the parallel section of the rod there is a tubular metallic grid that acts 8. Virgin PTFE tapered antenna as an extension to the waveguide. Inside the grid the waveguide is virgin PTFE, outside the grid the PTFE is carbon impregnated. At the end of the parallel section, there is a transition into a solid PTFE tapered rod which provides the impedance matching and focusing of the microwaves towards the product being measured. This antenna has the option for 100 mm or 250 mm nozzle lengths. As already discussed, the tapered section must be entirely within the vessel. 99
  • 105. Rod antenna design 5 Fig 5.25 This is a high temperature ceramic rod antenna design. There is temperature separation between the electronics and the signal coupling (similar to the high temperature horn antenna Fig 5.10). The ceramic rod has a sharper taper than the equivalent PTFE rod 1 2 3 4 1. Signal coupling 2. Ceramic waveguide 3. Process seal (graphite or tantalum) 4. Active tapered ceramic rod Rod antennas are available with the dielectric rod manufactured from ceramic (Al2O3). Ceramic has good chemical and thermal resistance. However, care must 100 be taken when installing ceramic rods because they are brittle and prone to accidental damage.
  • 106. 5. Radar antennas 3. Measuring tube antennas As discussed, conical horn antennas and dielectric rod antennas are used widely within the process industry. In general horn antennas are mechanically more robust and do not suffer as much from build up or heavy condensation. On the other hand, dielectric rods are smaller, weigh less and can be constructed from low cost but chemically resistant plastics such as PTFE and polypropylene. However, there are applications within the process industry where the installation of an antenna directly within a vessel is not suitable for reasons of vessel design or radar functionality. In these cases a measuring tube (bypass tube or a stand pipe within the vessel) may be an alternative. Bypass tube and stand pipes are used for the following reasons: · · · Highly agitated liquid surfaces a stilling tube ensures that the radar sees a calm surface with no scattering of the echo signal Low dielectric liquids such as liquefied petroleum gas (LPG) a stand pipe concentrates and guides the microwaves to the product surface giving the maximum signal strength from liquids with low levels of reflected energy Toxic and dangerous chemicals a stand pipe installation makes a small antenna size possible. This can be used to look through a full bore ball valve into the stand pipe. The instrument can be isolated from the process for maintenance · · · Small vessels - stand pipes or bypass tubes can be used for measurement in very small process vessels such as vacuum receivers. There may not be enough head space for a rod antenna or a suitable connection for a horn antenna. A small bore tube can be used with a radar Foam - a stilling tube can often prevent foam affecting the measurement Replacing existing floats and displacers - radar can be installed directly into existing bypass tubes 101
  • 107. Measuring tube radar 1 - horn antennas Fig 5.26 Installation of horn antenna radars into stand pipes or bypass tube DN50 DN80 DN100 ∅ 50 ∅ 80 ∅ 100 Horn antenna radars are most commonly used in measuring tube level applications. Stilling tube internal diameters can be 40 mm (1 ½"), 50 mm (2"), 80 mm (3"), 100 mm (4") and 150 mm (6"). Larger tubes are possible. Normally, the 40 mm and 50 mm tubes do not require a horn. The PTFE or ceramic waveguide impedance matching cone can be installed directly into the tube. 102 DN150 ∅ 150 For 80 mm and above, the appropriate horn antenna is attached and this is designed to fit inside the tube. As discussed in Chapter 2, Physics of radar and Chapter 6, Radar level installations, the linear polarization of the radar must be directed towards the tube breather hole or mixing slots, or towards the process connections in the case of a bypass tube.
  • 108. 5. Radar antennas Measuring tube radar 2 - offset rod antennas Fig 5.27 Offset rod antenna for use on 50 mm and 80 mm measuring tubes 1 1. HF cable 2. Signal coupling 3. PTFE faced flange 4. Offset short solid PTFE rod antenna 2 3 4 The standard length dielectric rod antennas should not be installed within measuring tubes. There is a high level of ‘ringing’ noise which severely reduces the efficiency of the antenna. However, a special design of short, offset rod antenna can be used on small diameter tubes (50 mm and 80 mm). This design is similar in construction to rod antenna design 3. All wetted parts are in PTFE and the short antenna is off centre. This asymmetric design produces improved signal to noise ratios within a measuring tube. 103
  • 109. Microwave velocity within measuring tube The speed of microwaves within a measuring tube is apparently slower when compared to the velocity in free space. The degree to which the running time slows down depends on the diameter of the tube and the wavelength of the signal. cwg = co x { 1- 2 } λ ( 1.71d )2 [Eq. 5.3] The microwaves bounce off the sides of the tube and small currents are induced in the walls of the tube. For a circular tube, or waveguide, the velocity change is calculated by the following equation : cwg co λ d is the speed of microwaves in the measuring tube / waveguide is the speed of light in free space is the wavelength of the microwaves is the diameter of the measuring tube Fig 5.28 The transit time of microwaves is slower within a stilling tube. This effect must be compensated within the software of the radar level transmitter 104
  • 110. 5. Radar antennas There are different modes of propagation of microwaves within a waveguide. However, an important value is the minimum diameter of pipe that will allow microwave propagation. The value of the critical diameter, dc , depends upon the wavelength λ of the microwaves: The higher the frequency of the microwaves, the smaller the minimum diameter of measuring tube that can be used. dc = Equation 5.4 shows the relationship between critical diameter and wavelength. For example, 5.8 GHz has a wavelength λ of ~ 52 mm. The minimum theoretical tube diameter is dc = 31 mm With a frequency of 26 GHz, a wavelength of 11.5 mm, the minimum tube diameter is dc = 6.75 mm. In practice the diameter should be higher. The diameter for 5.8 GHz should be at least 40 mm. λ 1.71 [Eq. 5.4] % speed of light, c 100 80 60 40 20 0 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 Tube diameter / wavelength, d / λ Fig 5.29 Graph showing the effect of measuring tube diameter on the propagation speed of microwaves Higher frequencies such as 26 GHz will be more focused within larger diameter stilling tubes. This will minimise false echoes from the stilling tube wall. The installation requirements of radar level transmitters in measuring tubes are covered in the next chapter. 105
  • 111. 4. Parabolic dish antennas Fig 5.30 Typical parabolic antenna 1. Feed from microwave module 2. Parabolic reflector secondary antenna 1 3. Primary antenna 4. Focus of parabolic reflector 2 3 4 The subject of this book is radar level measurement in process vessels. Although they are usually applied to custody transfer applications and not process vessel applications, the subject of antennas would not be complete without discussion of parabolic antennas. The parabolic antenna is well known to all. The parabolic form is widely seen from satellite television dishes and radio telescopes to car headlights and torch beams. 106 The main structure of a parabolic antenna is the parabolic reflector dish. This is usually of stainless steel construction and is designed to focus the microwaves as accurately as possible. The microwaves are fed through the centre of the dish to the primary antenna that is in front of the dish at the focus. The microwave energy is transmitted from the primary antenna back towards the parabolic dish, the secondary antenna, which reflects the energy and focuses it towards the product being measured.
  • 112. 5. Radar antennas The reflected energy is captured by the dish and focused back to the primary antenna for echo analysis. Parabolic antennas are used widely in custody transfer applications and are well proven in large storage tanks. The benefits of parabolic antennas in these applications are clear. The good focusing of the paraboloid shape ensures high antenna gain or directivity. Also this narrow beam angle results in higher sensitivity. However, parabolic antennas are large, heavy, relatively complex and expensive to manufacture. These factors limit the use of parabolic antennas in most process level applications. The central feed to the primary antenna at the focus of the dish causes a blind area directly in front of the antenna. This can reduce the antenna efficiency. Parabolic antennas have been applied to bitumen storage tanks where build up on the parabolic dish is said to cause minimum signal attenuation. If the primary antenna was coated in viscous product, this would cause a major problem to the signal strength. In conclusion, the parabolic antenna has a niche application in fiscal measurement of large, slow moving product tanks, but is not suitable for the arduous conditions that are prevalent in the wide variety of vessels within the process industries. Pic 1. Parabolic antennas have been around since the beginning of radar 107
  • 113. 5. Planar array antennas Fig 5.31 Planar antenna - side view 1 1. Electronics housing 2. Process flange 2 3. Antenna feed 4. Stainless steel back 5. Microwave absorbing material 3 6. Microwave patches 7. PTFE process seal 4 5 6 7 Planar array antennas were originally designed and built for aerospace radar applications. When the nose cone of a modern jet fighter is removed, it reveals a flat circular disk faced with dielectric material and covered with small slots instead of the more ‘traditional’ parabolic metal dish. This flat disk is typical of the planar array antennas which have been developed for use on radar level transmitters. Planar array antennas have the advantage of being relatively small and light in weight especially when compared with parabolic antennas. 108 The construction of a planar array antenna for a radar level transmitter is quite complex. The antenna is backed with a round stainless steel disk that provides rigidity and strength to the assembly. The steel disk is faced with a microwave absorbing material. This material ensures that the microwave energy is directed towards the process and that there is no ‘ringing’ noise interference from microwave energy bouncing off the steel back plate.
  • 114. 5. Radar antennas Fig 5.32 Cut away of planar array antenna for radar level transmitter 1. Stainless steel back to antenna provides rigidity 1 2 3 4 5 2. Microwave feed through antenna back into feed network to microwave patches 3. Microwave absorbing material prevents ringing from stainless steel back 4. Microwave patches with low dielectric layers between them focus the microwaves from each element of the array 5. PTFE process seal with anti-static elements The microwaves pass in a common feed from the microwave module through the stainless steel and absorption material to a feed network across the area of the planar antenna. A pattern of microwave patches are fed from this network. There is a pattern of microwave elements across the area of the antenna. Each element is built up of three or more microwave patches with dielectric material between. This forms a multiple microwave array with many individual elements transmitting from the face of the planar antenna. Finally, the microwave elements and the bonding materials that form the structure of the planar antenna are protected by a PTFE process seal covering the face of the antenna. Additional antistatic material is used for hazardous area applications. Planar antennas can be designed with good focusing of the microwaves and minimal side lobes. As well as applications within vessels, they can be used for measuring tube applications. 109
  • 115. Antenna energy patterns At the beginning of this chapter we stated that the definition of ‘beam angle’ is the angle at which the microwave energy measured at the centre line of the radar beam has reduced to 50% or minus 3 dB. We discussed directivity and antenna gain and stated that even the best designed antennas have side lobes of energy. The aim is to maximize the directivity and minimise the effect of side lobes. The metallic horn (or cone) antenna and the dielectric rod antenna are the most practical for process level measurement. The following pages show antenna radiation patterns for different antenna types, frequencies and sizes. These can be summarised as follows : · Larger horn antennas have more focused beam angles · Dielectric rod antennas have more side lobes than horn antennas · For more focusedofthe beam angle- the higher the frequency a given size horn antenna the 1. Comparison of horn antenna beam angle with horn diameter The following diagrams show the comparison of 100 mm, 150 mm and 250 mm (4",6" & 10") horn antennas at 5.8 GHz Fig 5.33 Horn antenna 100mm (4"), frequency 5.8GHz, beam angle 32° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 14,3 dB 60 120 150 30 180 -10 0 10 150 30 120 main lobe direction : 0,0 deg. angular width (3dB) : 32,1 deg. side lobe suppression : 16,9 dB 110 20 60 90 0
  • 116. 5. Radar antennas Fig 5.34 Horn antenna 150mm (6"), frequency 5.8GHz, Beam angle 27.9° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 15,4 dB 120 60 150 30 180 -10 0 10 150 20 0 30 120 60 : 0,0 deg. main lobe direction angular width (3dB) : 27,9 deg. side lobe suppression : 20,9 dB 90 Fig 5.35 Horn antenna 250mm (10"), frequency 5.8GHz, Beam angle 14.9° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 20,4 dB 120 60 30 150 180 0 10 20 150 30 0 30 120 main lobe direction : 0,0 deg. angular width (3dB) : 14,9 deg. side lobe suppression : 21,6 dB 60 90 111
  • 117. 2 Comparison of dielectric rod antenna with horn antenna The following show a 5.8 GHz horn antenna compared with a 5.8 GHz rod antenna. Although the beam angles are similar, the rod has more significant side lobes. Fig 5.36 Dielectric rod antenna, 5.8 GHz. Beam angle 32° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 15,2 dB 120 60 150 30 180 -10 0 10 150 20 0 30 120 60 main lobe direction : 0,0 deg. angular width (3dB) : 32,0 deg. side lobe suppression : 14,6 dB 90 Fig 5.37 150mm (6"), horn antenna, 5.8 GHz. Beam angle 27.9° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 15,4 dB 60 120 30 150 180 -10 0 10 150 30 120 main lobe direction : 0,0 deg. angular width (3dB) : 27,9 deg. side lobe suppression : 20,9 dB 112 20 60 90 0
  • 118. 5. Radar antennas 3 Frequency differences and beam angles The following diagrams show the beam angle of 26 GHz radar with a 40 mm (1½" ) and 80 mm (3") horn antenna. These should be compared with the previous 5.8 GHz horn antenna patterns. Fig 5.38 40 mm (1½") horn antenna, 26 GHz. Beam angle 18.2° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 19,3 dB 120 60 150 30 180 -10 0 10 150 20 0 30 120 60 main lobe direction : 0,0 deg. angular width (3dB) : 18,2 deg. side lobe suppression : 17,2 dB 90 Fig 5.39 80 mm (3") horn antenna, 26 GHz. Beam angle 9.4° Farfield E_Abs (Theta); Phi=90,0 deg. 90 Max.: 24,3 dB 120 60 150 30 180 0 10 20 150 30 0 30 120 main lobe direction : 0,0 deg. angular width (3dB) : 9,4 deg. side lobe suppression : 22,1 dB 60 90 113
  • 119. Inhalt Vorwort Danksagung Einleitung ix xi xiii 1. Geschichte des Radars 2. Physikalische Grundlagen des Radars 3. Radartypen 1. CW-Radar 2. FMCW-Radar 3. Pulsradar 1 13 33 33 36 39 4. Radar-Füllstandmessung 1. FMCW-Radar 2. Pulsradar 3. Frequenzwahl 4. Genauigkeit 5. Leistung 47 48 54 62 68 74 Teil I Teil II 5. Radarantennen 1. Hornantennen 2. Dielektrische Stabantennen 3. Standrohrantennen 4. Parabolantennen 5. Planarantennen Richtcharakteristik von Antennen 77 81 92 101 106 108 110 6. Installation A. Mechanischer Einbau 1. Flüssigkeitsanwendungen - Hornantenne 2. Flüssigkeitsanwendungen - Stabantenne 3. Allgemeine Einbauhinweise 4. Standrohre und Bypass-Rohre 5. Messung durch Behälterwand und Radarfenster 6. Messung von Schüttgütern mit Hornantennen B. Elektrische Anschlussvarianten 1. Nicht-Ex-Anwendungen 2. Geräte für Ex-Anendungen 115 115 115 117 120 127 134 139 141 141 144
  • 120. 6. Installation Mechanischer Einbau Der richtige Einbau ist für die Funktion eines Füllstandradars von sehr großer Bedeutung. Obwohl die Signalverarbeitungssoftware moderner Geräte inzwischen auch schlechte Echoverhältnisse zuverlässig auswerten kann, ist dies immer noch die wichtigste Vorausetzung für eine funktionierende Messung. 1. Flüssigkeitsanwendungen - Hornantenne Stutzen / Muffen Üblicherweise werden Radarsensoren auf einem Behälterstutzen oder einer Muffe installiert. Referenzpunkt für die Messung ist die Unterseite des Geräteflansches. Die Vorderkante der Hornantenne sollte immer mindestens 10 mm aus dem Stutzen heraus in den Behälter ragen. Korrekter Einbau Die Hornantenne eines Gerätes mit einem Flansch DN 150 (6") ist z.B. 205 mm lang. Ist der Montagestutzen deutlich länger als 195 mm, sollte eine Hohlleiterverlängerung verwendet werden. So kann garantiert werden, dass das Ende der Hornantenne über den Stutzen hinausragt. Falscher Einbau 10 mm Abb. 6.1 Abb.6.2 115
  • 121. Hohlleiterverlängerung und gebogene Hohlleiter Eine Hohlleiterverlängerung sollte verwendet werden, wenn ein Radargerät mit Hornantenne in einem langen Stutzen installiert wird. Hierfür wird ein Edelstahlrohr zwischen den PTFE / keramischen Hohlleiter im Flansch und der Hornantenne montiert. Es ist auch möglich, die Hohlleiterverlängerung für einen seitlichen Einbau des Gerätes abzubiegen. Der minimale Biegeradius für diesen Antennentyp ist 200 mm, der Winkel sollte nicht über 90° betragen. Bei der Verwendung eines gebogenen Hohlleiters ist die Ausrichtung der linearen Polarisation des Radars wichtig. Die Polarisationsrichtung des Radars sollte horizontal sein, wenn die Biegung nach unten verläuft. Verlängerte und gebogene Hohlleiter sind für Flüssigkeiten mit guten Reflexionseigenschaften geeignet. Sie sollten nicht bei Flüssigkeiten mit niedrigen DK-Werten oder bei Schüttgütern verwendet werden. Abb. 6.3: Einbau von Geräten mit Hornantenne. 116 Minimale Messdistanz bei Geräten mit Hornantenne Mit einer Hornantenne ist es normalerweise möglich, flüssige Medien bis an die Unterkante der Antenne zu messen. Dies ist allerdings nur möglich, wenn die Flüssigkeit gute Reflexionseigenschaften hat. Das Eintauchen der Antennen in die Flüssigkeit, eventuell sogar mit Anhaftungen, verursacht insbesondere bei 6,3 GHz-Geräten kaum Probleme.
  • 122. 6. Installation 2. Flüssigkeitsanwendungen - Stabantenne Stutzen / Muffen Eine PTFE-Stabantenne eignet sich gut bei chemisch aggressiven Produkten wie Säuren und Laugen. Sie wird oft in der chemischen und pharmazeutischen Industrie benutzt, wo Mischungen aus Lösungsmitteln, Säuren und Laugen alltäglich sind. Die PTFE-Stabantennen mit TriClamp und spaltfreier Dichtkonstruktion sind speziell für Anwendungen in der Lebensmittelindustrie und für sterile Behälter optimiert. Die Stabantenne wird für Flüssigkeiten und Schlämme, aber nicht für Schüttgutanwendungen benutzt. Der Sensor ist meistens in einem einfachen Stutzen oder in einer Gewindemuffe eingebaut. Radarsensoren mit Stabantenne werden passend für geschraubte Verbindungen wie 1½" (NPT oder G), Flanschanschlüsse von DN 50 (2") bis DN 150 (6") oder hygienische Lebensmittelanschlüsse geliefert. Beim Einbau ist wichtig, dass der komplette konische Teil der Antenne aus dem Stutzen in den Behälter ragt. Für den Einbau in langen Stutzen sind Stabantennen mit unterschiedlichen inaktiven Längen verfügbar. Typische Längen für diesen inaktiven Teil, und somit die maximale Länge des Stutzens, sind 100 mm und 250 mm. Abb. 6.4: Typische Einbau einer Stabantenne: Der aktive konische Teil der Antenne muss komplett in den Behälter ragen. Für längere Stutzen sollten Antennen mit inaktiver Länge verwendet werden. 117
  • 123. Falscher Einbau einer Stabantenne Wenn der konische Abschnitt einer Stabantenne in einem Stutzen montiert wird, erzeugen die abgestrahlten Mikro- wellen ein starkes Rauschen (Klingeln). Dies führt speziell im Nahbereich zu einer Verringerung der Messsicherheit. Abb. 6.5: Richtig: Antenne mit angepasstem inaktiven Teil für lange Stutzen. Normale Rauschkurve mit deutlichem Echo. Abb. 6.6: Falsch: Kurze Stabantenne in einem langen Stutzen. Produziert hohes „Klingeln“. Im Nahbereich kann dies sogar das Echo überdecken. 118
  • 124. 6. Installation Stabantenne direkt auf dem Behälter Radarsensoren mit Stabantenne können direkt in eine Öffnung in der Decke eines Tanks montiert werden. Dies kann entweder über einen Flansch oder ein Einschraubgewinde geschehen. Maximale Füllhöhe bei einer Stabantenne Wie bereits erklärt, ist es wichtig, dass der konische Abschnitt einer Stabantenne komplett innerhalb des Behälters ist. Die Gerätesoftware kann das „Klingeln“ bei einem falschen Einbau nicht eliminieren. Eine Erhöhung der Verstärkung würde dies noch weiter verschlechtern. Die Länge der Stabantenne ab dem Flansch bestimmt die maximale Befüllhöhe im Behälter. Im Idealfall sollte das flüssige Füllgut die Stabantenne nicht berühren. Allerdings ist dies manchmal unvermeidlich, hierbei muss Folgendes in Betracht gezogen werden. Anhaftungen auf der Stabantenne Wie schon erklärt, werden die Mikrowellen bei einer Stabantenne vom konischen Abschnitt des Stabs ausgesandt. Taucht nun der Stab in eine viskose Flüssigkeit ein, und das Produkt bildet auf der Antenne einen Überzug, so gefährdet dies die Messung. Bilden sich starke Anhaftungen, dann wird das Radar nicht mehr funktionieren. Berühren niedrigviskose Flüssigkeiten wie z.B. Lösungsmittel oder wasserbasierende Produkte die Stabantenne, kann dies sogar einen Selbstreinigungseffekt haben und die Messung bleibt stabil. Bei solchen Medien kann die Antenne bis zur Hälfte eintauchen. Jedoch ist auch hier schon mit deutlich verringerter Messsicherheit und Genauigkeit zu rechnen. Nach Möglichkeit sollte ein Eintauchen der Antenne gänzlich vermieden werden. Mechanische Belastung Es sollte beachtet werden, dass die PTFE-Antennen nur beschränkten mechanischen Belastungen widerstehen können. Beim Auftreten einer Querkraft kann sie sich biegen und verformen oder sogar brechen. Hat die Anwendung starke Füllgutbewegungen? Kann die Biegekraft Schaden am Stab verursachen? 119
  • 125. 3. Allgemeine Einbauhinweise: Horn- und Stabantenne bei Flüssigkeitsanwendungen Folgendes sollte bei der Montage eines Radargerätes mit Horn- oder Stabantenne auf einem Behälter berücksichtigt werden. Montage in Behältern mit gewölbtem Deckel Ein Radarsensor sollte nicht im Zentrum eines gewölbten Deckels oder zu nahe an der Gefäßwand montiert werden. Die ideale Position ist ungefähr ½ Radius von der Außenwand entfernt. Gewölbte Tankdeckel können sonst als parabolischer Reflektor wirken. Ist der Radarsensor im „Brennpunkt“ eines parabolischen Deckels montiert, empfängt er deutlich überhöhte Vielfachechos. Dies wird vermieden, wenn der Sensor wie zuvor beschrieben eingebaut wird. Paraboleffekt Wird ein Radarfüllstandmessgerät im Zentrum eines gewölbten Deckels montiert, empfängt der Sensor stark überhöhte Vielfachechos. Der Effekt dieser Vielfachechos kann deutlich auf der Echokurve betrachtet werden. Abb. 6.8 zeigt, dass das dritte Vielfache eine deutlich höhere Amplitude aufweist als das erste, tatsächliche Echo. Dieser Effekt kann auch in liegenden Rundtanks vorkommen. Vielfachechos können bei Pulsradar durch die Software erkannt werden, da sie zeitlich deutlich getrennt sind. Wie bereits in Kapitel 4 beschrieben, ist dies bei FMCW ein größeres Problem. Echokurve r/2 r Abb. 6.7: Die ideale Position für das Gerät ist bei Behältern mit gewölbtem Deckel bei der Hälfte des Radius. 120 Abb. 6.8: Dieser Effekt tritt auf, wenn das Gerät in der Spitze eines gewölbten Deckels montiert werden.
  • 126. 6. Installation Störechos Ebene Flächen, Einbauten z.B. Versteifungen oder auch Einbauten mit scharfen Kanten verursachen große Störechos. An diesen Objekten werden hohe Störamplituden produziert. Runde Profile hingegen produzieren eine diffuse Reflexion und somit nur geringe Störechos. Sie sind deshalb vom Gerät leichter zu verarbeiten als große Störechos, die von einer ebenen Fläche stammen. Können flache Reflexionsebenen im Messbereich des Radars nicht ver- mieden werden, sollten diese mit einem zur Seite ablenkenden Streublech versehen werden. Die dann mehrfach gebrochenen Radarsignale sind in der Amplitude deutlich kleiner und deshalb von der Software leichter zu verarbeiten. Diese Maßnahmen müssen umso gewissenhafter durchgeführt werden, je geringer der DK-Wert des Produkts ist und je höher die Genauigkeitsanforderungen sind. Abb. 6.9: Profile mit ebenen Flächen oder scharfen Ecken verursachen starke Störechos. Abb. 6.10: Durch diffuse Reflexion an runden Teilen werden deutlich geringere Störechos produziert. Abb. 6.11: Ein Streublech verteilt die Mikrowellenenergie zur Seite und reduziert damit die Störechoamplitude. 121
  • 127. Vermeiden vom Störechos Bei der Einbauposition des Radargerätes sollte darauf geachtet werden, dass sich keine Streben und kein Befüllstrom im Detektionsbereich des Radars befinden. Die folgenden Beispiele zeigen typische Messprobleme und wie sie vermieden werden können. Absätze Behälterprofile mit flachen Absätzen rechtwinklig zur Hauptstrahlrichtung des Radars erzeugen starke Störechos. Durch den Einbau eines Streublechs kann die Störechoamplitude deutlich reduziert werden, um somit eine zuverlässige Messung zu ermöglichen. Einbauten mit einer rechtwinkligen Fläche zum Sensor, z.B. Einlässe, Achsen, sollten mit einem „Dach“ versehen werden (Abb. 6.13). Hiermit wird das Radarsignal ebenfalls gestreut, die übrigen Störechos können von der Signalverarbeitungssoftware herausgefiltert werden. Abb. 6.12: Streublech an einem Absatz im Behälter. Abb. 6.13: Streublech auf Einbauten. 122
  • 128. 6. Installation Behältereinbauten Einbauten wie z.B. Streben, Leitern, Versteifungen und Sonden verursachen oft Störechos. Durch einen gute Wahl der Einbauposition können viele Störechos bereits im Vorfeld vermieden werden. Auch Schweißnähte im Behälter können Störechos produzieren. Speziell bei höherfrequenten Radargeräten, die nahe an der Wand montiert sind, können diese die Messung bei einem schlecht reflektierenden Produkt gefährden. Durch Anbringen von kleinen Blechen können diese Störechos verkleinert werden. Die Störamplitude sinkt und kann von der Signalverarbeitung besser verwertet werden. Bei der Herstellung des Behälters können Störechos durch Verschleifen der Schweißnähte minimiert werden. Abb. 6.14: Der Sensor sollte abseits von Einbauten, z.B. Leitern, montiert werden. Abb 6.15: Winkelbleche an Schweißnähten oder Versteifungen können Störechos reduzieren. 123
  • 129. Anhaftungen Ist der Radarsensor zu nahe an der Behälterwand montiert, können Produktanhaftungen Störechos erzeugen. Der Sensor sollte deshalb immer etwas Abstand zur Behälterwand haben. Der ideale Kompromiss ist ½ Radius. Abb. 6.16: Störechos durch Anhaftungen an der Behälterwand sollten vermieden werden. Polarisation Wie schon in Kapitel 2 besprochen, sind die Mikrowellen der VEGA Radargeräte linear polarisiert. Obwohl die Polarisation eine größere Bedeutung in Standrohren und Bypassrohren hat, kann sie auch bei Anwendung in „normalen“ Behältern von Bedeutung sein. Die Amplitude 124 von Störechos, z.B. von Streben oder der Behälterwand, kann oft durch Drehen des Radarsensors um 45º oder 90º reduziert werden. Die Richtung der Polarisation wird durch das Einkoppelsystem festgelegt, es ist am Gerät durch die Position des Typenschildes erkennbar.
  • 130. 6. Installation Ausrichtung des Radargerätes bei Flüssigkeitsanwendungen (Stab- oder Hornantenne) Bei Flüssigkeitsanwendungen muss Wird das Gerät angewinkelt, sinkt die das Radar-Füllstandmessgerät mögEchoamplitude und die Gefahr von lichst senkrecht nach unten zur zu Störechos wächst. messenden Oberfläche geführt werden. Abb. 6.17: Bei Messungen von Flüssigkeiten muss der Sensor senkrecht ausgerichtet sein. Fließende Produkte Ein Radarsensor sollte nicht direkt über oder in der Nähe einer Befüllung montiert werden. Dadurch wird ver- mieden, dass anstelle der Produktoberfläche der Befüllstrom gemessen wird. Abb. 6.18: Montieren Sie den Radarsensor abseits von Befüllströmen. 125
  • 131. Sensor zu nah an der Behälterwand Wird der Radarsensor zu nahe an der Behälterwand montiert, kann dies starke Interferenzen verursachen. Die Echos von Anhaftungen, Nieten oder Schweißnähten überlagern sich mit dem richtigen Echo. Es muss ausreichend Abstand vom Sensor zur Behälterwand eingehalten werden, um dies zu verhindern. Abhängig von der Antennengröße haben verschiedene Radarfüllstand- messgeräte unterschiedliche Öffnungswinkel (Kapitel 5: Radarantennen). Im Allgemeinen sollte darauf geachtet werden, dass sich die Behälterwand nicht innerhalb des 3dB-Öffnungswinkels der Antenne befindet. Bei ungünstigen Einbaubedingungen bzw. Störungen durch die Behälterwand können die Messverhältnisse durch Verändern der Polarisation optimiert werden. Abb. 6.19: Richtdiagramm einer Antenne mit 150 mm Durchmesser bei 6,3 GHz. 126
  • 132. 6. Installation 4. Standrohre und Bypassrohre Radar-Füllstandmessgeräte werden oft für Messungen in Standrohren oder, Bypassrohren eingesetzt. Diese Art von Installation kann bei Messungen mit Schaum, starken Turbulenzen, mechanisch komplexen Behältern oder bei Flüssigkeiten mit sehr niedrigem DKWert notwendig sein. Radar-Füllstandmessgeräte werden oft auch benutzt, um vorhandene Geräte in Rohren zu ersetzen, z.B. Verdränger und Schwimmer. Schaumbildung Ein dichter, leitfähiger Schaum auf dem Produkt kann die Messung stören. Unter diesen Bedingungen ist es wahrscheinlich, dass der Radarsensor die Oberfläche des Schaums messen wird. Es gibt aber auch Anwendungen mit Schaum geringer Dichte, der von Radarwellen problemlos durchdrungen wird. Allerdings kann hier keine generelle Aussage getroffen werden, deshalb muss bei Messungen mit Schaum stets mit Umsicht und Erfahrung vorgegangen werden. Lassen Sie sich bei solch einer Anwendung vom Sensorhersteller beraten. Flüssigkeiten mit sehr niedriger Dielektrizitätszahl Selbst nichtleitende Produkte und Flüssigkeiten mit äußerst niedriger Dielektrizitätszahl wie z.B. Flüssiggas können in Standrohren trotzdem genau und zuverlässig gemessen werden. Wie schon in Kapitel 5 erklärt, konzentriert das Standrohr die Mikrowellen und erzeugt so ein starkes Echo von der Produktoberfläche. Produkte mit Dielektrizitätszahlen bis zu 1,5 können so gemessen werden. Turbulente Produktoberfläche Starke Turbulenzen, verursacht durch Rührwerke oder heftige chemische Reaktionen, beeinflussen die Radarmessung. Ein Standrohr oder Bypassrohr mit hinreichender Größe erlaubt eine zuverlässige Messung sogar mit starken Turbulenzen im Behälter. Voraussetzung hierfür ist, dass das Produkt im Rohr nicht anhaftet. Leichte Anhaftungen verursachen jedoch in größeren Rohren, z.B. 100 mm Durchmesser, kaum Probleme. Allgemeine Hinweise zur Radarmessung in Rohren Ein Standrohr muss unten offen sein und sich über dem vollen Messbereich ausdehnen (d.h. von 0 % bis 100 % Füllstand). Zum Druckausgleich muss das Rohr über dem 100 % Punkt eine Bohrung besitzen. Ausgleichsbohrungen oder Schlitze müssen auf einer Achse liegen und dürfen maximal auf zwei gegenüberliegenden Seiten des Rohrs angebracht werden. Die Ausrichtung der Löcher zur Polarisation muss beachtet werden, bei VEGA-Sensoren müssen diese senkrecht unter dem Typschild angebracht sein. Als eine Alternative zum Standrohr im Gefäß kann ein Radarsensor auch außerhalb des Behälters auf einem Bypassrohr installiert werden. Die Polarisation muss wie in Abb. 6.21 dargestellt, zu den Prozessverbindungen ausgerichtet werden. 127
  • 133. E Abb. 6.20: Position von Entlüftungsbohrung und Polarisation auf einem Standrohr. E E Abb. 6.21: Polarisationsrichtung bei einem Bypassrohr. Abb. 6.22: Installation auf einem Bypassrohr. Radarsensoren können Verdrängersysteme und Schwimmer problemlos ersetzen. 128
  • 134. 6. Installation Polarisation Die Sensorpolarisation muss in einem Bypassrohr in Richtung der Prozessverbindungen und in einem Standrohr in Richtung der Ausgleichsbohrungen oder Schlitze ausgerichtet werden. Die Löcher oder Schlitze müssen auf einer Achse liegen. Eine korrekte Polarisation verbessert die Messung erheblich. Störechos werden dadurch reduziert und somit das Signal-Rausch-Verhältnis optimiert. Laufzeitänderung der Mikrowellen Wie bereits in Kapitel 2 und Kapitel 5 erklärt, reduziert sich in einem Standrohr, abhängig vom Durchmesser, der maximale Messbereich. Verursacht wird dies dadurch, dass sich die Mikrowellen im Rohr langsamer, als Lichtgeschwindigkeit ausbreiten. In einem Rohr mit 50 mm Durchmesser (2") verringert sich die Laufzeit um 20 % und die maximale Länge beträgt dadurch noch 16 m. Bei einem Rohr von 100 mm Durchmesser (4") reduziert sich die nutzbare Länge auf 19 m. Standrohr zur Messung von inhomogenen Produkten Abb. 6.23: Durch Schlitze wird eine gute Durchmischung von inhomogenen Produkten erreicht. Die Polarisation muss in Richtung der Schlitze ausgerichtet werden. 129
  • 135. Anhaftende Produkte Um Messprobleme und Messfehler bei der Messung von anhaftenden Produkten in Standrohren zu vermeiden, sollte das Rohr einen Innendurchmesser von mindestens 100 mm (4") haben. Sollen inhomogene Produkte oder Produkte gemessen werden die eine Trennschicht ausbilden, muss das Standrohr Löcher oder lange Schlitze haben. Diese Öffnungen stellen sicher, dass die Flüssigkeit durchmischt wird und, dass sie sich an den richtigen Füllstand angleicht. Je inhomogener das Produkt, desto mehr Öffnungen müssen vorhanden sein. Die Löcher und Schlitze müssen aus Gründen der Polarisation in zwei um 180º versetzten Reihen positioniert werden. Der Radarsensor muss so ausgerichtet werden, dass die Polarisation in Richtung der Löcher ausgerichtet ist. E Messrohr mit Kugelhahn Zur Abtrennung des Rohrs bzw. des Messgeräts vom Prozess kann ein Kugelhahn verwendet werden. Mit dem Kugelhahn ist es möglich, Wartungsarbeiten durchzuführen, ohne den Behälter zu öffnen. Dies ist bei Flüssiggas und giftigen Erzeugnissen besonders wichtig. Bei geöffnetem Ventil sollten möglichst keine Kanten im Durchlass zu sehen sein, dies würde sonst zu Störechos führen. E Abb. 6.25: Mit einem Kugelhahn kann der Radarsensor vom Behälter getrennt werden, ohne den Behälter zu öffnen, bzw. den Prozess zu stoppen. Abb. 6.24: Die Polarisation muss in Richtung der Schlitze oder Löcher ausgerichtet sein. 130
  • 136. 6. Installation Konstruktionsrichtlinien für Standrohre Diagramm 1 (Seite 132) Diagramm 2 (Seite 133) Für Messung in Stand- oder Bypassrohren werden Geräte mit Flanschgrößen DN50 (2"), DN80 (3"), DN100 (4") und DN 150 (6")benutzt. Diagramm 1 zeigt die Konstruktion eines Stand- oder Bypassrohrs mit einem Rohrdurchmesser und Flansch DN50. Das Standrohr muss innen glatt sein (Rauhigkeitswert Rz < 30). Ideal ist ein durchgehendes Rohr ohne Verbindungsstellen im Messbereich. Werden größere Rohrlängen benötigt, sollten die Teilstücke mit Vorschweißflanschen oder Rohrverschraubungen verbunden werden. Hierbei ist jedoch darauf zu achten, dass die Stoßstellen möglichst spaltfrei und ohne Durchmessersprung ausgeführt werden. Beim Schweißen darf kein Verzug entstehen, die Rohrstärke muss angepasst werden, um nicht durch das Rohr durchzuschweißen. Rauhigkeiten und Schweißnähte im Rohr müssen sorgfältig entfernt werden. Diese würden sonst Störechos verursachen und Anhaftungen begünstigen. Schlitze und Löcher müssen sorgfältig entgratet werden. Diagramm 2 zeigt die Konstruktion eines Standrohrs für einen Radarsensor mit einem DN100 (4") Flansch. Radarsensoren mit Flanschen von DN80 (3"), DN100 (4") und DN150 (6") müssen zur Messung im Standrohr mit einer Hornantenne ausgerüstet sein. Der Antennendurchmesser sollte hierbei möglichst nahe am Innendurchmesser des Rohrs liegen. Zur Messung in Rohren DN50 und DN80 sind spezielle Stabantennen vorhanden. Die Flanschverbindung zum Gerät ist nicht mehr kritisch, da sie hinter der Abstrahlebene der Antenne liegt. Bei starker Bewegung im Behälter (z.B. Rührwerk) muss das Standrohr entsprechend befestigt werden, dies gilt auch für sehr lange Rohre. Bei der Messung von Flüssigkeiten mit niedrigem DK-Wert kann oft der Nullpunkt nicht sicher gemessen werden, oder es kommt zu starken Messfehlern im Bodenbereich. Ausgelöst wird dies dadurch, dass das Echo des Behälterbodens hinter dem Rohrende ein stärkeres Echo erzeugt, als das Produkt selbst. In solchen Anwendungen kann der Einbau eines Streublechs am Ende des Rohrs von Vorteil sein. Die Mikrowellen werden hiermit zur Seite abgelenkt und das starke Bodenecho hierdurch vermieden. Allerdings geht dadurch am Rohrende Raum verloren, da außerhalb des Rohrs nicht gemessen werden kann. 131
  • 137. Diagramm 1 Radarsensor VEGAPULS 54 Flansch DN 50 Rohrdruchmesser 50 mm Vorschweißflansch 100% Schweißungs der Verbindungsmuffe 0.0…0.4 150…500 5…15 2.9…6 Verbindungsmuffe Rz ≤ 30 2.9 Schweißung des Vorschweißflansches 0.0…0.4 Vorschweißflansch 1.5…2 Löcher müssen gratfrei sein Halterung des Standrohres minimal messbare Füllhöhe (0%) 0% Ablenkplatte ~45˚ Alle Abmessungen in mm Abb. 6.26 132 Tankboden
  • 138. 6. Installation Diagramm 2 Radarsensor VEGAPULS 54 Flansch DN 100 Rohrdurchmesser 100 mm Schweißflansch Schweißung des Schweißflansches 100% 150…500 5…15 3.6 0.0…0.4 Schweißung der Verbindungsmuffe Verbindungsmuffe Rz ≤ 30 Vorschweißflansch 3.6 0.0…0.4 Schweißung des Vorschweiflansches 1.5…2 Löcher müssen gratfrei sein Halterung des Standrohres 0% AblenkPlatte minimal messbare Füllhöhe (0%) ~45˚ Behälterboden Alle Abmessungen in mm Abb. 6.27 133
  • 139. 5. Messung durch die Behälterwand und Radarfenster Die Mikrowellensignale von Radarfüllstandmessgeräten durchdringen dielektrische Materialien wie z.B. PTFE, Polypropylen und Glas. Dies ist für einige Anwendungen sehr wichtig, z.B. bei der Messung von hochreinen Flüssigkeiten in der Pharmaindustrie oder der Halbleiterfertigung, oder bei hochaggressiven Produkten in der chemischen Industrie. In diesen Fällen ist es aus Sicherheitsgründen und im Hinblick auf die Produktqualität von Vorteil wenn der Behälter geschlossen bleibt. Ein solche Messung ist bei Produkten mit guten Reflexionseigenschaften möglich, sie können bei geeignetem Behältermaterial direkt von oben, durch die Behälterdecke, gemessen werden. Produkte mit guter elektrischer Leitfähigkeit und mit einer Dielektrizitätszahl von mehr als 10 sind dafür geeignet. Bei Messungen in denen es prozess- oder produktbedingt zu starken Niederschlägen oder Kondensation an der Behälterdecke kommt, ist dieses Verfahren mit Vorsicht anzuwenden. Abb. 6.28: Gut reflektierende Medien können direkt durch die Behälterwand oder durch ein Messfenster gemessen werden. 134
  • 140. 6. Installation Reflexionen an der Behälterwand Wie Licht folgen auch Mikrowellen den Gesetzen der Reflexion. Obwohl bei geeignetem Behältermaterial der größte Teil der Energie durch die Behälterwand hindurch dringt, wird immer ein Teil dort reflektiert. Bei ebener Tankdecke und Aufsetzen des Radargerätes auf dem Tank wird dieser Teil der Sendeenergie direkt in die Antenne zurückreflektiert (Abb. 6.29). Dies führt zu erhöhtem Rauschen im Nahbereich. Abb. 6.29: Eine flache Behälterdecke produziert eine Störreflexion direkt zurück in die Antenne. Die Qualität der Messung wird verbessert, wenn das Radargerät über einem schrägen Bereich des Deckels (35º bis 50º) in einem Abstand von ca. 400 mm zum Behälter montiert wird. Der Winkel stellt sicher, dass die Reflexionen von der Tankwand nicht direkt in die Antenne strahlen und es somit nicht zu Störechos kommt. (Abb. 6.30) Abb. 6.30: Die Messung über einem angeschrägten Bereich des Behälterdeckels verbessert die Messung deut- Messung durch ein dielektrisches Fenster Mit einem Pulsradar kann auch durch „dielektrische Fenster“ in Metalltanks gemessen werden. Das Fenster muss groß genug und sollte im Idealfall auch angewinkelt sein. Auch hier sollte der Sensor auf Abstand zum Fenster montiert werden. Anmerkung: Prüfen Sie die Bestimmungen für den Einsatz von Radar-Füllstandmessgeräten außerhalb von geschlossenen Behältern in ihrem Land. Die geltenden Regeln können sehr unterschiedlich sein. Abb. 6.31: Optimale Installation für ein 6,3 GHz-Radar zur Messung durch ein dielektrisches Fenster. 135
  • 141. Messung durch ein dielektrisches Fenster In einigen Ländern ist es verboten Bei Messungen durch ein Fenster FMCW-Radar-Füllstandmessgeräte kann eine Verbesserung erzielt werden, außerhalb eines Metallgefäßes zu wenn die Scheibe eine konische Form betreiben. In solchen Fällen muss das erhält (siehe Abb. 6.32). Solch eine Gerät, um die Vorteile eines „dielekTrennscheibe kann bei geeigneter trischen Fensters“ nutzen zu können, in Dimensionierung als Linse wirken und einem metallischen Stutzen über einem die Mikrowellen zusätzlich fokusKunststoff oder Glasfenster installiert sieren. Diese Form begünstigt zusätzwerden (Abb. 6.32). Dies kann jedoch lich das Ablaufen und Abtropfen von einen hohen Störpegel verursachen. Kondensat. Radar-Sensor metallischer Stutzen konische Teflonscheibe Abb. 6.32 136
  • 142. 6. Installation Dimensionierung des dielektrischen Fensters Die Wahl der richtigen Material180º-Phasendrehung der Mikrowellen. dicke ist für die Messung durch ein Das zweite Echo, beim Verlassen des Fenster sehr wichtig. Fensters, besitzt keine Phasendrehung. Die entstehenden Interferenzen Hier geht es von einem dichteren in ein durch das Fenster bestehen aus zwei weniger dichtes Medium. Durch Wahl unterschiedlichen Echos. Das erste der Fensterdicke als λ/2 der Echo stammt von der äußeren Mikrowellenfrequenz löschen sich Oberfläche des Fenstermaterials, an der diese beiden Echos aus (siehe auch die Mikrowellen ins Fenster eindrinKapitel 2). gen. An dieser erste Oberfläche, dem Übergang von DK = 1 auf den DKWert des Fenstermaterials, gibt es eine Reflexion mit Phasendrehung von der Oberfläche Gesendete Welle Reflexion ohne Phasendrehung von der inneren Oberfläche D Kunststoffdeckel Sendesignal Reflexion mit Phasendrehung Reflexion ohne Phasendrehung { Gegenseitige Auslöschung Abb. 6.33: Die optimale Dicke des Fenstermaterials beträgt λ/2 der Radarfrequenz. 137
  • 143. Die Tabelle zeigt die optimale Dicke für die wichtigsten Kunststoffe und Gläser die zum Durchstrahlen geeignet sind. Es wird die optimale Dicke für 6,3 GHz und 26 GHz gezeigt. Fenstermaterialien für Radarsender: Frequenz 6,3 GHz zu durchdringendes Material PE Polyethylen PTFE (Teflon) PVDF Polyvinyl PP Polypropylen Borosylikat-Glas Rassotherm-Glas Labortherm-Glas Quarzglas POM Polyoxymethylen Polyester Plexiglas Polyacrylat PC Polycarbonat εr 2,3 2,1 ~7 2,3 5,5 4,6 8,1 ~4 3,7 4,6 3,1 ~2,8 optimale Dicke 15,5 16,5 9 15,5 10 11 8,5 12 12,5 11 13,5 14 D in mm (31; 46,5 …) (33; 49,5 …) (18; 27; 36 …) (31; 46,5 …) (20; 30; 40 …) (22; 33; 44 …) (17; 26,5; 34…) (24; 36; 48…) (25;37,5; 50 …) (22; 33; 44 …) (27; 40,5; 54 …) (28; 42 ...) Fenstermaterialien für Radarsender: Frequenz 26 GHz zu durchdringendes Material PE Polyethylen PTFE (Teflon) PVDF Polyvinyl PP Polypropylen Borosylikat-Glas Rassotherm-Glas Labortherm-Glas Quarzglas POM Polyoxymethylen Polyester Plexiglas Polyacrylat PC Polycarbonat εr 2,3 2,1 ~7 2,3 5,5 4,6 8,1 ~4 3,7 4,6 3,1 ~2,8 optimale Dicke 3,8 4 1,8 3,8 2,5 2,7 2 2,9 3 2,7 3,2 3,6 D in mm (7,6; 11,4 ...) (8,0; 12,0 ...) (3,6; 5,4 ...) (7,6; 11,4 ...) (5; 7,5 …) (5,4; 8,1 …) (4,0; 6,0; 8,0 …) (5,8; 8,7 …) (6,0; 9,0 ...) (5,4; 8,1 ...) (6,4; 9,6 ...) (7,2; 10,8 ...) Anmerkung: Die optimale Dicke kann auch durch Aufschichten einiger Lagen identischen Materials erreicht werden. Die Schichten müssen jedoch ohne Luftspalt aufeinander liegen. Vielfache der optimalen Dicke führen ebenfalls zu guten Ergebnissen, jedoch verursacht die Dicke des Fenstermaterials eine Signaldämpfung. 138
  • 144. 6. Installation 6. Messung von Schüttgütern mit Hornantennen Zur Messung von Schüttgütern werden fast ausschließlich Hornantennen verwendet. Dies schließt alle pneumatisch beförderten Erzeugnisse wie Pulver, Granulate und Körner ein. Die Stabantenne hat ihre Stärke in Flüssigkeitstanks. Die Oberflächen von Schüttgütern in Silos und Behältern sind selten flach. Bei Produkten wie z.B. Pulver oder Granulat sieht das Profil bei Befüllung und Entleerung zumeist unterschiedlich aus. Der Winkel des Schüttkegels hängt vom Produkt selbst, der Füll- und Entleermethode und von Form und Abmessungen des Silos ab. Radar-Füllstandmessgeräte, ebenso wie Ultraschallwandler, sollten außerhalb der Mitte zum tiefsten Punkt des Behälters ausgerichtet montiert werden. Auch hier sollte das Ende des Horns mindestens 10 mm in den Behälter ragen. Der Radarsensor wird angewinkelt montiert um immer möglichst senkrecht zur Produktoberfläche zu senden. So wird über die gesamte Füllhöhe die beste Echoamplitude erreicht. Der Radarsensor sollte abseits vom Befüllstrom und von Einbauten montiert werden um möglichst wenig Störechos zu erhalten. Abb 6.34: Für Schüttgutanwendungen werden Hornantennen verwendet. Die Antenne ist außerhalb der Mitte montiert und zum tiefsten Punkt im Silo ausgerichtet. Dies ergibt bei verschiedenen Schüttkegeln das beste Messergebnis. 139
  • 145. Abb. 6.35 und 6.36: Schüttkegel von typischen Schüttgutanwendungen beim Befüllen und Entleeren. Hohe Temperaturen und anhaftende Produkte Bei Anwendungen mit hohen TemHierzu wird der Flansch von zwei peraturen oder stark anhaftenden gegenüberliegenden Seiten bis zum Staubablagerungen auf der Antenne Konus der Teflonfüllung durchbohrt. sollte diese mit Druckluft oder StickAn diesen Stellen kann dann die Luftstoff gespült werden. bzw. Stickstoffspülung angeschlossen werden. Abb. 6.37: Luft- bzw. Stickstoffspülung zum Kühlen und Reinigen der Antenne. Luft- bzw. Stickstoff 140
  • 146. 6. Installation B. Elektrische Anschlussvarianten In den vergangenen Jahren hat sich die Auswahl an unterschiedlichen Radar-Füllstandmessgeräten erhöht. Zudem haben sich eine Vielzahl von elektrischen Anschlussmöglichkeiten für Standard- und Ex-Anwendungen auf dem Markt etabliert. Diese umfassen 4 … 20 mA- und verschiedene Feldbussensoren. Bei der Auswahl eines Radarsensors müssen die entsprechenden Verkabelungskosten berücksichtigt werden. Seit ihrer Markteinführung haben sich eigensichere ZweileiterRadarsensoren als vollwertiger Ersatz für traditionelle Sensoren wie z.B. Differenzdruckmessumformer oder Verdränger durchgesetzt. FMCWRadarsensoren benötigen jedoch noch immer die erhöhte Energie aus einer Vierleiterversorgung. In diesem Abschnitt werden die möglichen Beschaltungskonfigurationen für alle Arten von Radar betrachtet. 1. Nicht-Ex-Anwendungen a. 4 … 20 mA, Zweileiter-Radarsensor 4 … 20 mA, 24 VDC Abb. 6.38 b. Vierleiter-Radarsensor mit 4 … 20 mA Stromausgang 20/250 VAC / VDC 4 … 20 mA Abb. 6.39 c. HART®-Protokoll Die meisten Zweileiter- und Vierleiter-, 4 … 20 mA Radar-Füllstandmessgeräte sind mit dem HART®-Protokoll, aufmoduliert auf dem Stromsignal, verfügbar. Dadurch wird Folgendes möglich: - Fernparametrierung mit dem HART®-Handheld Programmiergerät - Einspeisung der HART®-Daten direkt in das Prozessleitsystem - multi-drop Betrieb mit bis zu 16 Sensoren parallel an einem Strang 141
  • 147. 142 verschiedene Industrie-StandardKommunikationen VEGALOG 571 mit bis zu 255 Sensoren VBUS Abb. 6.40 bis zu 15 Sensoren an einer Zweidrahtleitung d. Feldbus (VBUS) bis zu 15 Sensoren parallel auf zwei Drähten mit VEGALOG 571 und EV-Eingangskarten maximal 255 Messungen zusammenfassbar
  • 148. Segmentkoppler Profibus DP Profibus PA e. Feldbus (Profibus PA) max. 32 Sensoren gemeinsam an einem Segmentkoppler Abb. 6.41 6. Installation 143
  • 149. 2. Geräte für Ex-Anwendungen a. eigensicher ia, 4 … 20 mA, Zweileiter-Sensoren mit HART®-Protokoll Ex ia 4 … 20 mA, 24 VDC Zenerbarriere Ex-Bereich Nicht-Ex-Bereich Abb. 6.42 b. 4 … 20 mA, Zweileiter-EEx-d-ia Sensoren mit Verkabelung in erhöhter Sicherheit. - Versorgung 12 bis 36 VDC - Zener-Barriere in integriertem Ex-d Gehäuse, eigensicherer Gehäuseteil für Sensorelektronik und zur Bedienung - keine zusätzliche Trennbarriere erforderlich Ex ia Ex d Bedienung, Display und Elektronik eigensicher ausgeführt Ex-Bereich Abb. 6.43 144 4 … 20 mA, 24 VDC Ex e Zener barrier Nicht-Ex-Bereich
  • 150. 6. Installation c. Vierleiter, EEx d ia Versorgung - Versorgung 24 VDC - eigensicherer 4 … 20 mA Stromausgang Ex d 4 … 20 mA 24 VDC, Ex e eigensicher Zener barrier Nicht-Ex-Bereich Ex-Bereich Abb. 6.44 d. Vierleiter, 4 … 20 mA, Ex e Versorgung- Exd-Gehäuse Ex d 24 VDC, Ex e 4 … 20 mA Nicht-Ex-Bereich Ex-Bereich Abb.6.45 e. Vierleiter eigensicher (ib) mit Trennübertrager und Datenkoppler Display oder Signalverarbeitungseinheit Stromversorgung Ex d Stromversorgung & digitale Kommunikation 4 … 20 mA Nicht-Ex-Bereich Ex-Bereich Abb.6.46 145
  • 151. 146 verschiedene IndustrieStandard-Kommunikationen VEGALOG 571 mit bis zu 255 Sensoren Separate Spannungsversorgung Ex e Abb 6.47 f. Feldbus (VBUS) - max. 15 Sensoren an zwei Leitungen in Ex e, Verdrahtung in erhöhter Sicherheit - separate Energieversorgung der Sensoren in erhöhter Sicherheit VBUS bis zu 15 Sensoren an einer Zweidrahtleitung, Ex e
  • 152. verschiedene IndustrieStandard-Kommunikationen VEGALOG 571 Ex e Verkabelung Abb. 6.48 Fünf Sensoren an jeder Zweidrahtleitung versorgt durch diese Leitung g. Feldbus (VBUS) - max. 15 Sensoren Ex-e (je fünf Sensoren pro Strang, drei Stränge) pro VBUS-Karte - Verdrahtung in erhöhter Sicherheit ohne externe Versorgung VBUS 6. Installation 147
  • 153. 148 verschiedene IndustrieStandard-Kommunikationen VEGALOG 571 VBUS VBUS Fünf Sensoren an einer Zweidrahtleitung eigensicher Abb. 6.49 h. Feldbus (VBUS) - max. 15 Sensoren, eigensicher Ex ia, pro Ausgangskarte - je 5 Sensoren pro Zweileiter- Schleife, max. 3 Schleifen pro EV-Karte
  • 154. Profibus DP Segmentkoppler Profibus PA i. Feldbus (Profibus PA) - Ex ia eigensicher, max. 8 Sensoren pro Zweiader-Schleife - Verbindung über Segmentkoppler zu Profibus DP Abb. 6.50 Acht Sensoren an einer Zweidrahtleitung eigensicher 6. Installation 149