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Fundamentals of the RF Transmission and Reception of Digital Signals

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Part 1 of this deck discusses Digital Modulation and Part 2 focuses on Digital Demodulation. By Analog Devices, Inc.

Part 1 of this deck discusses Digital Modulation and Part 2 focuses on Digital Demodulation. By Analog Devices, Inc.

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  • The blue arrows indicate possible phase transitions. Note that 180° phase changes (through center of constellation in QPSK case) cause the envelope of the RF carrier waveform to go to zero for an instant. Also the instantaneous phase transitions result in a very wide bandwidth signal, normally the signal will need to be filtered (i.e. raised cosine) to suppress the sidelobes of the sin(x)/x response due to the digital BB square waves. However, the filtering causes a non-constant amplitude of the QPSK signal. This now requires a linear power amplifier , while for the unfiltered QPSK signal a nonlinear power amplifier would have sufficed. This is an example of the trade-off between spectral and power efficiency . Furthermore, if the required linear PA is not perfect, the non-linearity causes the spectrum to widen again - this is called spectral re-growth . The effect of this is an increase in adjacent channel power which causes ACI in a neighboring channel. One modified version of the QPSK modulation that avoids 180° phase changes is the  /4-QPSK modulation; it has at most 135° phase changes. Therefore, a less linear (i.e. more efficient) PA can be used. Another class of modulation schemes that is even more power efficient than the QPSK family are MSK (Minimum Shift Keying) signals. However, MSK signals require larger bandwidths than QPSK waveforms. But they belong to the class of constant envelope modulations which allow the use of highly efficient, non-linear Class C power amplifiers. GMSK is one popular example of this class of modulation schemes. NOTE : To increase the data rate for a fixed channel BW in a system like GSM, requires that the modulation needs to be changed; i.e. the constant envelope of GMSK won’t hold anymore. From the discussion above it should be obvious that now linear PAs are required !!! Therefore in the move from GSM to GSM EDGE, parts that can be helpful in PA linearization , l ike the AD8302, AD8347, AD8313 are in high demand.
  • 04/12/12
  • This analysis will help to understand how DC offsets are generated. Suppose Leakage from LO port to point A is 60dBc. There is finite leakage from LO to point B. The S12 of the LNA would somewhat determine leakage at point B. Also there is some direct leakage from LO port to point A. The -70dBm is the desired signal level. Gain from A to C is 30dB. Total Gain from Antenna to X is ~ 60dB to have some signal amplitude at ADC input. Therefore, the remaining gain (after point C) amplifies the offset voltage to saturate the following stages, prohibiting amplification of the desired signal.
  • 04/12/12
  • - VGA used to compensate for limited SFDR of low resolution ADCs. This becomes critical in the presence of a large blocker signal. The use of higher resolution ADCs may negate the need for a VGA function. - Accurate amplitude and phase conformance on I and Q channels is essential for high image rejection.
  • Transcript

    • 1. The World Leader in High-Performance Signal Processing Solutions FUNDAMENTALS OF THE RFTRANSMISSION AND RECEPTION OF DIGITAL SIGNALS
    • 2. The World Leader in High-Performance Signal Processing Solutions Part 1: Digital Modulation2
    • 3. Transmitting Bits Bit Stream 1 1 -1 1 1 -1 -1 -1 1 1 Bits Divide into 1 1 -1 1 1 -1 -1 -1 1 1 Symbols (2 bits per Symbol) -135° 45° Assign Phase 45° 135° -45° to Symbols 135° 45° Modulate Phases on to Carrier -135° -45°3
    • 4. Practical Digital Modulation using an IQ Modulator Filtered Bit Stream I IN 0 LO RF OUT LO (from PLL) 90 Q IN Filtered Bit Stream Looks like Amplitude Modulation (AM) but this signal is indeed phase modulated. Why the amplitude variations? Phase Splitter separates LO from PLL into “Quadrature” components of equal amplitude but 90 degrees out of phase Filtered bit streams from a dual DAC drive the I and Q inputs which are multiplied with the quadrature LOs The outputs of the two multipliers are combined to yield the modulated carrier This modulation coding scheme is called Quadrature Phase Shift Keying (QPSK) 4
    • 5. IQ Modulation in the Frequency Domain I IN 3 dB BW=Symbol Rate 0 LO RF OUT FLO 90 FLO Q IN 3 dB BW=Symbol Rate/2 I and Q baseband signals are mixed up to an IF or to RF. Modulated carrier bandwidth is twice the baseband bandwidth5
    • 6. Other Digital Phase Modulation Schemes m=2, n=1 m=4, n=2 m=8, n=3 BPSK – 1 bit/symbol QPSK- 2 bits/symbol 8-PSK – 3 bits/symbol m=16, n=4 m=64, n=8 64 QAM – 6 bits/symbol 16 QAM – 4 bits/symbol By allowing more I and Q levels (beyond -1 and +1), we can implement higher order QAM modulation schemes. Higher Order Modulation Schemes → Higher Data Rate. But Symbols are closer together → Requires higher Signal-to-Noise Ratio for demodulation Increasing “Symbol Rate” increases data rate but widens Spectrum6
    • 7. Error Vector Magnitude - EVMQ Magnitude Error (I/Q error mag) { Actual M ∑ Z (k ) − R(k ) 2 Signal EVM = k =1 M Unit = % 2 ∑ R(k ) k =1 φ Ideal (Reference) Signal Phase Error (I/Q error phase) I Noise and Imperfections in transmit and receive signal chains result in demodulated voltages which are displaced from their ideal location. Error Vector Magnitude expresses this dislocation Large EVM will result in Symbol Errors and degraded Bit Error Rate Higher Order Modulation Schemes → Symbols Closer Together → EVM More Critical7
    • 8. The Imperfect IQ Modulator Gain Imbalance IQ MOD (G1,G2,G3,G4) IIN Vofs1 Degrades G3 EVM Imbalance G1 In Phase 0 Splitter Vn 89.5 Degrades EVM G2 Q IN Vofs2 Noise risks G4 violation of Offset emissions Voltages LOIN regulations Cause LO Leakage to RFOUT8
    • 9. Dealing with IQ Modulator Imperfections DAC incorporates Gain, Phase and Offset Voltage adjustment functions DAC and IQ Modulator have matching bias levels (0.5 V), permitting a glue-less interface with no level shifting requirements Modulator correction functions can also be performed in the digital domain9
    • 10. How Distortion Impacts Transmitters Marker 1 [T1] RBW 30 kHz RF Att 20 dB Ref Lvl -10.73 dBm VBW 300 kHz -10 dBm 99.48897796 MHz SWT 84 ms Unit dBm 1 -10 1 [T1] -10.73 dBm A 99.48897796 MHz -20 CH PWR 8.11 dBm ACP Up -58.77 dB -30 ACP Low -59.27 dBACLR=58 dBc -40 1RM Adjacent -50 Channel -60 Leakage -70 Ratio Caused -80 By poor IMD -90 C0 C0 cl1 -100 cl1 cu1 cu1 -110 Center 100 MHz 3 MHz/ Span 30 MHz Date: 24.FEB.2006 12:00:50  No Blockers to worry about in Transmitter.  But excessive distortion creates Spectral Leakage into adjacent channels  Distortion can be caused by any component in the signal chain, not just the modulator10
    • 11. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -79.38 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 [T1] -79.38 dBm A 1.95950000 GHz -40 CH PWR -53.44 dBm ACP Up -41.74 dB -50 ACP Low -41.71 dB -60 1AVG 1RM -70 1 -80 -90 -100 ADJACENT MAIN ADJACENT CHANNEL CHANNEL CHANNEL -110 C0 C0 cl1 -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz11 Date: 9.NOV.2009 18:36:37
    • 12. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -60.22 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 [T1] -60.22 dBm A 1.95950000 GHz -40 CH PWR -35.08 dBm ACP Up -60.05 dB -50 ACP Low -60.01 dB 1 -60 1AVG 1RM -70 -80 -90 -100 ADJACENT MAIN ADJACENT CHANNEL CHANNEL CHANNEL -110 C0 C0 cl1 -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz12 Date: 9.NOV.2009 18:33:38
    • 13. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -33.52 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 1 [T1] -33.52 dBm A 1.95950000 GHz -40 CH PWR -8.92 dBm ACP Up -68.55 dB -50 ACP Low -71.69 dB -60 1AVG 1RM -70 -80 -90 -100 ADJACENT MAIN ADJACENT CHANNEL CHANNEL CHANNEL -110 C0 C0 cl1 -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz13 Date: 9.NOV.2009 18:10:08
    • 14. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -42.87 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 [T1] -42.87 dBm A 1.95950000 GHz -40 1 CH PWR -17.67 dBm ACP Up -73.47 dB -50 ACP Low -74.75 dB -60 1AVG 1RM -70 -80 -90 -100 -110 C0 C0 cl1 -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz14 Date: 9.NOV.2009 18:12:07
    • 15. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -36.78 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 1 [T1] -36.78 dBm A 1.95950000 GHz -40 CH PWR -11.53 dBm ACP Up -72.85 dB -50 ACP Low -74.71 dB -60 1AVG 1RM -70 -80 -90 -100 -110 C0 C0 cl1 -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz15 Date: 9.NOV.2009 19:14:23
    • 16. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -33.52 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 1 [T1] -33.52 dBm A 1.95950000 GHz -40 CH PWR -8.92 dBm ACP Up -68.55 dB -50 ACP Low -71.69 dB -60 1AVG 1RM -70 -80 -90 -100 -110 C0 C0 cl1 -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz16 Date: 9.NOV.2009 18:10:08
    • 17. What is happening here? 50 * Intercept SLOPE=1 of 0 Fundamentals Fundamentals and * * * Intermods * * IMD(dBc) (IP3) -50 SLOPE=3 *-100 * * Intermods * *-150 -20 -10 0 10 20 30 40 50  OIP3 Intercept(dBm) = PFUND – (IMD/2)  Knowing the OIP3 allows you to calculate Intermodulation Distortion (IMD) at any power level17  Many devices do not follow this rule
    • 18. Striking a Balance Poor SNR Excessive Distortion Marker 1 [T1] RBW 10 kHz RF Att 0 dB Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -33.52 dBm VBW 100 kHz Ref Lvl -79.38 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm -30 1 -30 1 [T1] -79.38 dBm 1 [T1] -33.52 dBm A A 1.95950000 GHz 1.95950000 GHz -40 -40 CH PWR -53.44 dBm CH PWR -8.92 dBm ACP Up -41.74 dB ACP Up -68.55 dB -50 ACP Low -41.71 dB -50 ACP Low -71.69 dB -60 -60 1AVG 1RM 1AVG 1RM -70 -70 1 -80 -80 -90 -90 -100 -100 -110 C0 -110 C0 C0 cl1 C0 -120 cl1 cl1 cu1 -120 cl1 cu1 cu1 -130 cu1 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHz Date: 9.NOV.2009 18:36:37 Date: 9.NOV.2009 18:10:08 We need to set our gains and levels so that we can strike a balance between SNR and Distortion This is why our customers simultaneously demand low noise and low distortion Gain is generally distributed throughout the channel to achieve this goal18
    • 19. Last Word on Distortion….. Marker 1 [T1] RBW 10 kHz RF Att 0 dB Ref Lvl -42.87 dBm VBW 100 kHz -30 dBm 1.95950000 GHz SWT 370 ms Unit dBm •During an IP3 -30 1 [T1] -42.87 dBm A sweep, at a certain -40 1 1.95950000 GHz CH PWR -17.67 dBm power level, the ACP Up -73.47 dB power of the IMD -50 ACP Low -74.75 dB tones will be equal -60 to the noise power 1AVG 1RM in a defined -70 Spurious bandwidth. The SNR -80 Free at this point is the Dynamic SFDR of the -90 Range component -100 •Don’t mix this up -110 with the SFDR of an C0 C0 cl1 ADC or DAC -120 cl1 cu1 cu1 -130 Center 1.96 GHz 1.46848 MHz/ Span 14.6848 MHzSFDR = (2/3)(IIP3-NF-10log(kTB)) Date: 9.NOV.2009 18:12:0719
    • 20. Key IQ Modulator Specifications Input IP3 (IIP3): Same as OIP3 but referred to input: Intermodulating Blockers can create IMD products that fall on the desired signal Noise Figure IP2: Figure of Merit for Second order Intermodulation Distortion. Poor IP2 can intermodulate with the desired signal and produce dc offsets LO Quadrature accuracy: Affects EVM/BER of recovered data20
    • 21. I/Q Modulator Key specifications Part Freq LO Sideband Noise P1dB OIP3 Specs P/N Isy Desc Vs(V) Package Number (MHz) (dBm) (dBc) (dBm/Hz) (dBm) (dBm) @ (MHz) dBc/Hz (mA) 5.1×6.4 AD8345 140-1000 Low Power I/Q Mod -42 -42 -154.5 2.5 25 800 N/A 2.7-5.5 65 TSSOP-16 5.1×6.4 AD8346 800-2500 Low Power I/Q Mod -42 -36 -147 -3 20 1900 N/A 2.7-5.5 45 TSSOP-16 5.1x6.4 AD8349 700-2700 Low Power I/Q Mod -45 -35 -155 7.6 21 900 N/A 4.75-5.5 135 TSSOP-16 7X7ADF9010 840-960 IQ Mod & Int-N PLL -40 -46 -158 10 24 900 -83 3.15-3.45 360 LFCSP-48 4×4ADL5370 300-1000 Narrowband IQ Mod -50 -41 -160 11.0 24 450 N/A 4.75-5.25 205 LFCSP-24 4×4ADL5371 500-1500 Narrowband IQ Mod -50 -55 -158.6 14.4 27 900 N/A 4.75-5.25 175 LFCSP-24 4×4ADL5372 1500-2500 Narrowband IQ Mod -45 -45 -158 14.2 27 1900 N/A 4.75-5.25 165 LFCSP-24 4x4ADL5373 2300-3000 Narrowband IQ Mod -32 -57 -157.1 13.8 26 2500 N/A 4.75-5.25 174 LFCSP-24 4×4ADL5374 3000-4000 Narrowband IQ Mod -32.8 -50 -159.6 12.0 22.8 3500 N/A 4.75-5.25 173 LFCSP-24 4×4ADL5375 400-6000 IQ Mod w Output Disable -46.2 -52.1 -160 9.4 26.8 900 N/A 4.75-5.25 200 LFCSP-24 4×4ADL5385 50-2200 2XLO Broadband IQ Mod -46 -50 -159 11.0 26 350 N/A 4.75-5.5 215 LFCSP-24 6×6ADL5386 50-2200 2XLO IQ Mod & VVA&AGC -38 -46 -160 11.1 25 350 N/A 4.75-5.5 230 LFCSP-40 6x6ADRF6701 750-1100 IQ Mod & Frac-N PLL&VCO -45 -40 -158 14 29 900 -93 4.75-5.25 260 LFCSP-40 6x6ADRF6702 1550-2150 IQ Mod & Frac-N PLL&VCO -40 -33 -158 14 26 1800 -90 4.75-5.25 260 LFCSP-40 6x6ADRF6703 2100-2600 IQ Mod & Frac-N PLL&VCO -40 -40 -158 15 33 2200 -93 4.75-5.5 260 LFCSP-40 6x6ADRF6704 2500-2900 IQ Mod & Frac-N PLL&VCO -41 -40 -158 15 31 2600 -92 4.75-5.5 260 LFCSP-40 8X8ADRF6750 950-1575 IQ Mod & Frac-N PLL&VCO -45 -45 -157 8.5 21 1200 -93 4.75-5.25 310 LFCSP-56 21
    • 22. The World Leader in High-Performance Signal Processing Solutions Part 2: Digital Demodulation22
    • 23. Recovering Data from a Digitally Modulated Carrier Iout 0 VREF 90 Qout 70 MHz VREF Comparators (real systems use Dual ADCs) 70 MHz Sine Wave  Reverse process to IQ Modulation  IQ Demodulator extracts phase (and amplitude) information from the modulated signal and presents it in XY (or IQ) format.  Apply I and Q outputs to an ADC or Comparator and bits can be recovered.23
    • 24. Critical IQ Demodulator Specs – LO to RF Leakage -60dBm -30dBm(~20mVp-p) -40dBm FLO ω A B C LNA ADC -70dBm Leakage Desired 0dBm Assume, ω Gain from A to C =30dB FLO LO to RF leakage ~ 60dB •If some of the LO leaks to the RF input, it mixes (multiplies) with itself in the mixer generating unwanted dc offsets on top of the recovered baseband data stream 24
    • 25. What is causing the poor quality of this demodulated Constellation? Symbol Decision Threshold If the symbol lands on the edge or outside of the box, bit errors will occur  Very poor LO Quadrature Phase Split (in DMOD)  Dc Offset of the complete constellation (probably LO to RF Leakage)  Noise has enlarged the footprint of the constellation points (poor Receiver Noise Figure)25
    • 26. Reading the Demodulated Constellation Signal Compression (signal chain is being over driven)
    • 27. Key IQ DMOD Specifications Input IP3 (IIP3): Same as OIP3 but referred to input: Intermodulating Blockers can create IMD products that fall on the desired signal Noise Figure IP2: Figure of Merit for Second order Intermodulation Distortion. Poor IP2 can intermodulate with the desired signal and produce dc offsets LO Quadrature accuracy: Affects EVM/BER of recovered data27
    • 28. IQ Demodulators VGA IQ Quadrature Noise Freq P1dB IIP3 Specs Isy VSPart No. Range 3dB BW Error Figure Package (MHz) (dBm) (dBm) @(MHz) (mA) (V) (dB) (MHz) (dB/deg) (dBm) 9.7x6.4 AD8347 800-2700 70 65 0.3/1º -2 +11.5 11 1900 64 2.7-5.5 TSSOP-28 9.7x6.4 AD8348 50-1000 44 125 0.25/0.5º +13 +28 10.75 380 48 2.7-5.5 TSSOP-28 4X4ADL5382 700-2700 N/A 370 0.05/0.2º 14.4 30.5 15.6 1900 220 4.75-5.25 LFCSP-24 4X4ADL5387 50-2000 N/A 240 0.05/0.2º +13 +31 12 140 180 5 LFCSP-24 4X4ADL5380 400-6000 N/A 390 0.07/0.25º 11.6 27.8 11.7 1900 245 5 LFCSP-24 8X8ADRF6850 100-1000 60 300 0.1/0.5º 12 22.5 11 800 350 3.15-3.45 LFCSP-56 28
    • 29. Application Example – Complete Direct Conversion Receiver  Direct Conversion Receiver has no IFs and no IF Filters  Variable gain after IQ DMOD is used to optimize the peak-to- peak swing of the signal for the ADCs29
    • 30. Receiver EVM vs Input power using ADF4350 PLL/VCO as LO source -10 -15Modulation Error Rate- using ADF4350 -20 PLL/VCO as LO source MER-dB -25 -30 -35 -40 -90 -80 -70 -60 -50 -40 -30 -20 Input Power (dBm)30
    • 31. An IQ DMOD-based Receiver Filtersand Amplifiers amplify signal and remove out-of-band blockers Variable gain after IQ DMOD is used to optimize the peak-to-peak swing of the signal for the ADCs When the input frequency to the IQ Modulator is also the receive frequency, we have a Direct Conversion Receiver (Zero IF) 31
    • 32. AD8348 IQ Demodulator with Integrated VGA Built-in VGA has 45 dB of gain control range VGA will still require external circuitry to implement AGC32

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